LTC3890IGN-1

FEATURES n n n n n n n n n n n n n n n n LTC3890-1 60V Low IQ, Dual, 2-Phase Synchronous Step-Down DC/DC Controller DESCRIPTION The LTC®3890-1 is a high performance dual step-down switching regulator DC/DC controller that drives all Nchannel synchronous power MOSFET stages. A constant frequency current mode architecture allows a phase-lockable frequency of up to 850kHz. Power loss and supply noise are minimized by operating the two controller output stages out-of-phase. The 50μA no-load quiescent current extends operating life in battery-powered systems. OPTI-LOOP® compensation allows the transient response to be optimized over a wide range of output capacitance and ESR values. A wide 4V to 60V input supply range encompasses a wide range of intermediate bus voltages and battery chemistries. Independent TRACK/SS pins for each controller ramp the output voltages during start-up. Current foldback limits MOSFET heat dissipation during short-circuit conditions. The PLLIN/MODE pin selects among Burst Mode operation, pulse skipping mode, or continuous conduction mode at light loads. For a leadless 32-pin QFN package with additional features of adjustable current limit, clock out, phase modulation and two PGOOD outputs, see the LTC3890 data sheet. L, LT, LTC, LTM, Linear Technology, Burst Mode, OPTI-LOOP and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents including 5481178, 5705919, 5929620, 6100678, 6144194, 6177787, 6304066, 6580258, 7230497. Wide VIN Range: 4V to 60V (65V Abs Max) Low Operating IQ: 50μA (One Channel On) Wide Output Voltage Range: 0.8V ≤ VOUT ≤ 24V RSENSE or DCR Current Sensing Out-of-Phase Controllers Reduce Required Input Capacitance and Power Supply Induced Noise Phase-Lockable Frequency (75kHz to 850kHz) Programmable Fixed Frequency (50kHz to 900kHz) Selectable Continuous, Pulse Skipping or Low Ripple Burst Mode® Operation at Light Loads Very Low Dropout Operation: 99% Duty Cycle Adjustable Output Voltage Soft-Start or Tracking Power Good Output Voltage Monitor Output Overvoltage Protection Low Shutdown IQ: 8V .......................................................100μA SENSE1+, SENSE2+, SENSE1– SENSE2– Voltages...................................... –0.3V to 28V PLLIN/MODE, FREQ Voltages .............. –0.3V to INTVCC EXTVCC ...................................................... –0.3V to 14V ITH1, ITH2, VFB1, VFB2 Voltages ................... –0.3V to 6V PGOOD1 Voltage ......................................... –0.3V to 6V TRACK/SS1, TRACK/SS2 Voltages .............. –0.3V to 6V Operating Junction Temperature Range (Note 2) ................................................. –40°C to 125°C Maximum Junction Temperature (Note 3) ............ 125°C Storage Temperature Range................... –65°C to 150°C SENSE2+ 11 VFB2 12 ITH2 13 TRACK/SS2 14 GN PACKAGE 28-LEAD PLASTIC SSOP TJMAX = 125°C, θJA = 90°C/W ORDER INFORMATION LEAD FREE FINISH LTC3890EGN-1#PBF LTC3890IGN-1#PBF TAPE AND REEL LTC3890EGN-1#TRPBF LTC3890IGN-1#TRPBF PART MARKING* LTC3890GN-1 LTC3890GN-1 PACKAGE DESCRIPTION 28-Lead Plastic SSOP 28-Lead Plastic SSOP TEMPERATURE RANGE –40°C to 125°C –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS SYMBOL VIN VFB1,2 PARAMETER Input Supply Operating Voltage Range Regulated Feedback Voltage The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TJ = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted. CONDITIONS ITH1,2 Voltage = 1.2V (Note 4) –40°C to 125°C –40°C to 85°C (Note 4) VIN = 4.5V to 60V (Note 4) (Note4) Measured in Servo Loop, ΔITH Voltage = 1.2V to 0.7V (Note4) Measured in Servo Loop, ΔITH Voltage = 1.2V to 2V l MIN 4 l TYP MAX 60 UNITS V V V nA %/V % 0.788 0.792 0.800 0.800 ±5 0.002 0.01 0.812 0.808 ±50 0.02 0.1 IFB1,2 VREFLNREG VLOADREG Feedback Current Reference Voltage Line Regulation Output Voltage Load Regulation l –0.01 –0.1 % 38901f 2 LTC3890-1 ELECTRICAL CHARACTERISTICS SYMBOL gm1,2 IQ PARAMETER Transconductance Amplifier gm Input DC Supply Current Pulse Skip or Forced Continuous Mode (One Channel On) Pulse Skip or Forced Continuous Mode (Both Channels On) Sleep Mode (One Channel On) The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TJ = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted. CONDITIONS ITH1,2 = 1.2V, Sink/Source = 5μA (Note 4) (Note 5) RUN1 = 5V and RUN2 = 0V or RUN1 = 0V and RUN2 = 5V, VFB1 = 0.83V (No Load) RUN1,2 = 5V, VFB1,2 = 0.83V (No Load) RUN1 = 5V and RUN2 = 0V or RUN1 = 0V and RUN2 = 5V, VFB1 = 0.83V (No Load) RUN1,2 = 5V, VFB1,2 = 0.83V (No Load) RUN1,2 = 0V INTVCC Ramping Up INTVCC Ramping Down Measured at VFB1,2, Relative to Regulated VFB1,2 Each Channel Each Channel VSENSE– < INTVCC – 0.5V VSENSE– > INTVCC + 0.5V In Dropout VTRACK1,2 = 0V VRUN1 Rising VRUN2 Rising VFB1,2 = 0.7V, VSENSE1–,2– = 3.3V, ILIM = 0 l l l l l MIN TYP 2 2 MAX UNITS mmho mA 2 50 75 mA μA Sleep Mode (Both Channels On) Shutdown UVLO VOVL ISENSE+ ISENSE– Undervoltage Lockout Feedback Overvoltage Protection SENSE+ Pin Current SENSE– Pins Current 60 14 3.6 7 4.0 3.8 10 100 25 4.2 4.0 13 ±1 ±1 μA μA V V % μA μA μA % μA V V mV mV Ω Ω Ω Ω ns ns ns ns ns ns ns 700 98 0.7 1.15 1.20 64 99 1.0 1.21 1.25 50 75 2.5 1.5 2.4 1.1 85 1.4 1.27 1.30 DFMAX ITRACK/SS1,2 VRUN1 On VRUN2 On VSENSE(MAX) Gate Driver TG1,2 BG1,2 Maximum Duty Factor Soft-Start Charge Current RUN1 Pin On Threshold RUN2 Pin On Threshold Maximum Current Sense Threshold Pull-Up On-Resistance Pull-Down On-Resistance Pull-Up On-Resistance Pull-Down On-Resistance TG Transition Time: Rise Time Fall Time BG Transition Time: Rise Time Fall Time Top Gate Off to Bottom Gate On Delay Synchronous Switch-On Delay Time Bottom Gate Off to Top Gate On Delay Top Switch-On Delay Time Minimum On-Time Internal VCC Voltage INTVCC Load Regulation Internal VCC Voltage VRUN1,2 Hyst RUN Pin Hysteresis TG1,2 tr TG1,2 tf BG1,2 tr BG1,2 tf TG/BG t1D BG/TG t1D tON(MIN) VINTVCCVIN VLDOVIN VINTVCCEXT (Note 6) CLOAD = 3300pF CLOAD = 3300pF (Note 6) CLOAD = 3300pF CLOAD = 3300pF CLOAD = 3300pF Each Driver CLOAD = 3300pF Each Driver (Note 7) 6V < VIN < 60V, VEXTVCC = 0V ICC = 0mA to 50mA, VEXTVCC = 0V 6V < VEXTVCC < 13V 4.85 4.85 25 25 25 25 30 30 95 5.1 0.7 5.1 5.35 1.1 5.35 INTVCC Linear Regulator V % V 38901f 3 LTC3890-1 ELECTRICAL CHARACTERISTICS SYMBOL VLDOEXT VEXTVCC VLDOHYS f25kΩ f65kΩ f105kΩ fLOW fHIGH fSYNC VPGL IPGOOD VPG PARAMETER INTVCC Load Regulation EXTVCC Switchover Voltage EXTVCC Hysteresis Programmable Frequency Programmable Frequency Programmable Frequency Low Fixed Frequency High Fixed Frequency Synchronizable Frequency PGOOD1 Voltage Low PGOOD1 Leakage Current PGOOD1 Trip Level RFREQ = 25k, PLLIN/MODE = DC Voltage RFREQ = 65k, PLLIN/MODE = DC Voltage RFREQ = 105k, PLLIN/MODE = DC Voltage VFREQ = 0V, PLLIN/MODE = DC Voltage VFREQ = INTVCC, PLLIN/MODE = DC Voltage PLLIN/MODE = External Clock IPGOOD = 2mA VPGOOD = 5V VFB with Respect to Set Regulated Voltage VFB Ramping Negative Hysteresis VFB with Respect to Set Regulated Voltage VFB Ramping Positive Hysteresis tPG Delay for Reporting a Fault –13 –10 2.5 10 2.5 25 l The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TJ = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted. CONDITIONS ICC = 0mA to 50mA, VEXTVCC = 8.5V EXTVCC Ramping Positive 4.5 MIN TYP 0.6 4.7 250 105 375 320 485 75 0.2 440 835 350 535 380 585 850 0.4 ±1 –7 505 MAX 1.1 4.9 UNITS % V mV kHz kHz kHz kHz kHz kHz V μA % % % % μs Oscillator and Phase-Locked Loop PGOOD1 Output 7 13 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Ratings for extended periods may affect device reliability and lifetime. Note 2: The LTC3890E-1 is guaranteed to meet performance specifications from 0°C to 125°C. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3890I-1 is guaranteed over the full –40°C to 125°C operating junction temperature range. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD • 90°C/W) Note 4: The LTC3890-1 is tested in a feedback loop that servos VITH1,2 to a specified voltage and measures the resultant VFB1,2. The specification at 85°C is not tested in production. This specification is assured by design, characterization and correlation to production testing at 125°C. Note 5: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications information. Note 6: Rise and fall times are measured using 10% and 90% levels. Delay times are measured using 50% levels. Note 7: The minimum on-time condition is specified for an inductor peakto-peak ripple current ≥ of IMAX (See Minimum On-Time Considerations in the Applications Information section). 38901f 4 LTC3890-1 TYPICAL PERFORMANCE CHARACTERISTICS Efficiency and Power Loss vs Output Current 100 VIN = 12V 90 VOUT = 3.3V 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.0001 PULSE-SKIPPING LOSS BURST LOSS 10 CCM LOSS 100 10000 BURST EFFICIENCY 1000 POWER LOSS (mW) EFFICIENCY (%) 100 90 80 70 60 50 40 30 20 10 0.1 10 38901 G01 Efficiency vs Output Current 100 VOUT = 8.5V VOUT = 3.3V 98 96 EFFICIENCY (%) 94 92 90 88 86 84 Burst Mode OPERATION 82 80 10 38901 G02 Efficiency vs Input Voltage VOUT2 = 8.5V VOUT1 = 3.3V CCM EFFICIENCY PULSE-SKIPPING EFFICIENCY 0.001 0.01 0.1 1 OUTPUT CURRENT (A) 1 VIN = 12V 0 0.0001 0.001 0.01 0.1 1 OUTPUT CURRENT (A) FIGURE 13 CIRCUIT ILOAD = 2A 0 5 10 15 20 25 30 35 40 45 50 55 60 INPUT VOLTAGE (V) 38901 G03 FIGURE 13 CIRCUIT FIGURE 13 CIRCUIT Load Step Burst Mode Operation VOUT 100mV/DIV ACCOUPLED VOUT 100mV/DIV ACCOUPLED Load Step Pulse-Skipping Mode VOUT 100mV/DIV ACCOUPLED Load Step Forced Continuous Mode IL 2A/DIV 50μs/DIV VIN = 12V VOUT = 3.3V FIGURE 13 CIRCUIT 38901 G04 IL 2A/DIV 50μs/DIV VIN = 12V VOUT = 3.3V FIGURE 13 CIRCUIT 38901 G05 IL 2A/DIV 50μs/DIV VIN = 12V VOUT = 3.3V FIGURE 13 CIRCUIT 38901 G06 Inductor Current at Light Load FORCED CONTINUOUS MODE Soft Start-Up Tracking Start-Up VOUT2 2V/DIV VOUT2 2V/DIV Burst Mode OPERATION 1A/DIV PULSE-SKIPPING MODE 5μs/DIV VIN = 12V VOUT = 3.3V ILOAD = 200μA 38901 G07 VOUT1 2V/DIV VOUT1 2V/DIV 2ms/DIV FIGURE 13 CIRCUIT 38901 G08 2ms/DIV FIGURE 13 CIRCUIT 38901 G09 38901f 5 LTC3890-1 TYPICAL PERFORMANCE CHARACTERISTICS Total Input Supply Current vs Input Voltage 300 250 SUPPLY CURRENT (μA) 200 300μA 150 100 NO LOAD 50 0 5 10 15 20 25 30 35 40 45 50 55 60 65 INPUT VOLTAGE (V) 38901 G10 EXTVCC Switchover and INTVCC Voltages vs Temperature 6.0 5.8 EXTVCC AND INTVCC VOLTAGE (V) 5.6 5.4 5.2 5.0 4.8 4.6 4.4 4.2 4.0 –45 4.8 –20 55 30 5 80 TEMPERATURE (°C) 105 130 INTVCC EXTVCC RISING EXTVCC FALLING INTVCC VOLTAGE (A) 5.1 5.2 INTVCC Line Regulation 5.0 4.9 5 10 15 20 25 30 35 40 45 50 55 60 65 INPUT VOLTAGE (V) 38901 G12 38901 G11 Maximum Current Sense Voltage vs ITH Voltage 80 CURRENT SENSE THESHOLD (mV) 60 40 20 0 –20 FORCED CONTINUOUS MODE –40 0 0.2 0.4 0.6 0.8 VITH (V) 1.0 1.2 1.4 –700 –800 SENSE– CURRENT (A) PULSE SKIPPING MODE Burst Mode OPERATION 0 –100 –200 –300 –400 –500 –600 SENSE– Pin Input Bias Current MAXIMUM CURRENT SENSE VOLTAGE (mV) 90 85 80 75 70 65 60 Maximum Current Sense Threshold vs Duty Cycle 0 5 10 15 20 INPUT VOLTAGE (V) 25 38901 G14 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) 38901 G15 38901 G13 Foldback Current Limit MAXIMUM CURRENT SENSE VOLTAGE (mV) 80 70 QUIESCENT CURRENT (μA) 60 50 40 30 20 10 0 0 100 200 300 400 500 600 FEEDBACK VOLTAGE (MV) 700 800 3890 G16 Quiescent Current vs Temperature 80 75 70 65 60 55 50 45 40 –45 –20 4.95 5 80 55 30 TEMPERATURE (°C) 105 130 INVCC VOLTAGE (V) 5.15 5.20 INTVCC vs Load Current VIN = 12V 5.10 EXTVCC = 0V 5.05 EXTVCC = 8.5V 5.00 0 20 40 60 80 100 120 140 160 180 200 LOAD CURRENT (mA) 38901 G18 38901 G17 38901f 6 LTC3890-1 TYPICAL PERFORMANCE CHARACTERISTICS TRACK/SS Pull-Up Current vs Temperature 1.10 1.40 1.35 TRACK/SS CURRENT (μA) RUN PIN VOLTAGE (V) 1.05 1.30 1.25 1.20 1.15 1.10 1.05 0.90 –45 –20 80 5 30 55 TEMPERATURE (°C) 105 130 1.00 –45 –20 55 30 80 5 TEMPERATURE (°C) 105 130 RUN1 FALLING RUN2 FALLING RUN1 RISING REGULATED FEEDBACK VOLTAGE (mV) Shutdown (RUN) Threshold vs Temperature 800 806 804 802 800 798 796 794 Regulated Feedback Voltage vs Temperature RUN2 RISING 1.00 0.95 792 –45 –20 5 80 55 30 TEMPERATURE (°C) 105 130 38901 G19 38901 G20 38901 G21 SENSE– Pin Total Input Bias Current vs Temperature 50 0 VOUT < INTVCC – 0.5V –50 –100 –150 –200 –250 –300 –350 –400 –450 –500 –550 –600 VOUT > INTVCC – 0.5V –650 –700 –750 –800 –45 –25 –5 15 35 55 75 95 115 TEMPERATURE (°C) 38901 G22 Shutdown Current vs Input Voltage 30 25 SHUTDOWN CURRENT (μA) FREQUENCY (kHz) 20 15 10 5 0 5 10 15 20 25 30 35 40 45 50 55 60 65 INPUT VOLTAGE (V) 38901 G23 Oscillator Frequency vs Temperature 600 550 500 450 400 FREQ = GND 350 300 –45 –20 FREQ = INTVCC SENSE – CURRENT (μA) 55 30 80 5 TEMPERATURE (°C) 105 130 38901 G24 Undervoltage Lockout Threshold vs Temperature 4.4 4.3 4.2 INTVCC VOLTAGE (V) 4.1 4.0 3.9 3.8 3.7 3.6 3.5 3.4 –45 –20 55 30 5 80 TEMPERATURE (°C) 105 130 344 FALLING RISING OSCILLATOR FREQUENCY (kHz) 354 356 Oscillator Frequency vs Input Voltage FREQ = GND SHUTDOWN CURRENT (μA) 22 20 18 16 14 12 10 Shutdown Current vs Temperature VIN = 12V 352 350 348 346 5 10 15 20 25 30 35 40 45 50 55 60 65 INPUT VOLTAGE (V) 38901 G26 8 –45 –20 5 30 55 80 TEMPERATURE (°C) 105 130 38901 G25 38901 G27 38901f 7 LTC3890-1 PIN FUNCTIONS ITH1, ITH2 (Pin 1, Pin 13): Error Amplifier Outputs and Switching Regulator Compensation Points. Each associated channel’s current comparator trip point increases with this control voltage. VFB1, VFB2 (Pin 2, Pin 12): Receives the remotely sensed feedback voltage for each controller from an external resistive divider across the output. SENSE1+, (Pin 3, Pin 11): The (+) input to the differential current comparators are normally connected to DCR sensing networks or current sensing resistors. The ITH pin voltage and controlled offsets between the SENSE– and SENSE+ pins in conjunction with RSENSE set the current trip threshold. SENSE2+ at light loads. Pulling this pin to ground selects Burst Mode operation. An internal 100k resistor to ground also invokes Burst Mode Operation when the pin is floated. Tying this pin to INTVCC forces continuous inductor current operation. Tying this pin to a voltage greater than 1.2V and less than INTVCC – 1.3V selects pulse skipping operation. SGND (Pin 7): Small-signal ground common to both controllers, must be routed separately from high current grounds to the common (–) terminals of the CIN capacitors. RUN1, RUN2 (Pin 8, Pin 9): Digital Run Control Inputs for Each Controller. Forcing either of these pins below 1.2V shuts down that controller. Forcing both of these pins below 0.7V shuts down the entire LTC3890-1, reducing quiescent current to approximately 14μA. INTVCC (Pin 19): Output of the Internal Linear Low Dropout Regulator. The driver and control circuits are powered from this voltage source. Must be decoupled to power ground with a minimum of 4.7μF ceramic or other low ESR capacitor. Do not use the INTVCC pin for any other purpose. EXTVCC (Pin 20): External Power Input to an Internal LDO Connected to INTVCC. This LDO supplies INTVCC power, bypassing the internal LDO powered from VIN whenever EXTVCC is higher than 4.7V. See EXTVCC Connection in the Applications Information section. Do not exceed 14V on this pin. PGND (Pin 21): Driver Power Ground. Connects to the sources of bottom (synchronous) N-channel MOSFETs and the (–) terminal(s) of CIN. VIN (Pin 22): Main Supply Pin. A bypass capacitor should be tied between this pin and the signal ground pin. SENSE1–, SENSE2– (Pin 4, Pin 10): The (–) Input to the Differential Current Comparators. When greater than INTVCC – 0.5V, the SENSE– pin supplies current to the current comparator. FREQ (Pin 5): The Frequency Control Pin for the Internal VCO. Connecting the pin to GND forces the VCO to a fixed low frequency of 350kHz. Connecting the pin to INTVCC forces the VCO to a fixed high frequency of 535kHz. Other frequencies between 50kHz and 900kHz can be programmed using a resistor between FREQ and GND. An internal 20μA pull-up current develops the voltage to be used by the VCO to control the frequency. PLLIN/MODE (Pin 6): External Synchronization Input to Phase Detector and Forced Continuous Mode Input. When an external clock is applied to this pin, the phase-locked loop will force the rising TG1 signal to be synchronized with the rising edge of the external clock. When not synchronizing to an external clock, this input, which acts on both controllers, determines how the LTC3890-1 operates 38901f 8 LTC3890-1 PIN FUNCTIONS BG1, BG2 (Pin 23, Pin 18): High Current Gate Drives for Bottom (Synchronous) N-Channel MOSFETs. Voltage swing at these pins is from ground to INTVCC. BOOST1, BOOST2 (Pin 24, Pin 17): Bootstrapped Supplies to the Topside Floating Drivers. Capacitors are connected between the BOOST and SW pins and Schottky diodes are tied between the BOOST and INTVCC pins. Voltage swing at the BOOST pins is from INTVCC to (VIN + INTVCC). SW1, SW2 (Pin 25, Pin 16): Switch Node Connections to Inductors. TG1, TG2 (Pin 26, Pin 15): High Current Gate Drives for Top N-Channel MOSFETs. These are the outputs of floating drivers with a voltage swing equal to INTVCC – 0.5V superimposed on the switch node voltage SW. PGOOD1 (Pin 27): Open-Drain Logic Output. PGOOD1 is pulled to ground when the voltage on the VFB1 pin is not within ±10% of its set point. TRACK/SS1, TRACK/SS2 (Pin 28, Pin 14): External Tracking and Soft-Start Input. The LTC3890-1 regulates the VFB1,2 voltage to the smaller of 0.8V or the voltage on the TRACK/SS1,2 pin. An internal 1μA pull-up current source is connected to this pin. A capacitor to ground at this pin sets the ramp time to final regulated output voltage. Alternatively, a resistor divider on another voltage supply connected to this pin allows the LTC3890-1 output to track the other supply during start-up. 38901f 9 LTC3890-1 FUNCTIONAL DIAGRAM INTVCC DUPLICATE FOR SECOND CONTROLLER CHANNEL BOOST DB CB D SW INTVCC BG PGND VCO CLK2 CLK1 0.425V SLEEP L RSENSE – + CIN VIN FREQ 20μA 3mV PLLIN/MODE 100k SLOPE COMP SYNC DET 2.7V 0.65V VIN EXTVCC OV 5.1V LDO EN + 5.1V LDO EN 7μA (RUN1) 0.5μA (RUN2) SHDN RST 2(VFB) 11V FOLDBACK 4.7V SHDN SGND INTVCC RUN 10 + – EA + – – –+ +– + CLP + – + + – – – PGOOD1 0.88V VFB1 0.72V S R Q Q DROP OUT DET TOP BOT TOP ON TG SHDN SWITCH LOGIC BOT COUT VOUT PFD ICMP IR SENSE+ SENSE– VFB 0.80V TRACK/SS RA RB 0.88V ITH CC CC2 1μA TRACK/SS RC CSS 38901 FD 38901f LTC3890-1 OPERATION (Refer to the Functional Diagram) Main Control Loop The LTC3890-1 uses a constant frequency, current mode step-down architecture with the two controller channels operating 180 degrees out of phase. During normal operation, each external top MOSFET is turned on when the clock for that channel sets the RS latch, and is turned off when the main current comparator, ICMP, resets the RS latch. The peak inductor current at which ICMP trips and resets the latch is controlled by the voltage on the ITH pin, which is the output of the error amplifier, EA. The error amplifier compares the output voltage feedback signal at the VFB pin, (which is generated with an external resistor divider connected across the output voltage, VOUT , to ground) to the internal 0.800V reference voltage. When the load current increases, it causes a slight decrease in VFB relative to the reference, which causes the EA to increase the ITH voltage until the average inductor current matches the new load current. After the top MOSFET is turned off each cycle, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by the current comparator IR, or the beginning of the next clock cycle. INTVCC/EXTVCC Power Power for the top and bottom MOSFET drivers and most other internal circuitry is derived from the INTVCC pin. When the EXTVCC pin is left open or tied to a voltage less than 4.7V, the VIN LDO (low dropout linear regulator) supplies 5.1V from VIN to INTVCC. If EXTVCC is taken above 4.7V, the VIN LDO is turned off and an EXTVCC LDO is turned on. Once enabled, the EXTVCC LDO supplies 5.1V from EXTVCC to INTVCC. Using the EXTVCC pin allows the INTVCC power to be derived from a high efficiency external source such as one of the LTC3890-1 switching regulator outputs. Each top MOSFET driver is biased from the floating bootstrap capacitor CB, which normally recharges during each cycle through an external diode when the top MOSFET turns off. If the input voltage, VIN, decreases to a voltage close to VOUT , the loop may enter dropout and attempt to turn on the top MOSFET continuously. The dropout detector detects this and forces the top MOSFET off for about one-twelfth of the clock period every tenth cycle to allow CB to recharge. Shutdown and Start-Up (RUN1, RUN2 and TRACK/ SS1, TRACK/SS2 Pins) The two channels of the LTC3890-1 can be independently shut down using the RUN1 and RUN2 pins. Pulling either of these pins below 1.2V shuts down the main control loop for that controller. Pulling both pins below 0.7V disables both controllers and most internal circuits, including the INTVCC LDOs. In this state, the LTC3890-1 draws only 14μA of quiescent current. Releasing either RUN pin allows a small internal current to pull up the pin to enable that controller. The RUN1 pin has a 7μA pull-up current while the RUN2 pin has a smaller 0.5μA. The 7μA current on RUN1 is designed to be large enough so that the RUN1 pin can be safely floated (to always enable the controller) without worry of condensation or other small board leakage pulling the pin down. This is ideal for always-on applications where one or both controllers are enabled continuously and never shut down. The RUN pin may be externally pulled up or driven directly by logic. When driving the RUN pin with a low impedance source, do not exceed the absolute maximum rating of 8V. The RUN pin has an internal 11V voltage clamp that allows the RUN pin to be connected through a resistor to a higher voltage (for example, VIN), so long as the maximum current into the RUN pin does not exceed 100μA. The start-up of each controller’s output voltage VOUT is controlled by the voltage on the TRACK/SS pin for that channel. When the voltage on the TRACK/SS pin is less than the 0.8V internal reference, the LTC3890-1 regulates the VFB voltage to the TRACK/SS pin voltage instead of the 0.8V reference. This allows the TRACK/SS pin to be used to program a soft-start by connecting an external capacitor from the TRACK/SS pin to SGND. An internal 1μA pull-up current charges this capacitor creating a voltage ramp on the TRACK/SS pin. As the TRACK/SS voltage rises linearly from 0V to 0.8V (and beyond up to 5V), the output voltage VOUT rises smoothly from zero to its final value. Alternatively the TRACK/SS pin can be used to cause the start-up of VOUT to track that of another supply. Typically, this requires connecting to the TRACK/SS pin an external resistor divider from the other supply to ground (see Applications Information section). 38901f 11 LTC3890-1 OPERATION (Refer to the Functional Diagram) Light Load Current Operation (Burst Mode Operation, Pulse Skipping, or Forced Continuous Mode) (PLLIN/MODE Pin) The LTC3890-1 can be enabled to enter high efficiency Burst Mode operation, constant frequency pulse skipping mode, or forced continuous conduction mode at low load currents. To select Burst Mode operation, tie the PLLIN/ MODE pin to a DC voltage below 0.8V (e.g., SGND). To select forced continuous operation, tie the PLLIN/MODE pin to INTVCC. To select pulse-skipping mode, tie the PLLIN/MODE pin to a DC voltage greater than 1.2V and less than INTVCC – 1.3V. When a controller is enabled for Burst Mode operation, the minimum peak current in the inductor is set to approximately 25% of the maximum sense voltage even though the voltage on the ITH pin indicates a lower value. If the average inductor current is higher than the load current, the error amplifier, EA, will decrease the voltage on the ITH pin. When the ITH voltage drops below 0.425V, the internal sleep signal goes high (enabling sleep mode) and both external MOSFETs are turned off. The ITH pin is then disconnected from the output of the EA and parked at 0.450V. In sleep mode, much of the internal circuitry is turned off, reducing the quiescent current that the LTC3890-1 draws. If one channel is shut down and the other channel is in sleep mode, the LTC3890-1 draws only 50μA of quiescent current. If both channels are in sleep mode, the LTC3890-1 draws only 60μA of quiescent current. In sleep mode, the load current is supplied by the output capacitor. As the output voltage decreases, the EA’s output begins to rise. When the output voltage drops enough, the ITH pin is reconnected to the output of the EA, the sleep signal goes low, and the controller resumes normal operation by turning on the top external MOSFET on the next cycle of the internal oscillator. When a controller is enabled for Burst Mode operation, the inductor current is not allowed to reverse. The reverse current comparator, IR, turns off the bottom external MOSFET just before the inductor current reaches zero, preventing it from reversing and going negative. Thus, the controller operates in discontinuous operation. In forced continuous operation or clocked by an external clock source to use the phase-locked loop (see Frequency Selection and Phase-Locked Loop section), the inductor current is allowed to reverse at light loads or under large transient conditions. The peak inductor current is determined by the voltage on the ITH pin, just as in normal operation. In this mode, the efficiency at light loads is lower than in Burst Mode operation. However, continuous operation has the advantage of lower output voltage ripple and less interference to audio circuitry. In forced continuous mode, the output ripple is independent of load current. When the PLLIN/MODE pin is connected for pulse skipping mode, the LTC3890-1 operates in PWM pulse skipping mode at light loads. In this mode, constant frequency operation is maintained down to approximately 1% of designed maximum output current. At very light loads, the current comparator, ICMP, may remain tripped for several cycles and force the external top MOSFET to stay off for the same number of cycles (i.e., skipping pulses). The inductor current is not allowed to reverse (discontinuous operation). This mode, like forced continuous operation, exhibits low output ripple as well as low audio noise and reduced RF interference as compared to Burst Mode operation. It provides higher low current efficiency than forced continuous mode, but not nearly as high as Burst Mode operation. Frequency Selection and Phase-Locked Loop (FREQ and PLLIN/MODE Pins) The selection of switching frequency is a trade-off between efficiency and component size. Low frequency operation increases efficiency by reducing MOSFET switching losses, but requires larger inductance and/or capacitance to maintain low output ripple voltage. The switching frequency of the LTC3890-1’s controllers can be selected using the FREQ pin. 38901f 12 LTC3890-1 OPERATION (Refer to the Functional Diagram) If the PLLIN/MODE pin is not being driven by an external clock source, the FREQ pin can be tied to SGND, tied to INTVCC or programmed through an external resistor. Tying FREQ to SGND selects 350kHz while tying FREQ to INTVCC selects 535kHz. Placing a resistor between FREQ and SGND allows the frequency to be programmed between 50kHz and 900kHz, as shown in Figure 10. A phase-locked loop (PLL) is available on the LTC3890-1 to synchronize the internal oscillator to an external clock source that is connected to the PLLIN/MODE pin. The LTC3890-1’s phase detector adjusts the voltage (through an internal lowpass filter) of the VCO input to align the turn-on of controller 1’s external top MOSFET to the rising edge of the synchronizing signal. Thus, the turn-on of controller 2’s external top MOSFET is 180 degrees out of phase to the rising edge of the external clock source. The VCO input voltage is prebiased to the operating frequency set by the FREQ pin before the external clock is applied. A resistor connected between the FREQ pin and SGND can prebias VCO’s input voltage to the desired frequency. If prebiased near the external clock frequency, the PLL loop only needs to make slight changes to the VCO input in order to synchronize the rising edge of the external clock’s to the rising edge of TG1. The ability to prebias the loop filter allows the PLL to lock-in rapidly without deviating far from the desired frequency. The typical capture range of the phase-locked loop is from approximately 55kHz to 1MHz, with a guarantee to be between 75kHz and 850kHz. In other words, the LTC3890-1’s PLL is guaranteed to lock to an external clock source whose frequency is between 75kHz and 850kHz. The typical input clock thresholds on the PLLIN/MODE pin are 1.6V (rising) and 1.2V (falling). Output Overvoltage Protection An overvoltage comparator guards against transient overshoots as well as other more serious conditions that may overvoltage the output. When the VFB pin rises by more than 10% above its regulation point of 0.800V, the top MOSFET is turned off and the bottom MOSFET is turned on until the overvoltage condition is cleared. Power Good (PGOOD1 Pin) The PGOOD1 pin is connected to an open drain of an internal N-channel MOSFET. The MOSFET turns on and pulls the PGOOD1 pin low when the corresponding VFB1 pin voltage is not within ±10% of the 0.8V reference voltage. The PGOOD1 pin is also pulled low when the corresponding RUN1 pin is low (shut down). When the VFB1 pin voltage is within the ±10% requirement, the MOSFET is turned off and the pin is allowed to be pulled up by an external resistor to a source no greater than 6V. Foldback Current When the output voltage falls to less than 70% of its nominal level, foldback current limiting is activated, progressively lowering the peak current limit in proportion to the severity of the overcurrent or short-circuit condition. Foldback current limiting is disabled during the soft-start interval (as long as the VFB voltage is keeping up with the TRACK/SS voltage). Theory and Benefits of 2-Phase Operation Why the need for 2-phase operation? Up until the 2-phase family, constant-frequency dual switching regulators operated both channels in phase (i.e., single phase operation). This means that both switches turned on at the same time, causing current pulses of up to twice the amplitude of those for one regulator to be drawn from the input capacitor and battery. These large amplitude current pulses increased the total RMS current flowing from the input capacitor, requiring the use of more expensive input capacitors and increasing both EMI and losses in the input capacitor and battery. With 2-phase operation, the two channels of the dual switching regulator are operated 180 degrees out of phase. This effectively interleaves the current pulses drawn by the switches, greatly reducing the overlap time where they add 38901f 13 LTC3890-1 OPERATION (Refer to the Functional Diagram) together. The result is a significant reduction in total RMS input current, which in turn allows less expensive input capacitors to be used, reduces shielding requirements for EMI and improves real world operating efficiency. Figure 1 compares the input waveforms for a representative single-phase dual switching regulator to the LTC3890-1 2-phase dual switching regulator. An actual measurement of the RMS input current under these conditions shows that 2-phase operation dropped the input current from 2.53ARMS to 1.55ARMS. While this is an impressive reduction in itself, remember that the power losses are proportional to IRMS2, meaning that the actual power wasted is reduced by a factor of 2.66. The reduced input ripple voltage also means less power is lost in the input power path, which could include batteries, switches, trace/connector resistances and protection circuitry. Improvements in both conducted and radiated EMI also directly accrue as a result of the reduced RMS input current and voltage. Of course, the improvement afforded by 2-phase operation is a function of the dual switching regulator’s relative duty cycles which, in turn, are dependent upon the input voltage VIN (Duty Cycle = VOUT/VIN). Figure 2 shows how the RMS input current varies for single phase and 2-phase operation for 3.3V and 5V regulators over a wide input voltage range. It can readily be seen that the advantages of 2-phase operation are not just limited to a narrow operating range, for most applications is that 2-phase operation will reduce the input capacitor requirement to that for just one channel operating at maximum current and 50% duty cycle. The schematic on the first page is a basic LTC3890-1 application circuit. External component selection is driven by the load requirement, and begins with the selection of RSENSE and the inductor value. Next, the power MOSFETs are selected. Finally, CIN and COUT are selected 3.0 2.5 INPUT RMS CURRENT (A) 2.0 1.5 1.0 0.5 0 2-PHASE DUAL CONTROLLER SINGLE PHASE DUAL CONTROLLER VO1 = 5V/3A VO2 = 3.3V/3A 0 10 20 30 INPUT VOLTAGE (V) 40 38901 F02 Figure 2. RMS Input Current Comparison 5V SWITCH 20V/DIV 3.3V SWITCH 20V/DIV INPUT CURRENT 5A/DIV INPUT VOLTAGE 500mV/DIV IIN(MEAS) = 2.53ARMS IIN(MEAS) = 1.55ARMS 38901 F01 Figure 1. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for Dual Switching Regulators Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input Ripple with the 2-Phase Regulator Allows Less Expensive Input Capacitors, Reduces Shielding Requirements for EMI and Improves Efficiency 38901f 14 LTC3890-1 APPLICATIONS INFORMATION The Typical Application on the first page is a basic LTC3890-1 application circuit. LTC3890-1 can be configured to use either DCR (inductor resistance) sensing or low value resistor sensing. The choice between the two current sensing schemes is largely a design trade-off between cost, power consumption, and accuracy. DCR sensing is becoming popular because it saves expensive current sensing resistors and is more power efficient, especially in high current applications. However, current sensing resistors provide the most accurate current limits for the controller. Other external component selection is driven by the load requirement, and begins with the selection of RSENSE (if RSENSE is used) and inductor value. Next, the power MOSFETs and Schottky diodes are selected. Finally, input and output capacitors are selected. SENSE+ and SENSE– Pins and pins are the inputs to the curThe rent comparators. The common mode voltage range on these pins is 0V to 28V (abs max), enabling the LTC3857 to regulate output voltages up to a nominal 24V (allowing margin for tolerances and transients). The SENSE+ pin is high impedance over the full common mode range, drawing at most ±1μA. This high impedance allows the current comparators to be used in inductor DCR sensing. The impedance of the SENSE– pin changes depending on the common mode voltage. When SENSE– is less than INTVCC – 0.5V, a small current of less than 1μA flows out of the pin. When SENSE– is above INTVCC + 0.5V, a higher current (~700μA) flows into the pin. Between INTVCC – 0.5V and INTVCC + 0.5V, the current transitions from the smaller current to the higher current. Filter components mutual to the sense lines should be placed close to the LTC3890-1, and the sense lines should run close together to a Kelvin connection underneath the current sense element (shown in Figure 3). Sensing current elsewhere can effectively add parasitic inductance and capacitance to the current sense element, degrading the information at the sense terminals and making the SENSE+ SENSE– programmed current limit unpredictable. If inductor DCR sensing is used (Figure 4b), sense resistor R1 should be placed close to the switching node, to prevent noise from coupling into sensitive small-signal nodes. TO SENSE FILTER, NEXT TO THE CONTROLLER COUT 38901 F03 INDUCTOR OR RSENSE Figure 3. Sense Lines Placement with Inductor or Sense Resistor VIN INTVCC BOOST TG SW LTC3890-1 BG SENSE+ SENSE– SGND 38901 F04a VIN RSENSE VOUT PLACE CAPACITOR NEAR SENSE PINS (4a) Using a Resistor to Sense Current VIN INTVCC BOOST TG SW LTC3890-1 BG SENSE+ C1* SENSE– SGND *PLACE C1 NEAR SENSE PINS (R1||R2) • C1 = L DCR RSENSE(EQ) = DCR R2 R1 + R2 38901 F04b VIN INDUCTOR L DCR VOUT R1 R2 (4b) Using the Inductor DCR to Sense Current Figure 4. Current Sensing Methods 38901f 15 LTC3890-1 APPLICATIONS INFORMATION Low Value Resistor Current Sensing A typical sensing circuit using a discrete resistor is shown in Figure 4a. RSENSE is chosen based on the required output current. The current comparator has a maximum threshold VSENSE(MAX). The current comparator threshold voltage sets the peak of the inductor current, yielding a maximum average output current, IMAX, equal to the peak value less half the peak-to-peak ripple current, ΔIL. To calculate the sense resistor value, use the equation: RSENSE = VSENSE(MAX) IMAX + ΔIL 2 If the external (R1||R2) • C1 time constant is chosen to be exactly equal to the L/DCR time constant, the voltage drop across the external capacitor is equal to the drop across the inductor DCR multiplied by R2/(R1 + R2). R2 scales the voltage across the sense terminals for applications where the DCR is greater than the target sense resistor value. To properly dimension the external filter components, the DCR of the inductor must be known. It can be measured using a good RLC meter, but the DCR tolerance is not always the same and varies with temperature; consult the manufacturers’ data sheets for detailed information. Using the inductor ripple current value from the Inductor Value Calculation section, the target sense resistor value is: RSENSE(EQUIV) = VSENSE(MAX) IMAX + ΔIL 2 To ensure that the application will deliver full load current over the full operating temperature range, choose the minimum value for the Maximum Current Sense Threshold (VSENSE(MAX)). When using the controller in very low dropout conditions, the maximum output current level will be reduced due to the internal compensation required to meet stability criterion for buck regulators operating at greater than 50% duty factor. A curve is provided in the Typical Performance Characteristics section to estimate this reduction in peak inductor current depending upon the operating duty factor. Inductor DCR Sensing For applications requiring the highest possible efficiency at high load currents, the LTC3890-1 is capable of sensing the voltage drop across the inductor DCR, as shown in Figure 4b. The DCR of the inductor represents the small amount of DC resistance of the copper wire, which can be less than 1mΩ for today’s low value, high current inductors. In a high current application requiring such an inductor, power loss through a sense resistor would cost several points of efficiency compared to inductor DCR sensing. To ensure that the application will deliver full load current over the full operating temperature range, choose the minimum value for the Maximum Current Sense Threshold (VSENSE(MAX)). Next, determine the DCR of the inductor. When provided, use the manufacturer’s maximum value, usually given at 20°C. Increase this value to account for the temperature coefficient of copper resistance, which is approximately 0.4%/°C. A conservative value for TL(MAX) is 100°C. To scale the maximum inductor DCR to the desired sense resistor value (RD), use the divider ratio: RD = RSENSE(EQUIV ) DCRMAX at TL(MAX ) C1 is usually selected to be in the range of 0.1μF to 0.47μF. This forces R1|| R2 to around 2k, reducing error that might have been caused by the SENSE+ pin’s ±1μA current. 38901f 16 LTC3890-1 APPLICATIONS INFORMATION The equivalent resistance R1|| R2 is scaled to the room temperature inductance and maximum DCR: R1|| R2 = L DCR  at   20°C • C1 Accepting larger values of ΔIL allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is ΔIL =0.3(IMAX). The maximum ΔIL occurs at the maximum input voltage. The inductor value also has secondary effects. The transition to Burst Mode operation begins when the average inductor current required results in a peak current below 25% of the current limit determined by RSENSE. Lower inductor values (higher ΔIL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to decrease. Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite or molypermalloy cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance value selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss and are preferred for high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates hard, which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Power MOSFET and Schottky Diode (Optional) Selection Two external power MOSFETs must be selected for each controller in the LTC3890-1: one N-channel MOSFET for the top (main) switch, and one N-channel MOSFET for the bottom (synchronous) switch. ( ) The sense resistor values are: R1 = R1 • RD R1|| R2 ;  R2 = RD 1 – RD The maximum power loss in R1 is related to duty cycle, and will occur in continuous mode at the maximum input voltage: PLOSS  R1 = ( VIN(MAX) – VOUT ) • VOUT R1 Ensure that R1 has a power rating higher than this value. If high efficiency is necessary at light loads, consider this power loss when deciding whether to use DCR sensing or sense resistors. Light load power loss can be modestly higher with a DCR network than with a sense resistor, due to the extra switching losses incurred through R1. However, DCR sensing eliminates a sense resistor, reduces conduction losses and provides higher efficiency at heavy loads. Peak efficiency is about the same with either method. Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of smaller inductor and capacitor values. So why would anyone ever choose to operate at lower frequencies with larger components? The answer is efficiency. A higher frequency generally results in lower efficiency because of MOSFET switching and gate charge losses. In addition to this basic trade-off, the effect of inductor value on ripple current and low current operation must also be considered. The inductor value has a direct effect on ripple current. The inductor ripple current, ΔIL, decreases with higher inductance or higher frequency and increases with higher VIN: IL = V 1 VOUT 1– OUT VIN ( f) (L) 38901f 17 LTC3890-1 APPLICATIONS INFORMATION The peak-to-peak drive levels are set by the INTVCC voltage. This voltage is typically 5.1V during start-up (see EXTVCC Pin Connection). Consequently, logic-level threshold MOSFETs must be used in most applications. Pay close attention to the BVDSS specification for the MOSFETs as well. Selection criteria for the power MOSFETs include the on-resistance, RDS(ON), Miller capacitance, CMILLER, input voltage and maximum output current. Miller capacitance, CMILLER, can be approximated from the gate charge curve usually provided on the MOSFET manufacturers’ data sheet. CMILLER is equal to the increase in gate charge along the horizontal axis while the curve is approximately flat divided by the specified change in VDS. This result is then multiplied by the ratio of the application applied VDS to the Gate charge curve specified VDS. When the IC is operating in continuous mode the duty cycles for the top and bottom MOSFETs are given by: Main Switch Duty  Cycle = VOUT VIN VIN − VOUT VIN where δ is the temperature dependency of RDS(ON) and RDR (approximately 2Ω) is the effective driver resistance at the MOSFET’s Miller threshold voltage. V THMIN is the typical MOSFET minimum threshold voltage. Both MOSFETs have I2R losses while the topside N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For VIN < 20V the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CMILLER actually provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage when the top switch duty factor is low or during a short-circuit when the synchronous switch is on close to 100% of the period. The term (1+ δ) is generally given for a MOSFET in the form of a normalized RDS(ON) vs Temperature curve, but δ = 0.005/°C can be used as an approximation for low voltage MOSFETs. The optional Schottky diodes D3 and D4 shown in Figure 11 conduct during the dead-time between the conduction of the two power MOSFETs. This prevents the body diode of the bottom MOSFET from turning on, storing charge during the dead-time and requiring a reverse recovery period that could cost as much as 3% in efficiency at high VIN. A 1A to 3A Schottky is generally a good compromise for both regions of operation due to the relatively small average current. Larger diodes result in additional transition losses due to their larger junction capacitance. CIN and COUT Selection The selection of CIN is simplified by the 2-phase architecture and its impact on the worst-case RMS current drawn through the input network (battery/fuse/capacitor). It can be shown that the worst-case capacitor RMS current occurs when only one controller is operating. The controller with the highest (VOUT)(IOUT) product needs to be used in the formula shown in Equation 1 to determine the maximum Synchronous  Switch Duty  Cycle = The MOSFET power dissipations at maximum output current are given by: PMAIN = 2 VOUT (IMAX ) (1+ VIN )RDS(ON) + ( VIN) 2 IMAX (RDR ) (CMILLER ) • 2 1 1 + ( f) VINTVCC – VTHMIN VTHMIN PSYNC = 2 VIN – VOUT (IMAX ) (1+ VIN )RDS(ON) 38901f 18 LTC3890-1 APPLICATIONS INFORMATION RMS capacitor current requirement. Increasing the output current drawn from the other controller will actually decrease the input RMS ripple current from its maximum value. The out-of-phase technique typically reduces the input capacitor’s RMS ripple current by a factor of 30% to 70% when compared to a single phase power supply solution. In continuous mode, the source current of the top MOSFET is a square wave of duty cycle (VOUT)/(VIN). To prevent large voltage transients, a low ESR capacitor sized for the maximum RMS current of one channel must be used. The maximum RMS capacitor current is given by: CIN Required IRMS ≈ IMAX ⎡ ⎤1/ 2 ⎣( VOUT ) ( VIN – VOUT )⎦ (1) VIN 1cm of each other and share a common CIN(s). Separating the drains and CIN may produce undesirable voltage and current resonances at VIN. A small (0.1μF to 1μF) bypass capacitor between the chip VIN pin and ground, placed close to the LTC3890-1, is also suggested. A 10Ω resistor placed between CIN (C1) and the VIN pin provides further isolation between the two channels. The selection of COUT is driven by the effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (ΔVOUT) is approximated by: VOUT IL ESR + 1 8 • f • COUT This formula has a maximum at VIN = 2VOUT , where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturers’ ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may be paralleled to meet size or height requirements in the design. Due to the high operating frequency of the LTC3890-1, ceramic capacitors can also be used for CIN. Always consult the manufacturer if there is any question. The benefit of the LTC3890-1 2-phase operation can be calculated by using Equation 1 for the higher power controller and then calculating the loss that would have resulted if both controller channels switched on at the same time. The total RMS power lost is lower when both controllers are operating due to the reduced overlap of current pulses required through the input capacitor’s ESR. This is why the input capacitor’s requirement calculated above for the worst-case controller is adequate for the dual controller design. Also, the input protection fuse resistance, battery resistance, and PC board trace resistance losses are also reduced due to the reduced peak currents in a 2-phase system. The overall benefit of a multiphase design will only be fully realized when the source impedance of the power supply/battery is included in the efficiency testing. The drains of the top MOSFETs should be placed within where f is the operating frequency, COUT is the output capacitance and ΔIL is the ripple current in the inductor. The output ripple is highest at maximum input voltage since ΔIL increases with input voltage. Setting Output Voltage The LTC3890-1 output voltages are each set by an external feedback resistor divider carefully placed across the output, as shown in Figure 5. The regulated output voltage is determined by: VOUT = 0.8V 1+ RB RA To improve the frequency response, a feedforward capacitor, CFF , may be used. Great care should be taken to route the VFB line away from noise sources, such as the inductor or the SW line. VOUT 1/2 LTC3890-1 VFB RA 38901 F05 RB CFF Figure 5. Setting Output Voltage 38901f 19 LTC3890-1 APPLICATIONS INFORMATION Tracking and Soft-Start (TRACK/SS Pins) The start-up of each VOUT is controlled by the voltage on the respective TRACK/SS pin. When the voltage on the TRACK/SS pin is less than the internal 0.8V reference, the LTC3890-1 regulates the VFB pin voltage to the voltage on the TRACK/SS pin instead of 0.8V. The TRACK/SS pin can be used to program an external soft-start function or to allow VOUT to track another supply during start-up. Soft-start is enabled by simply connecting a capacitor from the TRACK/SS pin to ground, as shown in Figure 6. An internal 1μA current source charges the capacitor, providing a linear ramping voltage at the TRACK/SS pin. The LTC3890-1 will regulate the VFB pin (and hence VOUT) according to the voltage on the TRACK/SS pin, allowing VOUT to rise smoothly from 0V to its final regulated value. The total soft-start time will be approximately: tSS = CSS • 0.8V 1μA 38901 F07b VX(MASTER) OUTPUT VOLTAGE VOUT(SLAVE) 38901 F07a TIME (7a) Coincident Tracking VX(MASTER) OUTPUT VOLTAGE VOUT(SLAVE) TIME 1/2 LTC3890-1 TRACK/SS CSS SGND 38901 F06 (7b) Ratiometric Tracking Figure 7. Two Different Modes of Output Voltage Tracking Vx VOUT Figure 6. Using the TRACK/SS Pin to Program Soft-Start RB RA RTRACKB 1/2 LTC3890-1 VFB Alternatively, the TRACK/SS pin can be used to track two (or more) supplies during start-up, as shown qualitatively in Figures 7a and 7b. To do this, a resistor divider should be connected from the master supply (V X) to the TRACK/ SS pin of the slave supply (VOUT), as shown in Figure 8. During start-up VOUT will track V X according to the ratio set by the resistor divider: + RTRACKB R VX RA = • TRACKA VOUT RTRACKA RA + RB For coincident tracking (VOUT = V X during start-up): RA = RTRACKA RB = RTRACKB TRACK/SS RTRACKA 38901 F08 Figure 8. Using the TRACK/SS Pin for Tracking INTVCC Regulators The LTC3890-1 features two separate internal P-channel low dropout linear regulators (LDO) that supply power at the INTVCC pin from either the VIN supply pin or the EXTVCC pin depending on the connection of the EXTVCC pin. INTVCC powers the gate drivers and much of the LTC3890-1’s internal circuitry. The VIN LDO and the EXTVCC 38901f 20 LTC3890-1 APPLICATIONS INFORMATION LDO regulate INTVCC to 5.1V. Each of these can supply a peak current of 50mA and must be bypassed to ground with a minimum of 4.7μF ceramic capacitor. No matter what type of bulk capacitor is used, an additional 1μF ceramic capacitor placed directly adjacent to the INTVCC and PGND IC pins is highly recommended. Good bypassing is needed to supply the high transient currents required by the MOSFET gate drivers and to prevent interaction between the channels. High input voltage applications in which large MOSFETs are being driven at high frequencies may cause the maximum junction temperature rating for the LTC3890-1 to be exceeded. The INTVCC current, which is dominated by the gate charge current, may be supplied by either the VIN LDO or the EXTVCC LDO. When the voltage on the EXTVCC pin is less than 4.7V, the VIN LDO is enabled. Power dissipation for the IC in this case is highest and is equal to VIN • IINTVCC. The gate charge current is dependent on operating frequency as discussed in the Efficiency Considerations section. The junction temperature can be estimated by using the equations given in Note 3 of the Electrical Characteristics. For example, the LTC3890-1 INTVCC current is limited to less than 15mA from a 40V supply when not using the EXTVCC supply at 70°C ambient temperature: TJ = 70°C + (15mA)(40V)(90°C/W) = 125°C To prevent the maximum junction temperature from being exceeded, the input supply current must be checked while operating in forced continuous mode (PLLIN/MODE = INTVCC) at maximum VIN. When the voltage applied to EXTVCC rises above 4.7V, the VIN LDO is turned off and the EXTVCC LDO is enabled. The EXTVCC LDO remains on as long as the voltage applied to EXTVCC remains above 4.5V. The EXTVCC LDO attempts to regulate the INTVCC voltage to 5.1V, so while EXTVCC is less than 5.1V, the LDO is in dropout and the INTVCC voltage is approximately equal to EXTVCC. When EXTVCC is greater than 5.1V, up to an absolute maximum of 14V, INTVCC is regulated to 5.1V. Using the EXTVCC LDO allows the MOSFET driver and control power to be derived from one of the LTC3890-1’s switching regulator outputs (4.7V ≤ VOUT ≤ 14V) during normal operation and from the VIN LDO when the output is out of regulation (e.g., start-up, short-circuit). If more current is required through the EXTVCC LDO than is specified, an external Schottky diode can be added between the EXTVCC and INTVCC pins. In this case, do not apply more than 6V to the EXTVCC pin and make sure that EXTVCC ≤ VIN. Significant efficiency and thermal gains can be realized by powering INTVCC from the output, since the VIN current resulting from the driver and control currents will be scaled by a factor of (Duty Cycle)/(Switcher Efficiency). For 5V to 14V regulator outputs, this means connecting the EXTVCC pin directly to VOUT . Tying the EXTVCC pin to an 8.5V supply reduces the junction temperature in the previous example from 125°C to: TJ = 70°C + (15mA)(8.5V)(90°C/W) = 82°C However, for 3.3V and other low voltage outputs, additional circuitry is required to derive INTVCC power from the output. The following list summarizes the four possible connections for EXTVCC: 1. EXTVCC Left Open (or Grounded). This will cause INTVCC to be powered from the internal 5.1V regulator resulting in an efficiency penalty of up to 10% at high input voltages. 2. EXTVCC Connected directly to VOUT . This is the normal connection for a 5V to 14V regulator and provides the highest efficiency. 3. EXTVCC Connected to an External supply. If an external supply is available in the 5V to 14V range, it may be used to power EXTVCC providing it is compatible with the MOSFET gate drive requirements. Ensure that EXTVCC < VIN. 4. EXTVCC Connected to an Output-Derived Boost Network. For 3.3V and other low voltage regulators, efficiency gains can still be realized by connecting EXTVCC to an output-derived voltage that has been boosted to greater than 4.7V. This can be done with the capacitive charge pump shown in Figure 9. Ensure that EXTVCC < VIN. 38901f 21 LTC3890-1 APPLICATIONS INFORMATION CIN VIN MTOP TG1 1/2 LTC3890-1 EXTVCC SW MBOT BG1 D COUT L NDS7002 RSENSE VOUT BAT85 BAT85 BAT85 output voltage falls below 70% of its nominal output level, then the maximum sense voltage is progressively lowered from 100% to 45% of its maximum selected value. Under short-circuit conditions with very low duty cycles, the LTC3890-1 will begin cycle skipping in order to limit the short-circuit current. In this situation the bottom MOSFET will be dissipating most of the power but less than in normal operation. The short-circuit ripple current is determined by the minimum on-time. tON(MIN), of the LTC3890-1 (≈90ns), the input voltage and inductor value: IL(SC) = tON(MIN) VIN L PGND 38901 F09 Figure 9. Capacitive Charge Pump for EXTVCC The resulting average short-circuit current is: ISC = 45% • ILIM(MAX) – 1 I 2 L(SC) Topside MOSFET Driver Supply (CB, DB) External bootstrap capacitors, CB, connected to the BOOST pins supply the gate drive voltages for the topside MOSFETs. Capacitor CB in the Functional Diagram is charged though external diode DB from INTVCC when the SW pin is low. When one of the topside MOSFETs is to be turned on, the driver places the CB voltage across the gate-source of the desired MOSFET. This enhances the top MOSFET switch and turns it on. The switch node voltage, SW, rises to VIN and the BOOST pin follows. With the topside MOSFET on, the boost voltage is above the input supply: VBOOST = VIN + VINTVCC. The value of the boost capacitor, CB, needs to be 100 times that of the total input capacitance of the topside MOSFET(s). The reverse breakdown of the external Schottky diode must be greater than VIN(MAX). When adjusting the gate drive level, the final arbiter is the total input current for the regulator. If a change is made and the input current decreases, then the efficiency has improved. If there is no change in input current, then there is no change in efficiency. Fault Conditions: Current Limit and Current Foldback The LTC3890-1 includes current foldback to help limit load current when the output is shorted to ground. If the Fault Conditions: Overvoltage Protection (Crowbar) The overvoltage crowbar is designed to blow a system input fuse when the output voltage of the regulator rises much higher than nominal levels. The crowbar causes huge currents to flow, that blow the fuse to protect against a shorted top MOSFET if the short occurs while the controller is operating. A comparator monitors the output for overvoltage conditions. The comparator detects faults greater than 10% above the nominal output voltage. When this condition is sensed, the top MOSFET is turned off and the bottom MOSFET is turned on until the overvoltage condition is cleared. The bottom MOSFET remains on continuously for as long as the overvoltage condition persists; if VOUT returns to a safe level, normal operation automatically resumes. A shorted top MOSFET will result in a high current condition which will open the system fuse. The switching regulator will regulate properly with a leaky top MOSFET by altering the duty cycle to accommodate the leakage. 38901f 22 LTC3890-1 APPLICATIONS INFORMATION Phase-Locked Loop and Frequency Synchronization The LTC3890-1 has an internal phase-locked loop (PLL) comprised of a phase frequency detector, a lowpass filter, and a voltage-controlled oscillator (VCO). This allows the turn-on of the top MOSFET of controller 1 to be locked to the rising edge of an external clock signal applied to the PLLIN/MODE pin. The turn-on of controller 2’s top MOSFET is thus 180 degrees out of phase with the external clock. The phase detector is an edge sensitive digital type that provides zero degrees phase shift between the external and internal oscillators. This type of phase detector does not exhibit false lock to harmonics of the external clock. If the external clock frequency is greater than the internal oscillator’s frequency, fOSC, then current is sourced continuously from the phase detector output, pulling up the VCO input. When the external clock frequency is less than fOSC, current is sunk continuously, pulling down the VCO input. If the external and internal frequencies are the same but exhibit a phase difference, the current sources turn on for an amount of time corresponding to the phase difference. The voltage at the VCO input is adjusted until the phase and frequency of the internal and external oscillators are identical. At the stable operating point, the phase detector output is high impedance and the internal filter capacitor, CLP, holds the voltage at the VCO input. Note that the LTC3890-1 can only be synchronized to an external clock whose frequency is within range of the LTC3890-1’s internal VCO, which is nominally 55kHz to 1MHz. This is guaranteed to be between 75kHz and 850kHz. Typically, the external clock (on the PLLIN/MODE pin) input high threshold is 1.6V, while the input low threshold is 1.1V. Rapid phase locking can be achieved by using the FREQ pin to set a free-running frequency near the desired synchronization frequency. The VCO’s input voltage is prebiased at a frequency corresponding to the frequency set by the FREQ pin. Once prebiased, the PLL only needs to adjust the frequency slightly to achieve phase lock and synchronization. Although it is not required that the free-running frequency be near external clock frequency, doing so will 1000 900 800 FREQUENCY (kHz) 700 600 500 400 300 200 100 0 15 25 35 45 55 65 75 85 95 105 115 125 FREQ PIN RESISTOR (kΩ) 38901 F10 Figure 10. Relationship Between Oscillator Frequency and Resistor Value at the FREQ Pin prevent the operating frequency from passing through a large range of frequencies as the PLL locks. Table 2 summarizes the different states in which the FREQ pin can be used. Table 2 FREQ PIN 0V INTVCC Resistor Any of the Above PLLIN/MODE PIN DC Voltage DC Voltage DC Voltage External Clock FREQUENCY 350kHz 535kHz 50kHz to 900kHz Phase Locked to External Clock Minimum On-Time Considerations Minimum on-time, tON(MIN), is the smallest time duration that the LTC3890-1 is capable of turning on the top MOSFET. It is determined by internal timing delays and the gate charge required to turn on the top MOSFET. Low duty cycle applications may approach this minimum on-time limit and care should be taken to ensure that: tON(MIN) < VOUT VIN f () If the duty cycle falls below what can be accommodated by the minimum on-time, the controller will begin to skip cycles. The output voltage will continue to be regulated, but the ripple voltage and current will increase. 38901f 23 LTC3890-1 APPLICATIONS INFORMATION The minimum on-time for the LTC3890-1 is approximately 90ns. However, as the peak sense voltage decreases the minimum on-time gradually increases up to about TBDns. This is of particular concern in forced continuous applications with low ripple current at light loads. If the duty cycle drops below the minimum on-time limit in this situation, a significant amount of cycle skipping can occur with correspondingly larger current and voltage ripple. Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC3890-1 circuits: 1) IC VIN current, 2) INTVCC regulator current, 3) I2R losses, 4) topside MOSFET transition losses. 1. The VIN current is the DC supply current given in the Electrical Characteristics table, which excludes MOSFET driver and control currents. VIN current typically results in a small (1μF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT , causing a rapid drop in VOUT . No regulator can alter its delivery of current quickly enough to prevent this sudden step change in output voltage if the load switch resistance is low and it is driven quickly. If the ratio of CLOAD to COUT is greater than 1:50, the switch rise time should be controlled so that the load rise time is limited to approximately 25 • CLOAD. Thus a 10μF capacitor would require a 250μs rise time, limiting the charging current to about 200mA. 38901f 25 LTC3890-1 APPLICATIONS INFORMATION Design Example As a design example for one channel, assume VIN = 12V(nominal), VIN = 22V (max), VOUT = 3.3V, IMAX = 5A, VSENSE(MAX) = 75mV and f = 350kHz. The inductance value is chosen first based on a 30% ripple current assumption. The highest value of ripple current occurs at the maximum input voltage. Tie the FREQ pin to GND, generating 350kHz operation. The minimum inductance for 30% ripple current is: V V IL = OUT 1– OUT ( f) (L) VIN(NOM) A 4.7μH inductor will produce 29% ripple current. The peak inductor current will be the maximum DC value plus one half the ripple current, or 5.73A. Increasing the ripple current will also help ensure that the minimum on-time of 95ns is not violated. The minimum on-time occurs at maximum VIN: tON(MIN) = VOUT VIN(MAX ) ( f) = 3.3V = 429ns 22V (350kHz ) A short-circuit to ground will result in a folded back current of: ISC = 34mV 1 95ns (22V ) – = 3.18A 2 4.7μH 0.01 with a typical value of RDS(ON) and δ = (0.005/°C)(25°C) = 0.125. The resulting power dissipated in the bottom MOSFET is: PSYNC = (3.18A ) (1.125) (0.022Ω) = 250mW which is less than under full-load conditions. CIN is chosen for an RMS current rating of at least 3A at temperature assuming only this channel is on. COUT is chosen with an ESR of 0.02Ω for low output ripple. The output ripple in continuous mode will be highest at the maximum input voltage. The output voltage ripple due to ESR is approximately: VORIPPLE = RESR (ΔIL) = 0.02Ω(1.45A) = 29mVP-P PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the IC. These items are also illustrated graphically in the layout diagram of Figure 11. Figure 12 illustrates the current waveforms present in the various branches of the 2-phase synchronous regulators operating in the continuous mode. Check the following in your layout: 1. Are the top N-channel MOSFETs MTOP1 and MTOP2 located within 1cm of each other with a common drain connection at CIN? Do not attempt to split the input decoupling for the two channels as it can cause a large resonant loop. 2. Are the signal and power grounds kept separate? The combined IC signal ground pin and the ground return of CINTVCC must return to the combined COUT (–) terminals. The path formed by the top N-channel MOSFET, Schottky diode and the CIN capacitor should have short 2 The equivalent RSENSE resistor value can be calculated by using the minimum value for the maximum current sense threshold (43mV): RSENSE ≤ 64mV ≈ 0.01Ω 5.73A Choosing 1% resistors: RA = 25k and RB = 78.7k yields an output voltage of 3.32V. The power dissipation on the topside MOSFET can be easily estimated. Choosing a Fairchild FDS6982S dual MOSFET results in: RDS(ON) = 0.035Ω/0.022Ω, CMILLER = 215pF. At maximum input voltage with T(estimated) = 50°C: PMAIN = 2 3.3V (5A) 1+ (0.005) (50°C – 25°C) 22V 2 (0.035 ) + (22V) 5A (2.5 ) (215pF ) • 2 1 1 + (350kHz) = 331mW 5V – 2.3V 2.3V 38901f 26 LTC3890-1 APPLICATIONS INFORMATION leads and PC trace lengths. The output capacitor (–) terminals should be connected as close as possible to the (–) terminals of the input capacitor by placing the capacitors next to each other and away from the Schottky loop described above. 3. Do the LTC3890-1 VFB pins’ resistive dividers connect to the (+) terminals of COUT? The resistive divider must be connected between the (+) terminal of COUT and signal ground. The feedback resistor connections should not be along the high current input feeds from the input capacitor(s). 4. Are the SENSE– and SENSE+ leads routed together with minimum PC trace spacing? The filter capacitor between SENSE+ and SENSE– should be as close as possible to the IC. Ensure accurate current sensing with Kelvin connections at the SENSE resistor. 5. Is the INTVCC decoupling capacitor connected close to the IC, between the INTVCC and the power ground pins? This capacitor carries the MOSFET drivers’ current peaks. An additional 1μF ceramic capacitor placed immediately next to the INTVCC and PGND pins can help improve noise performance substantially. 6. Keep the switching nodes (SW1, SW2), top gate nodes (TG1, TG2), and boost nodes (BOOST1, BOOST2) away from sensitive small-signal nodes, especially from the opposites channel’s voltage and current sensing feedback pins. All of these nodes have very large and fast moving signals and therefore should be kept on the output side of the LTC3890-1 and occupy minimum PC trace area. 7. Use a modified star ground technique: a low impedance, large copper area central grounding point on the same side of the PC board as the input and output capacitors with tie-ins for the bottom of the INTVCC decoupling capacitor, the bottom of the voltage feedback resistive divider and the SGND pin of the IC. PC Board Layout Debugging Start with one controller on at a time. It is helpful to use a DC-50MHz current probe to monitor the current in the inductor while testing the circuit. Monitor the output switching node (SW pin) to synchronize the oscilloscope to the internal oscillator and probe the actual output voltage as well. Check for proper performance over the operating voltage and current range expected in the application. The frequency of operation should be maintained over the input voltage range down to dropout and until the output load drops below the low current operation threshold—typically 15% of the maximum designed current level in Burst Mode operation. The duty cycle percentage should be maintained from cycle to cycle in a well-designed, low noise PCB implementation. Variation in the duty cycle at a subharmonic rate can suggest noise pickup at the current or voltage sensing inputs or inadequate loop compensation. Overcompensation of the loop can be used to tame a poor PC layout if regulator bandwidth optimization is not required. Only after each controller is checked for its individual performance should both controllers be turned on at the same time. A particularly difficult region of operation is when one controller channel is nearing its current comparator trip point when the other channel is turning on its top MOSFET. This occurs around 50% duty cycle on either channel due to the phasing of the internal clocks and may cause minor duty cycle jitter. Reduce VIN from its nominal level to verify operation of the regulator in dropout. Check the operation of the undervoltage lockout circuit by further lowering VIN while monitoring the outputs to verify operation. Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems coincide with high input voltages and low output currents, look for capacitive coupling between the BOOST, SW, TG, 38901f 27 LTC3890-1 APPLICATIONS INFORMATION and possibly BG connections and the sensitive voltage and current pins. The capacitor placed across the current sensing pins needs to be placed immediately adjacent to the pins of the IC. This capacitor helps to minimize the effects of differential noise injection due to high frequency capacitive coupling. If problems are encountered with high current output loading at lower input voltages, look for inductive coupling between CIN, Schottky and the top MOSFET components to the sensitive current and voltage sensing traces. In addition, investigate common ground path voltage pickup between these components and the SGND pin of the IC. An embarrassing problem, which can be missed in an otherwise properly working switching regulator, results when the current sensing leads are hooked up backwards. The output voltage under this improper hookup will still be maintained but the advantages of current mode control will not be realized. Compensation of the voltage loop will be much more sensitive to component selection. This behavior can be investigated by temporarily shorting out the current sensing resistor—don’t worry, the regulator will still maintain control of the output voltage. 38901f 28 LTC3890-1 APPLICATIONS INFORMATION ITH1 VFB1 SENSE1+ TRACK/SS1 PGOOD1 TG1 RPU1 VPULL-UP PGOOD1 L1 RSENSE VOUT1 SW1 SENSE1– LTC3890-1 BOOST1 BG1 FREQ fIN PLLIN/MODE RUN1 RUN2 SGND SENSE2– SENSE2+ VFB2 ITH2 TRACK/SS2 EXTVCC INTVCC BG2 BOOST2 VIN CB1 M1 M2 D1 RIN CVIN PGND 1μF CERAMIC COUT1 + GND + CINTVCC VIN 1μF CERAMIC M3 CB2 M4 CIN + COUT2 D2 SW2 TG2 L2 RSENSE VOUT2 38901 F11 Figure 11. Recommended Printed Circuit Layout Diagram 38901f 29 LTC3890-1 APPLICATIONS INFORMATION SW1 L1 RSENSE1 VOUT1 D1 COUT1 RL1 VIN RIN CIN SW2 L2 RSENSE2 VOUT2 BOLD LINES INDICATE HIGH SWITCHING CURRENT. KEEP LINES TO A MINIMUM LENGTH. D2 COUT2 RL2 38901 F12 Figure 12. Branch Current Waveforms 38901f 30 LTC3890-1 TYPICAL APPLICATIONS SENSE1+ RB1 100k RA1 31.6k CITH1A 100pF CITH1 1000pF RITH1 34.8k ITH1 CSS1 0.01μF LTC3890-1 TRACK/SS1 TG1 D1 VIN PLLIN/MODE SGND EXTVCC RUN1 RUN2 FREQ TRACK/SS2 ITH2 RA2 10.5k VFB2 RB2 100k C2 1nF SENSE2+ SENSE2– BG2 MBOT2 INTVCC PGND D2 TG2 CB2 0.1μF BOOST2 SW2 L2 8μH RSENSE2 10mΩ VOUT2 8.5V 3A COUT2 330μF MTOP2 CINT 4.7μF CIN 220μF VIN 9V TO 60V C1 1nF SENSE1– VFB1 BG1 SW1 BOOST1 CB1 0.1μF MTOP1 MBOT1 L1 4.7μH RSENSE1 8mΩ COUT1 470μF VOUT1 3.3V 5A 100k PGOOD1 INTVCC VOUT2 RFREQ 41.2k CSS2 0.01μF CITH2 470pF RITH2 34.8k 38901 TA07a MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB L1: COILCRAFT SER1360-472KL L2: COILCRAFT SER1360-802KL COUT1: SANYO 6TPE470M COUT2: SANYO 10TPE330M D1, D2: DFLS1100 Figure 13. High Efficiency Dual 8.5V/3.3V Step-Down Converter Efficiency and Power Loss vs Output Current 100 VIN = 12V 90 VOUT = 3.3V 80 EEFICIENCY (%) 70 60 50 40 30 20 10 0 0.0001 PULSE-SKIPPING LOSS BURST LOSS 10 CCM LOSS 100 10000 BURST EFFICIENCY 1000 POWER LOSS (mW) EEFICIENCY (%) 100 90 80 70 60 50 40 30 20 10 0.1 10 38901 TA07b Efficiency vs Load Current 100 VOUT = 8.5V VOUT = 3.3V 98 96 EEFICIENCY (%) 94 92 90 88 86 84 82 80 10 38901 TA07c Efficiency vs Input Voltage VOUT2 = 8.5V VOUT1 = 3.3V CCM EFFICIENCY PULSE-SKIPPING EFFICIENCY 0.001 0.01 0.1 1 OUTPUT CURRENT (A) 1 VIN = 12V 0 0.0001 0.001 0.01 0.1 1 OUTPUT CURRENT (A) ILOAD = 2A 0 5 10 15 20 25 30 35 40 45 50 55 60 INPUT VOLTAGE (V) 38901 TA07d 38901f 31 LTC3890-1 TYPICAL APPLICATIONS High Efficiency 8.5V Dual-Phase Step-Down Converter SENSE1+ RB1 100k RA1 10.5k CITH1A 100pF C1 1nF SENSE1– VFB1 BG1 SW1 ITH1 CITH1 C SS1 0.01μF 470pF BOOST1 LTC3890-1 TRACK/SS1 TG1 D1 INTVCC RMODE 100k PLLIN/MODE SGND VIN RRUN V 1000k OUT EXTVCC RUN1 RUN2 FREQ TRACK/SS2 BOOST2 ITH2 SW2 BG2 PGND D2 TG2 VIN INTVCC INTVCC 100k PGOOD1 MBOT1 L1 8μH RSENSE1 10mΩ CB1 0.1μF MTOP1 VIN CIN 9V TO 60V 220μF CINT 4.7μF VOUT1 8.5V 6A COUT1 330μF RITH1 34.8k RFREQ 41.2k CITH2 100pF MTOP2 CB2 0.1μF L2 8μH RSENSE2 10mΩ COUT2 330μF VFB2 SENSE2– C2 1nF SENSE2+ MBOT2 38901 TA03 MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB L1, L2: COILCRAFT SER1360-802KL COUT1, COUT2: SANYO 10TPE330M D1, D2: DFLS1100 38901f 32 LTC3890-1 TYPICAL APPLICATIONS High Efficiency Dual 12V/5V Step-Down Converter SENSE1+ RB1 100k RA1 6.98k CITH1A 100pF CITH1 470pF RITH1 34.8k ITH1 CSS1 0.01μF LTC3890-1 TRACK/SS1 TG1 D1 VIN PLLIN/MODE SGND EXTVCC RUN1 RUN2 FREQ CSS2 0.01μF TRACK/SS2 CITH2 470pF RITH2 20k ITH2 BG2 RA2 18.7k VFB2 SENSE2– RB2 100k C2 1nF SENSE2+ BOOST2 SW2 INTVCC PGND D2 TG2 C1 1nF SENSE1– VFB1 BG1 SW1 BOOST1 100k PGOOD1 INTVCC MBOT1 L1 8μH CB1 0.47μF MTOP1 RSENSE1 9mΩ COUT1 180μF VOUT1 12V 3A CIN 220μF CINT 4.7μF VIN 12.5V TO 60V RFREQ 41.2k MTOP2 CB2 0.47μF L2 4.7μH RSENSE2 10mΩ VOUT2 5V 5A COUT2 470μF MBOT2 38901 TA04 MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB L1: COILCRAFT SER1360-802KL L2: COILCRAFT SER1360-472KL COUT1: 16SVP180MX COUT2: SANYO 6TPE470M D1, D2: DFLS1100 38901f 33 LTC3890-1 TYPICAL APPLICATIONS High Efficiency Dual 24V/5V Step-Down Converter RB1 487k CF1 33pF RA1 16.9k CITH1A 100pF CITH1 680pF RITH1 46k C1 1nF SENSE1+ SENSE1– VFB1 BG1 SW1 ITH1 BOOST1 LTC3890-1 TRACK/SS1 TG1 D1 VIN PLLIN/MODE SGND EXTVCC RUN1 RUN2 FREQ INTVCC PGND D2 TG2 100k PGOOD1 INTVCC MBOT1 L1 22μH RSENSE1 25mΩ VOUT1 24V 1A COUT1 22μF ×2 CERAMIC CSS1 0.01μF CB1 0.47μF MTOP1 VIN 28V TO 60V CIN 220μF CINT 4.7μF RFREQ 60k CSS2 0.01μF CITH2 470pF RITH2 20k MTOP2 CB2 0.47μF L2 4.7μH RSENSE2 10mΩ TRACK/SS2 ITH2 BOOST2 SW2 BG2 VOUT2 5V 5A COUT2 470μF MBOT2 RA2 18.7k VFB2 RB2 100k C2 1nF SENSE2– SENSE2+ 38901 TA05 MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB L1: SUMIDA CDR7D43MN L2: COILCRAFT SER1360-472KL COUT1: KEMET T525D476MO16E035 COUT2: SANYO 6TPE470M D1, D2: DFLS1100 38901f 34 LTC3890-1 PACKAGE DESCRIPTION GN Package 28-Lead Plastic SSOP (Narrow .150 Inch) (Reference LTC DWG # 05-08-1641) .386 – .393* (9.804 – 9.982) 28 27 26 25 24 23 22 21 20 19 18 17 1615 .045 .005 .033 (0.838) REF .254 MIN .150 – .165 .229 – .244 (5.817 – 6.198) .150 – .157** (3.810 – 3.988) .0165 .0015 RECOMMENDED SOLDER PAD LAYOUT .0250 BSC 1 45 23 4 56 7 8 9 10 11 12 13 14 .004 – .0098 (0.102 – 0.249) .015 .004 (0.38 0.10) .0075 – .0098 (0.19 – 0.25) 0 – 8 TYP .0532 – .0688 (1.35 – 1.75) .016 – .050 (0.406 – 1.270) NOTE: 1. CONTROLLING DIMENSION: INCHES INCHES 2. DIMENSIONS ARE IN (MILLIMETERS) .008 – .012 (0.203 – 0.305) TYP .0250 (0.635) BSC GN28 (SSOP) 0204 3. DRAWING NOT TO SCALE *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 38901f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 35 LTC3890-1 TYPICAL APPLICATION High Efficiency Dual 12V/3.3V Step-Down Converter SENSE1+ RB1 100k RA1 6.98k CITH1A 100pF CITH1 470pF RITH1 34.8k ITH1 CSS1 0.01μF LTC3890-1 TRACK/SS1 TG1 D1 VIN PLLIN/MODE SGND EXTVCC RUN1 RUN2 FREQ CSS2 0.01μF TRACK/SS2 CITH2 1000pF CITH2A 100pF RITH2 34.8k ITH2 BG2 MBOT2 BOOST2 SW2 INTVCC PGND D2 TG2 CB2 0.47μF L2 4.7μH RSENSE2 10mΩ VOUT2 3.3V 5A COUT2 470μF MTOP2 CINT 4.7μF CIN 220μF VIN 12.5V TO 60V C1 1nF SENSE1– VFB1 BG1 SW1 BOOST1 CB1 0.47μF MTOP1 MBOT1 L1 8μH RSENSE1 9mΩ COUT1 180μF VOUT1 12V 3A 100k PGOOD1 INTVCC RFREQ 41.2k RA2 31.6k VFB2 SENSE2– RB2 100k C2 1nF SENSE2+ MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB L1: COILCRAFT SER1360-802KL L2: COILCRAFT SER1360-472KL COUT1: 16SVP180MX COUT2: SANYO 6TPE470M D1, D2: DFLS1100 38901 TA06 RELATED PARTS PART NUMBER DESCRIPTION LTC3857/LTC3857-1 Low IQ, Dual Output 2-Phase Synchronous Step-Down DC/DC Controllers with 99% Duty Cycle LTC3858/LTC3858-1 Low IQ, Dual Output 2-Phase Synchronous Step-Down DC/DC Controllers with 99% Duty Cycle LTC3868/LTC3868-1 Low IQ, Dual Output 2-Phase Synchronous Step-Down DC/DC Controller with 99% Duty Cycle LTC3834/LTC3834-1 Low IQ, Synchronous Step-Down DC/DC Controller with 99% Duty Cycle LTC3835/LTC3835-1 Low IQ, Synchronous Step-Down DC/DC Controller with 99% Duty Cycle LT3845 Low IQ, High Voltage Synchronous Step-Down DC/DC Controller COMMENTS Phase-Lockable Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 24V, IQ = 50μA Phase-Lockable Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 24V, IQ = 170μA Phase-Lockable Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 24V, 0.8V ≤ VOUT ≤ 14V, IQ = 170μA Phase-Lockable Fixed Frequency 140kHz to 650kHz, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 10V, IQ = 30μA Phase-Lockable Fixed Frequency 140kHz to 650kHz, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 10V, IQ = 80μA Adjustable Fixed Frequency 100kHz to 500kHz, 4V ≤ VIN ≤ 60V, 1.23V ≤ VOUT ≤ 36V, IQ = 120μA, TSSOP-16 38901f 36 Linear Technology Corporation (408) 432-1900 ● LT 0210 • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2010