LTC6409CUDBTRMPBF

FeaTures n n n n n n n n n n n n LTC6409 10GHz GBW, 1.1nV/√Hz Differential Amplifier/ADC Driver DescripTion The LTC®6409 is a very high speed, low distortion, differential amplifier. Its input common mode range includes ground, so that a ground-referenced input signal can be DC-coupled, level-shifted, and converted to drive an ADC differentially. The gain and feedback resistors are external, so that the exact gain and frequency response can be tailored to each application. For example, the amplifier could be externally compensated in a no-overshoot configuration, which is desired in certain time-domain applications. The LTC6409 is stable in a differential gain of 1. This allows for a low output noise in applications where gain is not desired. It draws 52mA of supply current and has a hardware shutdown feature which reduces current consumption to 100µA. The LTC6409 is available in a compact 3mm × 2mm 10-pin leadless QFN package and operates over a –40°C to 125°C temperature range. L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. 10GHz Gain-Bandwidth Product 88dB SFDR at 100MHz, 2VP-P 1.1nV/√Hz Input Noise Density Input Range Includes Ground External Resistors Set Gain (Min 1V/V) 3300V/µs Differential Slew Rate 52mA Supply Current 2.7V to 5.25V Supply Voltage Range Fully Differential Input and Output Adjustable Output Common Mode Voltage Low Power Shutdown Small 10-Lead 3mm × 2mm × 0.75mm QFN Package applicaTions n n n n n Differential Pipeline ADC Driver High-Speed Data-Acquisition Cards Automated Test Equipment Time Domain Reflexometry Communications Receivers Typical applicaTion DC-Coupled Interface from a Ground-Referenced Single-Ended Input to an LTC2262-14 ADC 1.3pF VIN LTC6409 Driving LTC2262-14 ADC, fIN = 70MHz, –1dBFS, fS = 150MHz, 4096-Point FFT 0 VS = 3.3V –10 V OUTDIFF = 1.8VP-P –20 HD2 = –86.5dBc HD3 = –89.4dBc –30 SFDR = 81.6dB –40 SNR = 71.1dB –50 –60 –70 –80 –90 –100 –110 –120 0 10 20 30 40 50 FREQUENCY (MHz) 60 70 6409 TA01b 150 150 1.8V 3.3V 39pF 33.2 10 AIN+ AIN– VDD AMPLITUDE (dBFS) –+ VOCM = 0.9V LTC6409 +– 150 150 33.2 10 39pF LTC2262-14 ADC GND 6409 TA01 1.3pF 6409fa 1 LTC6409 absoluTe MaxiMuM raTings (Note 1) pin conFiguraTion TOP VIEW V– 10 –OUT +IN 1 2 3 SHDN 4 V+ 5 VOCM V+ 9 11,V– V– 8 7 6 +OUT –IN Total Supply Voltage (V+ – V–) .................................5.5V Input Current (+IN, –IN, VOCM, SHDN) (Note 2) ................................................................ ±10mA Output Short-Circuit Duration (Note 3) ............ Indefinite Operating Temperature Range (Note 4).................................................. – 40°C to 125°C Specified Temperature Range (Note 5).................................................. – 40°C to 125°C Maximum Junction Temperature .......................... 150°C Storage Temperature Range .................. – 65°C to 150°C UDB PACKAGE 10-LEAD (3mm × 2mm) PLASTIC QFN TJMAX = 150°C, θJA = 138°C/W, θJC = 5.2°C/W EXPOSED PAD (PIN 11) CONNECTED TO V– orDer inForMaTion Lead Free Finish TAPE AND REEL (MINI) LTC6409CUDB#TRMPBF LTC6409IUDB#TRMPBF LTC6409HUDB#TRMPBF TAPE AND REEL LTC6409CUDB#TRPBF LTC6409IUDB#TRPBF PART MARKING* LFPF LFPF PACKAGE DESCRIPTION 10-Lead (3mm × 2mm) Plastic QFN 10-Lead (3mm × 2mm) Plastic QFN SPECIFIED TEMPERATURE RANGE 0°C to 70°C –40°C to 85°C –40°C to 125°C LTC6409HUDB#TRPBF LFPF 10-Lead (3mm × 2mm) Plastic QFN TRM = 500 pieces. *Temperature grades are identified by a label on the shipping container. Consult LTC Marketing for parts specified with wider operating temperature ranges. Consult LTC Marketing for information on lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. V+ = 5V, V– = 0V, VCM = VOCM = VICM = 1.25V, VSHDN = open. VS is defined as (V+ – V–). VOUTCM is defined as (V+OUT + V–OUT)/2. VICM is defined as (V+IN + V–IN)/2. VOUTDIFF is defined as (V+OUT – V–OUT). SYMBOL PARAMETER VOSDIFF Differential Offset Voltage (Input Referred) CONDITIONS VS = 3V VS = 3V VS = 5V VS = 5V VS = 3V VS = 5V VS = 3V VS = 5V VS = 3V VS = 5V Common Mode Differential Mode Differential Mode f = 1MHz, Not Including RI/RF Noise f = 1MHz, Not Including RI/RF Noise Shunt-Terminated to 50Ω, RS = 50Ω, RI = 25Ω, RF = 10kΩ l elecTrical characTerisTics MIN TYP ±300 ±300 MAX ±1000 ±1200 ±1100 ±1400 UNITS µV µV µV µV µV/°C µV/°C l l l l l l l ΔVOSDIFF Differential Offset Voltage Drift (Input Referred) ΔT IB IOS RIN CIN en in NF Input Bias Current (Note 6) Input Offset Current (Note 6) Input Resistance Input Capacitance Differential Input Noise Voltage Density Input Noise Current Density Noise Figure at 100MHz 2 2 –140 –160 –62 –70 ±2 ±2 165 860 0.5 1.1 8.8 6.9 0 0 ±10 ±10 µA µA µA µA kΩ Ω pF nV/√Hz pA/√Hz dB 6409fa 2 LTC6409 elecTrical characTerisTics SYMBOL PARAMETER enVOCM VICMR (Note 7) CMRRI (Note 8) Common Mode Noise Voltage Density Input Signal Common Mode Range Input Common Mode Rejection Ratio (Input Referred) ΔVICM/ΔVOSDIFF The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. V+ = 5V, V– = 0V, VCM = VOCM = VICM = 1.25V, VSHDN = open. VS is defined as (V+ – V–). VOUTCM is defined as (V+OUT + V–OUT)/2. VICM is defined as (V+IN + V–IN)/2. VOUTDIFF is defined as (V+OUT – V–OUT). CONDITIONS f = 10MHz VS = 3V VS = 5V VS = 3V, VICM from 0V to 1.5V VS = 5V, VICM from 0V to 3.5V VS = 3V, VOCM from 0.5V to 1.5V VS = 5V, VOCM from 0.5V to 3.5V VS = 2.7V to 5.25V VS = 2.7V to 5.25V l l l l l l l l l MIN 0 0 75 75 55 60 60 55 2.7 TYP 12 MAX 1.5 3.5 UNITS nV/√Hz V V dB dB dB dB dB dB 90 90 80 85 85 70 5.25 1 1 ±0.1 ±0.1 –65 –70 ±1 ±1 4 ±0.3 ±0.3 –50 –50 ±5 ±6 CMRRIO Output Common Mode Rejection Ratio (Input (Note 8) Referred) ΔVOCM/ΔVOSDIFF PSRR (Note 9) Differential Power Supply Rejection (ΔVS/ΔVOSDIFF) PSRRCM Output Common Mode Power Supply Rejection (Note 9) (ΔVS/ΔVOSCM) Supply Voltage Range (Note 10) VS GCM ΔGCM BAL Common Mode Gain (ΔVOUTCM/ΔVOCM) Common Mode Gain Error, 100 × (GCM – 1) Output Balance (ΔVOUTCM/ ΔVOUTDIFF) Common Mode Offset Voltage (VOUTCM – VOCM) Common Mode Offset Voltage Drift V V/V V/V % % dB dB mV mV µV/°C VS = 3V, VOCM from 0.5V to 1.5V VS = 5V, VOCM from 0.5V to 3.5V VS = 3V, VOCM from 0.5V to 1.5V VS = 5V, VOCM from 0.5V to 3.5V ΔVOUTDIFF = 2V Single-Ended Input Differential Input VS = 3V VS = 5V l l l l l l l l l VOSCM ΔVOSCM ΔT VOUTCMR Output Signal Common Mode Range (Note 7) (Voltage Range for the VOCM Pin) RINVOCM Input Resistance, VOCM Pin VOCM VOUT Self-Biased Voltage at the VOCM Pin Output Voltage, High, Either Output Pin VS = 3V VS = 5V VS = 3V, VOCM = Open VS = 5V, VOCM = Open VS = 3V, IL = 0 VS = 3V, IL = –20mA VS = 5V, IL = 0 VS = 5V, IL = –20mA VS = 3V, 5V; IL = 0 VS = 3V, 5V; IL = 20mA VS = 3V VS = 5V l l l l l l l l l l l l 0.5 0.5 30 0.9 1.85 1.8 3.85 3.8 40 0.85 1.25 2 1.95 4 3.95 0.06 0.2 ±50 ±70 ±70 ±95 65 52 1.5 3.5 50 1.6 V V KΩ V V V V V V Output Voltage, Low, Either Output Pin ISC AVOL IS ISHDN RSHDN VIL VIH tON tOFF Output Short-Circuit Current, Either Output Pin (Note 11) Large-Signal Open Loop Voltage Gain Supply Current 0.15 0.4 V V mA mA dB l 56 58 500 185 0.6 mA mA µA KΩ V V ns ns Supply Current in Shutdown SHDN Pull-Up Resistor SHDN Input Logic Low SHDN Input Logic High Turn-On Time Turn-Off Time VSHDN ≤ 0.6V VSHDN = 0V to 0.5V l l l l 100 115 1.4 160 80 150 6409fa 3 LTC6409 elecTrical characTerisTics SYMBOL PARAMETER SR Slew Rate The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. V+ = 5V, V– = 0V, VCM = VOCM = VICM = 1.25V, VSHDN = open. VS is defined as (V+ – V–). VOUTCM is defined as (V+OUT + V–OUT)/2. VICM is defined as (V+IN + V–IN)/2. VOUTDIFF is defined as (V+OUT – V–OUT). CONDITIONS Differential Output, VOUTDIFF = 4VP-P +OUT Rising (–OUT Falling) +OUT Falling (–OUT Rising) RI = 25Ω, RF = 10kΩ, fTEST = 100MHz l MIN TYP 3300 1720 1580 MAX UNITS V/µs V/µs V/µs GHz GHz GHz MHz MHz GBW f–3dB f0.1dB FPBW HD2 HD3 Gain-Bandwidth Product –3dB Frequency Frequency for 0.1dB Flatness Full Power Bandwidth 25MHz Distortion 9.5 8 10 2 600 550 RI = RF = 150Ω, RLOAD = 400Ω, CF = 1.3pF RI = RF = 150Ω, RLOAD = 400Ω , CF = 1.3pF VOUTDIFF = 2VP-P Differential Input, VOUTDIFF = 2VP-P, RI = RF = 150Ω, RLOAD = 400Ω 2nd Harmonic 3rd Harmonic Differential Input, VOUTDIFF = 2VP-P, RI = RF = 150Ω, RLOAD = 400Ω 2nd Harmonic 3rd Harmonic Single-Ended Input, VOUTDIFF = 2VP-P, RI = RF = 150Ω, RLOAD = 400Ω 2nd Harmonic 3rd Harmonic Single-Ended Input, VOUTDIFF = 2VP-P, RI = RF = 150Ω, RLOAD = 400Ω 2nd Harmonic 3rd Harmonic VOUTDIFF = 2VP-P Envelope, RI = RF = 150Ω, RLOAD = 400Ω VOUTDIFF = 2VP-P Envelope, RI = RF = 150Ω, RLOAD = 400Ω VOUTDIFF = 2VP-P Envelope, RI = RF = 150Ω, RLOAD = 400Ω –104 –106 dBc dBc 100MHz Distortion –93 –88 dBc dBc HD2 HD3 25MHz Distortion –101 –103 dBc dBc 100MHz Distortion –88 –93 –110 –98 –88 59 53 48 dBc dBc dBc dBc dBc dBm dBm dBm IMD3 3rd Order IMD at 25MHz f1 = 24.9MHz, f2 = 25.1MHz 3rd Order IMD at 100MHz f1 = 99.9MHz, f2 = 100.1MHz 3rd Order IMD at 140MHz f1 = 139.9MHz, f2 = 140.1MHz OIP3 Equivalent OIP3 at 25MHz (Note 12) Equivalent OIP3 at 100MHz (Note 12) Equivalent OIP3 at 140MHz (Note 12) Settling Time VOUTDIFF = 2VP-P Step, RI = RF = 150Ω, RLOAD = 400Ω 1% Settling tS 1.9 ns Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: Input pins (+IN, –IN, VOCM, and SHDN) are protected by steering diodes to either supply. If the inputs should exceed either supply voltage, the input current should be limited to less than 10mA. In addition, the inputs +IN, –IN are protected by a pair of back-to-back diodes. If the differential input voltage exceeds 1.4V, the input current should be limited to less than 10mA. Note 3: A heat sink may be required to keep the junction temperature below the absolute maximum rating when the output is shorted indefinitely. Note 4: The LTC6409C/LTC6409I are guaranteed functional over the temperature range of –40°C to 85°C. The LTC6409H is guaranteed functional over the temperature range of –40°C to 125°C. Note 5: The LTC6409C is guaranteed to meet specified performance from 0°C to 70°C. The LTC6409C is designed, characterized and expected to meet specified performance from –40°C to 85°C, but is not tested or QA sampled at these temperatures. The LTC6409I is guaranteed to meet specified performance from –40°C to 85°C. The LTC6409H is guaranteed to meet specified performance from –40°C to 125°C. Note 6: Input bias current is defined as the average of the input currents flowing into the inputs (–IN and +IN). Input offset current is defined as the difference between the input currents (IOS = IB+ – IB–). 6409fa 4 LTC6409 elecTrical characTerisTics Note 7: Input common mode range is tested by testing at both VICM = 1.25V and at the Electrical Characteristics table limits to verify that the differential offset (VOSDIFF) and the common mode offset (VOSCM) have not deviated by more than ±1mV and ±2mV respectively from the VICM = 1.25V case. The voltage range for the output common mode range is tested by applying a voltage on the VOCM pin and testing at both VOCM = 1.25V and at the Electrical Characteristics table limits to verify that the common mode offset (VOSCM) has not deviated by more than ±6mV from the VOCM = 1.25V case. Note 8: Input CMRR is defined as the ratio of the change in the input common mode voltage at the pins +IN or –IN to the change in differential input referred offset voltage. Output CMRR is defined as the ratio of the change in the voltage at the VOCM pin to the change in differential input referred offset voltage. This specification is strongly dependent on feedback ratio matching between the two outputs and their respective inputs and it is difficult to measure actual amplifier performance (See Effects of Resistor Pair Mismatch in the Applications Information section of this data sheet). For a better indicator of actual amplifier performance independent of feedback component matching, refer to the PSRR specification. Note 9: Differential power supply rejection (PSRR) is defined as the ratio of the change in supply voltage to the change in differential input referred offset voltage. Common mode power supply rejection (PSRRCM) is defined as the ratio of the change in supply voltage to the change in the output common mode offset voltage. Note 10: Supply voltage range is guaranteed by power supply rejection ratio test. Note 11: Extended operation with the output shorted may cause the junction temperature to exceed the 150°C limit. Note 12: Refer to Relationship Between Different Linearity Metrics in the Applications Information section of this data sheet for information on how to calculate an equivalent OIP3 from IMD3 measurements. Typical perForMance characTerisTics Differential Input Offset Voltage vs Temperature 1.5 2.0 Differential Input Offset Voltage vs Input Common Mode Voltage COMMON MODE OFFSET VOLTAGE (mV) VS = 5V VOCM = 1.25V 1.5 RI = RF = 150 0.1% FEEDBACK NETWORK RESISTORS REPRESENTATIVE UNIT 1.0 0.5 0 –0.5 2.5 2.0 1.5 1.0 0.5 0 Common Mode Offset Voltage vs Temperature DIFFERENTIAL VOS (mV) DIFFERENTIAL VOS (mV) 1.0 0.5 VS = 5V VOCM = VICM = 1.25V RI = RF = 150 FIVE REPRESENTATIVE UNITS 0 VS = 5V VOCM = VICM = 1.25V RI = RF = 150 FIVE REPRESENTATIVE UNITS –0.5 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 –1.0 TA = 85°C TA = 70°C TA = 25°C TA = 0°C TA = –40°C 0 0.5 1 1.5 2 2.5 3 3.5 INPUT COMMON MODE VOLTAGE (V) 4 –0.5 –50 –25 6409 G01 6409 G02 0 25 50 75 TEMPERATURE (°C) 100 125 6409 G03 Supply Current vs Supply Voltage 60 55 TOTAL SUPPLY CURRENT (mA) 50 45 40 35 30 25 20 15 10 5 0 TA = 125°C TA = 85°C TA = 70°C TA = 25°C TA = 0°C TA = –40°C 0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 SUPPLY VOLTAGE (V) 6409 G04 Supply Current vs SHDN Voltage 60 55 TOTAL SUPPLY CURRENT (mA) 50 45 40 35 30 25 20 15 10 5 0 0 0.5 1 1.5 2 2.5 3 3.5 SHDN VOLTAGE (V) TA = 125°C TA = 85°C TA = 70°C TA = 25°C TA = 0°C TA = –40°C 4 4.5 5 VS = 5V SHUTDOWN SUPPLY CURRENT (µA) 140 120 100 80 60 40 20 0 Shutdown Supply Current vs Supply Voltage TA = 125°C TA = 85°C TA = 70°C TA = 25°C TA = 0°C TA = –40°C VSHDN = OPEN VSHDN = V– 0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 SUPPLY VOLTAGE (V) 6409 G06 6409 G05 6409fa 5 LTC6409 Typical perForMance characTerisTics Differential Output Voltage Noise vs Frequency 1000 VOLTAGE NOISE DENSITY (nV/√Hz) INPUT VOLTAGE NOISE DENSITY (nV/√Hz) VS = 5V RI = RF = 150 INCLUDES RI/RF NOISE 1000 Input Noise Density vs Frequency VS = 5V 1000 INPUT CURRENT NOISE DENSITY (pA/√Hz) 1000 Differential Output Impedance vs Frequency VS = 5V RI = RF = 150 100 100 100 OUTPUT IMPEDANCE ( ) 100 10 in 10 en 10 1 10 0.1 1 1 1k 1M FREQUENCY (Hz) 1G 6409 G07 1 1 1k 1M FREQUENCY (Hz) 1 1G 6409 G18 0.01 1 10 100 1000 FREQUENCY (MHz) 10000 6409 G09 CMRR vs Frequency 100 90 80 90 70 PSRR (dB) 60 50 40 30 20 10000 6409 G10 Differential PSRR vs Frequency Small Signal Step Response –OUT +OUT CMRR (dB) 80 20mV/DIV 70 VS = 5V 60 VOCM = 1.25V RI = RF = 150 , CF = 1.3pF 0.1% FEEDBACK NETWORK RESISTORS 50 1 10 100 1000 FREQUENCY (MHz) VS = 5V VOCM = VICM = 1.25V RLOAD = 400 10 VS = 5V 1 10 100 1000 FREQUENCY (MHz) 10000 6409 G11 RI = RF = 150 , CF = 1.3pF CL = 0pF VIN = 200mVP-P, DIFFERENTIAL 2ns/DIV 6409 G12 Large Signal Step Response 4.0 –OUT 3.5 3.0 VOLTAGE (V) 2.5 2.0 1.5 1.0 0.5 6409 G13 Overdriven Output Transient Response –OUT 0.2V/DIV +OUT VS = 5V RLOAD = 400 VIN = 2VP-P, DIFFERENTIAL 2ns/DIV VS = 5V VOCM = 1.25V RLOAD = 200 TO GROUND PER OUTPUT 0 +OUT 20ns/DIV 6409 G14 6409fa 6 LTC6409 Typical perForMance characTerisTics Frequency Response vs Closed Loop Gain 60 50 40 30 GAIN (dB) 20 10 0 –10 AV = 400 AV = 100 AV = 20 AV = 10 AV = 5 AV = 2 AV = 1 AV (V/V) RI ( ) 1 2 5 10 20 100 400 150 100 50 50 25 25 25 RF ( ) CF (pF) 150 1.3 200 1 250 0.8 500 0.4 500 0.4 2.5k 0 10k 0 20 Frequency Response vs Load Capacitance CL = 0pF CL = 0.5pF CL = 1pF CL = 1.5pF CL = 2pF 10 GAIN (dB) 0 –10 VS = 5V VOCM = VICM = 1.25V RLOAD = 400 –20 RI = RF = 150 , CF = 1.3pF CAPACITOR VALUES ARE FROM EACH OUTPUT TO GROUND. NO SERIES RESISTORS ARE USED. –30 10 100 1000 FREQUENCY (MHz) VS = 5V –20 VOCM = VICM = 1.25V RLOAD = 400 –30 1 10 100 1000 FREQUENCY (MHz) 10000 6409 G15 10000 6409 G16 Gain 0.1dB Flatness 0.5 0.4 0.3 SLEW RATE (V/µs) 0.2 GAIN (dB) 0.1 0 –0.1 –0.2 VS = 5V VOCM = VICM = 1.25V –0.4 RLOAD = 400 RI = RF = 150 , CF = 1.3pF –0.5 1 10 100 1000 FREQUENCY (MHz) –0.3 3400 3375 3350 3325 3300 3275 3250 3225 10000 6409 G17 Slew Rate vs Temperature VS = 5V 3200 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 6409 G08 Harmonic Distortion vs Frequency –50 VS = 5V VOCM = VICM = 1.25V –60 R LOAD = 400 RI = RF = 150 –70 VOUTDIFF = 2VP-P DIFFERENTIAL INPUTS –80 HD2 –90 –100 –110 –120 1 10 100 FREQUENCY (MHz) 1000 6409 G19 Harmonic Distortion vs Output Common Mode Voltage –30 –40 –50 DISTORTION (dBc) –60 –70 –80 –90 –100 –110 0.5 HD2 HD3 VS = 5V fIN = 100MHz RLOAD = 400 RI = RF = 150 VOUTDIFF = 2VP-P DIFFERENTIAL INPUTS –80 Harmonic Distortion vs Input Amplitude VS = 5V VOCM = VICM = 1.25V fIN = 100MHz RLOAD = 400 –90 RI = RF = 150 DIFFERENTIAL INPUTS –100 HD3 DISTORTION (dBc) DISTORTION (dBc) HD3 HD2 –110 1 1.5 2 2.5 3 3.5 OUTPUT COMMON MODE VOLTAGE (V) 6409 G20 –120 –2 –4 (0.4VP-P) 0 2 4 6 INPUT AMPLITUDE (dBm) 8 10 (2VP-P) 6409 G21 6409fa 7 LTC6409 Typical perForMance characTerisTics Harmonic Distortion vs Frequency –50 VS = 5V VOCM = VICM = 1.25V –60 R LOAD = 400 RI = RF = 150 –70 VOUTDIFF = 2VP-P SINGLE-ENDED INPUT –80 –90 HD2 –100 –110 –120 1 HD3 10 100 FREQUENCY (MHz) 1000 6409 G22 Harmonic Distortion vs Output VS = 5V VOCM VICM = 1.25V Common Mode=Voltage –30 –40 –50 DISTORTION (dBc) –60 –70 –80 –90 –100 –110 0.5 HD3 HD2 RLOAD = 400 RI 5 R VS = = V F = 150 VOUTDIFF = fIN = 100MHz2VP-P SINGLE-ENDED INPUT RLOAD = 400 RI = RF = 150 VOUTDIFF = 2VP-P SINGLE-ENDED INPUT –80 Harmonic Distortion vs Input Amplitude VS = 5V VOCM = VICM = 1.25V fIN = 100MHz –90 DISTORTION (dBc) HD2 DISTORTION (dBc) –100 HD3 –110 RLOAD = 400 RI = RF = 150 SINGLE-ENDED INPUT 0 2 4 6 INPUT AMPLITUDE (dBm) 8 10 (2VP-P) 6409 G24 1 1.5 2 2.5 3 3.5 OUTPUT COMMON MODE VOLTAGE (V) 6409 G23 –120 –2 –4 (0.4VP-P) Intermodulation Distortion vs Frequency –50 VS = 5V VOCM = VICM = 1.25V –60 R LOAD = 400 RI = RF = 150 –70 2 TONES, 200kHz TONE SPACING, 2VP-P COMPOSITE –80 DIFFERENTIAL INPUTS –90 –100 –110 –120 10 100 FREQUENCY (MHz) 1000 6409 G25 Intermodulation Distortion vs Output Common Mode Voltage –30 VS = 5V –40 fIN = 100MHz RLOAD = 400 RI = RF = 150 –50 2 TONES, 200kHz TONE SPACING, 2VP-P COMPOSITE –60 DIFFERENTIAL INPUTS –70 –80 –90 –100 –110 0.5 1 1.5 2 2.5 3 3.5 OUTPUT COMMON MODE VOLTAGE (V) 6409 G26 Intermodulation Distortion vs Input Amplitude –80 VS = 5V VOCM = VICM = 1.25V fIN = 100MHz RLOAD = 400 –90 RI = RF = 150 2 TONES, 200kHz TONE SPACING DIFFERENTIAL INPUTS –100 THIRD ORDER IMD (dBc) THIRD ORDER IMD (dBc) THIRD ORDER IMD (dBc) –110 –120 2 (0.8VP-P) 4 6 8 INPUT AMPLITUDE (dBm) 10 (2VP-P) 6409 G27 pin FuncTions +IN, –IN (Pins 2, 6): Non-Inverting and Inverting Input Pins. SHDN (Pin 3): When SHDN is floating or directly tied to V+, the LTC6409 is in the normal (active) operating mode. When the SHDN pin is connected to V–, the part is disabled and draws approximately 100µA of supply current. V+, V– (Pins 4, 9 and Pins 8, 10): Positive and Negative Power Supply Pins. Similar pins should be connected to the same voltage. VOCM (Pin 5): Output Common Mode Reference Voltage. The voltage on this pin sets the output common mode voltage level. If left floating, an internal resistor divider develops a default voltage of 1.25V with a 5V supply. +OUT, –OUT (Pins 7, 1): Differential Output Pins. Exposed Pad (Pin 11): Tie the bottom pad to V–. If split supplies are used, DO NOT tie the pad to ground. 6409fa 8 LTC6409 block DiagraM 2 +IN 1 –OUT V– SHDN V+ 200k 5 VOCM V– 4 V+ V+ –IN +OUT 6409 BD 3 V+ 10 V– + V+ 9 V+ 50k – V– V– 8 V– 6 7 applicaTions inForMaTion Functional Description The LTC6409 is a small outline, wideband, high speed, low noise, and low distortion fully-differential amplifier with accurate output phase balancing. The amplifier is optimized to drive low voltage, single-supply, differential input analogto-digital converters (ADCs). The LTC6409 input common mode range includes ground, which makes it ideal to DC-couple and convert ground-referenced, single-ended signals into differential signals that are referenced to the user-supplied output common mode voltage. This is ideal for driving these differential ADCs. The balanced differential nature of the amplifier also provides even-order harmonic distortion cancellation, and low susceptibility to common mode noise (like power supply noise). The LTC6409 can operate with a single-ended input and differential output, or with a differential input and differential output. The outputs of the LTC6409 are capable of swinging from close-to-ground to 1V below V+. They can source or sink up to approximately 70mA of current. Load capacitances should be decoupled with at least 10Ω of series resistance from each output. Input Pin Protection The LTC6409 input stage is protected against differential input voltages which exceed 1.4V by two pairs of series diodes connected back to back between +IN and –IN. Moreover, the input pins, as well as VOCM and SHDN pins, have clamping diodes to either power supply. If these pins are driven to voltages which exceed either supply, the current should be limited to 10mA to prevent damage to the IC. SHDN Pin The SHDN pin is a CMOS logic input with a 150k internal pull-up resistor. If the pin is driven low, the LTC6409 powers down. If the pin is left unconnected or driven high, the part is in normal active operation. Some care should be taken to control leakage currents at this pin to prevent inadvertently putting the LTC6409 into shutdown. The turn-on and turn-off time between the shutdown and active states is typically less than 200ns. General Amplifier Applications In Figure 1, the gain to VOUTDIFF from VINP and VINM is given by: VOUTDIFF = V+OUT – V–OUT ≈ RF • ( VINP – VINM ) RI (1) Note from Equation (1), the differential output voltage (V+OUT – V–OUT) is completely independent of input and output common mode voltages, or the voltage at the common mode pin. This makes the LTC6409 ideally 6409fa 9 LTC6409 applicaTions inForMaTion RI V+IN RF + VINP V–OUT – VVOCM + VOCM that can be processed is even wider. The input common mode range at the op amp inputs depends on the circuit configuration (gain), VOCM and VCM (refer to Figure 1). For fully differential input applications, where VINP = –VINM, the common mode input is approximately: VICM = V+IN + V–IN RI RF ≈ VOCM • + VCM • 2 RI + RF RI + RF + VCM – – VINM – + RI V–IN RF 6409 F01 V+OUT Figure 1. Circuit for Common Mode Range suited for pre-amplification, level shifting and conversion of single-ended signals to differential output signals for driving differential input ADCs. Output Common Mode and VOCM Pin The output common mode voltage is defined as the average of the two outputs: VOUTCM = VOCM = V+OUT + V–OUT 2 With single-ended inputs, there is an input signal component to the input common mode voltage. Applying only VINP (setting VINM to zero), the input common mode voltage is approximately: VICM = V+IN + V–IN RI ≈ VOCM • + 2 RI + RF RF V RF VCM • + INP • RI + RF 2 RI + RF (2) This means that if, for example, the input signal (VINP) is a sine, an attenuated version of that sine signal also appears at the op amp inputs. Input Impedance and Loading Effects The low frequency input impedance looking into the VINP or VINM input of Figure 1 depends on how the inputs are driven. For fully differential input sources (VINP = –VINM), the input impedance seen at either input is simply: RINP = RINM = RI For single-ended inputs, because of the signal imbalance at the input, the input impedance actually increases over the balanced differential case. The input impedance looking into either input is: RINP = RINM = RF 1 1– • 2 RI + RF RI As the equation shows, the output common mode voltage is independent of the input common mode voltage, and is instead determined by the voltage on the VOCM pin, by means of an internal common mode feedback loop. If the VOCM pin is left open, an internal resistor divider develops a default voltage of 1.25V with a 5V supply. The VOCM pin can be overdriven to another voltage if desired. For example, when driving an ADC, if the ADC makes a reference available for setting the common mode voltage, it can be directly tied to the VOCM pin, as long as the ADC is capable of driving the 40k input resistance presented by the VOCM pin. The Electrical Characteristics table specifies the valid range that can be applied to the VOCM pin (VOUTCMR). Input Common Mode Voltage Range The LTC6409’s input common mode voltage (VICM) is defined as the average of the two input pins, V+IN and V–IN. The valid range that can be used for VICM has been specified in the Electrical Characteristics table (VICMR). However, due to external resistive divider action of the gain and feedback resistors, the effective range of signals Input signal sources with non-zero output impedances can also cause feedback imbalance between the pair of feedback networks. For the best performance, it is recommended that the input source output impedance be compensated. If input impedance matching is required by the source, 6409fa 10 LTC6409 applicaTions inForMaTion a termination resistor RT should be chosen (see Figure 2) such that: R •R R T = INM S RINM – RS According to Figure 2, the input impedance looking into the differential amp (RINM) reflects the single-ended source case, given above. Also, R2 is chosen as: R2 = R T ||RS = R T • RS R T + RS RINM RS VS RT RI RF VINP RI2 V+IN RF2 + – VVOCM V–OUT + VOCM – VINM – RI1 V–IN RF1 6409 F03 + V+OUT Figure 3. Real-World Application with Feedback Resistor Pair Mismatch Δb is defined as the difference in the feedback factors: ∆β = + – RI2 RI1 – RI2 + RF2 RI1 + RF1 – RT CHOSEN SO THAT RT || RINM = RS R2 CHOSEN TO BALANCE RT || RS + RI RF Here, VCM and VINDIFF are defined as the average and the difference of the two input voltages VINP and VINM, respectively: 6409 F02 VCM = R2 = RS || RT VINP + VINM 2 VINDIFF = VINP – VINM Figure 2. Optimal Compensation for Signal Source Impedance Effects of Resistor Pair Mismatch Figure 3 shows a circuit diagram which takes into consideration that real world resistors will not match perfectly. Assuming infinite open loop gain, the differential output relationship is given by the equation: VOUTDIFF = V+OUT – V–OUT ≈ VINDIFF • VCM • ∆β ∆β – VOCM • β AVG β AVG RF + RI When the feedback ratios mismatch (Δb), common mode to differential conversion occurs. Setting the differential input to zero (VINDIFF = 0), the degree of common mode to differential conversion is given by the equation: VOUTDIFF = V+OUT – V–OUT ≈ (VCM – VOCM ) • ∆β (3) β AVG where RF is the average of RF1, and RF2, and RI is the average of RI1, and RI2. bAVG is defined as the average feedback factor from the outputs to their respective inputs: RI2  1  RI1 β AVG = •  + 2  RI1 + RF1 RI2 + RF2   In general, the degree of feedback pair mismatch is a source of common mode to differential conversion of both signals and noise. Using 0.1% resistors or better will mitigate most problems and will provide about 54dB worst case of common mode rejection. A low impedance ground plane should be used as a reference for both the input signal source and the VOCM pin. There may be concern on how feedback factor mismatch affects distortion. Feedback factor mismatch from using 1% resistors or better, has a negligible effect on distortion. However, in single supply level shifting applications where there is a voltage difference between the input common mode voltage and the output common mode voltage, 6409fa 11 LTC6409 applicaTions inForMaTion resistor mismatch can make the apparent voltage offset of the amplifier appear worse than specified. The apparent input referred offset induced by feedback factor mismatch is derived from Equation (3): VOSDIFF(APPARENT) ≈ (VCM – VOCM) • Δb Using the LTC6409 in a single 5V supply application with 0.1% resistors, the input common mode grounded, and the VOCM pin biased at 1.25V, the worst case mismatch can induce 1.25mV of apparent offset voltage. Noise and Noise Figure The LTC6409’s differential input referred voltage and current noise densities are 1.1nV/√Hz and 8.8pA/√Hz, respectively. In addition to the noise generated by the amplifier, the surrounding feedback resistors also contribute noise. A simplified noise model is shown in Figure 4. The output noise generated by both the amplifier and the feedback components is given by the equation:   RF   2 eni •  1+   + 2 • (in • RF ) +  RI     eno =  2  R 2 •  enRI • F  + 2 • enRF2 RI   If the circuits surrounding the amplifier are well balanced, common mode noise (enVOCM) of the amplifier does not appear in the differential output noise equation given above. A plot of this equation and a plot of the noise generated by the feedback components for the LTC6409 are shown in Figure 5. The LTC6409’s input referred voltage noise contributes the equivalent noise of a 75Ω resistor. When the feedback network is comprised of resistors whose values are larger than this, the output noise is resistor noise and amplifier current noise dominant. For feedback networks consisting of resistors with values smaller than 75Ω, the output noise is voltage noise dominant (see Figure 5). 2 in–2 enRI2 RI in+2 RF enRF2 + VOCM – eno2 enRI2 eni2 RI RF enRF2 6409 F04 Figure 4. Simplified Noise Model 1000 NOISE DENSITY (nV/√Hz) 100 TOTAL (AMPLIFIER AND FEEDBACK NETWORK) OUTPUT NOISE 10 FEEDBACK NETWORK NOISE 1 0.1 10 100 1000 RI = RF ( ) 10000 6409 F05 Figure 5. LTC6409 Output Noise vs Noise Contributed by Feedback Network Alone Lower resistor values always result in lower noise at the penalty of increased distortion due to increased loading by the feedback network on the output. Higher resistor values will result in higher output noise, but typically improved distortion due to less loading on the output. For this reason, when LTC6409 is configured in a differential gain of 1, using feedback resistors of at least 150Ω is recommended. To calculate noise figure (NF), a source resistance and the noise it generates should also come into consideration. Figure 6 shows a noise model for the amplifier which includes the source resistance (RS). To generalize the 6409fa 12 LTC6409 applicaTions inForMaTion enRI2 RI in+2 RF enRF2 Finally, noise figure can be obtained as:  eno2 NF = 10log  1+  eno2(RS)      RS RT + VOCM in–2 – eno2 enRS2 enRT2 enRI2 eni2 RI RF enRF2 6409 F06 Figure 6. A More General Noise Model Including Source and Termination Resistors Figure 7 specifies the measured total output noise (eno), excluding the noise contribution of source resistance, and noise figure (NF) of LTC6409 configured at closed loop gains (AV = RF /RI) of 1V/V, 2V/V and 5V/V. The circuits in the left column use termination resistors and transformers to match to the 50Ω source resistance, while the circuits in the right column do not have such matching. For simplicity, DC-blocking and bypass capacitors have not been shown in the circuits, as they do not affect the noise results. Relationship Between Different Linearity Metrics Linearity is, of course, an important consideration in many amplifier applications. This section relates the intermodulation distortion of fully differential amplifiers to other linearity metrics commonly used in RF style blocks. Intercept points are specifications that have long been used as key design criteria in the RF communications world as a metric for the intermodulation distortion performance of a device in the signal chain (e.g., amplifiers, mixers, etc.). Intercept points, like noise figures, can be easily cascaded back and forth through a signal chain to determine the overall performance of a receiver chain, thus resulting in simpler system-level calculations. Traditionally, these systems use primarily single-ended RF amplifiers as gain blocks designed to operate in a 50Ω environment, just like the rest of the receiver chain. Since intercept points are given in dBm, this implies an associated impedance of 50Ω. However, for LTC6409 as a differential feedback amplifier with low output impedance, a 50Ω resistive load is not required (unlike an RF amplifier). This distinction is important when evaluating the intercept point for LTC6409. In fact, the LTC6409 yields optimum distortion performance when loaded with 200Ω to 1kΩ (at each output), very similar to the input impedance of an ADC. As a result, terminating calculation, a termination resistor (RT) is included and its noise contribution is taken into account. Now, the total output noise power (excluding the noise contribution of RS) is calculated as:       RF 2   + 2 • (in • RF ) + eno2 = eni •  1+   R +  R T ||RS    I    2         RF  + 2 • enRF2 + 2 •  enRI •  R T ||RS    RI +    2     R enRT • F RI     2RI ||RS •   R T + ( 2RI ||RS )    2 2 2 Meanwhile, the output noise power due to noise of RS is given by: eno2(RS) = enRS    R •F RI   2RI ||R T •   RS + ( 2RI ||R T )    2 6409fa 13 LTC6409 applicaTions inForMaTion 1.3pF 1.3pF 150 1:4 150 150 150 50 VIN + 600 VOCM + – eno = 4.70nV/√Hz NF = 14.41dB 50 VIN + + – 150 VOCM eno = 5.88nV/√Hz NF = 17.59dB – 150 150 1.3pF – 150 1.3pF 1pF 1pF 100 1:4 200 100 200 50 VIN + VOCM + – 100 eno = 5.77nV/√Hz NF = 10.43dB 50 VIN + + – 100 VOCM eno = 9.76nV/√Hz NF = 16.66dB – 200 1pF – 200 1pF 0.4pF 0.8pF 100 1:4 500 50 250 50 VIN + VOCM + – 100 eno = 11.69nV/√Hz NF = 8.81dB 50 VIN + + – 50 VOCM eno = 14.23nV/√Hz NF = 13.56dB – 500 0.4pF – 250 0.8pF 6409 F07 Figure 7. LTC6409 Measured Output Noise and Noise Figure at Different Closed Loop Gains with and without Source Impedance Matching the input of the ADC to 50Ω can actually be detrimental to system performance. The definition of 3rd order intermodulation distortion (IMD3) is shown in Figure 8. Also, a graphical representation of how to relate IMD3 to output/input 3rd order intercept points (OIP3/IIP3) has been depicted in Figure 9. Based on this figure, Equation (4) gives the definition of the intercept point, relative to the intermodulation distortion. OIP3 = PO + IMD3 2 (4) PO is the output power of each of the two tones at which IMD3 is measured, as shown in Figure 9. It is calculated in dBm as:  V2  PDIFF PO = 10log  –3   2 • RL • 10  (5) where RL is the differential load resistance, and VPDIFF is the differential peak voltage for a single tone. Normally, intermodulation distortion is specified for a benchmark composite differential peak of 2VP-P at the output of the 6409fa 14 LTC6409 applicaTions inForMaTion ∆f = f2 – f1 = f1 – (2f1 – f2) = (2f2 – f1) – f2 PO PO results in a lower intercept point. Therefore, it is important to consider the impedance seen by the output of the LTC6409 when working with intercept points. Comparing linearity specifications between different amplifier types becomes easier when a common impedance level is assumed. For this reason, the intercept points for LTC6409 are reported normalized to a 50Ω load impedance. This is the reason why OIP3 in the Electrical Characteristics table is 4dBm more than half the absolute value of IMD3. If the top half of the LTC6409 demo board (DC1591A, shown in Figure 12) is used to measure IMD3 and OIP3, one should make sure to properly convert the power seen at the differential output of the amplifier to the power that appears at the single-ended output of the demo board. Figure 10 shows an equivalent representation of the top half of the demo board. This view ignores the DC-blocking and bypass capacitors, which do not affect the analysis here. The transmission line transformers (used mainly for impedance matching) are modeled here as ideal 4:1 impedance transformers together with a –1dB block. This separates the insertion loss of the transformer from its ideal behavior. The 100Ω resistors at the LTC6409 output create a differential 200Ω resistance, which is an impedance match for the reflected RL. As previously mentioned, IMD3 is measured for 2VP-P differential peak (i.e. 10dBm) at the output of the LTC6409, corresponding to 1VP-P (i.e. 4dBm) at each output alone. From LTC6409 output (location A in Figure 10) to the input of the output transformer (location B), there is a voltage attenuation of 1/2 (or –6dB) formed by the resistive divider CF RF POWER IMD3 = PS – PO PS PS 2f1 – f2 f1 f2 2f2 – f1 FREQUENCY 6409 F08 Figure 8. Definition of IMD3 POUT (dBm) OIP3 PO PS 1× IMD3 3× IIP3 PIN (dBm) 6409 F10 Figure 9. Graphical Representation of the Relationship between IMD3 and OIP3 amplifier, implying that each single tone is 1VP-P, resulting in VPDIFF = 0.5V. Using RL = 50Ω as the associated impedance, PO is calculated to be close to 4dBm. As seen in Equation (5), when a higher impedance is used, the same level of intermodulation distortion performance RS 50 RT 1dB LOSS IDEAL 1:4 RT RI RI RF CF LTC6409 A 100 B IDEAL 4:1 1dB LOSS 6409 F10 C RL 50 + – VS 100 Figure 10. Equivalent Schematic of the Top Half of the LTC6409 Demo Board 6409fa 15 LTC6409 applicaTions inForMaTion between the RL • 4 = 200Ω differential resistance seen at location B and the 200Ω formed by the two 100Ω matching resistors at the LTC6409 output. Thus, the differential power at location B is 10 – 6 = 4dBm. Since the transformer ratio is 4:1 and it has an insertion loss of about 1dB, the power at location C (across RL) is calculated to be 4 – 6 – 1 = –3dBm. This means that IMD3 should be measured while the power at the output of the demo board is –3dBm which is equivalent to having 2VP-P differential peak (or 10dBm) at the output of the LTC6409. GBW vs f–3dB Gain-bandwidth product (GBW) and –3dB frequency (f–3dB) have been both specified in the Electrical Characteristics table as two different metrics for the speed of the LTC6409. GBW is obtained by measuring the gain of the amplifier at a specific frequency (fTEST) and calculate gain • fTEST. To measure gain, the feedback factor (i.e. b = RI/(RI + RF)) is chosen sufficiently small so that the feedback loop does not limit the available gain of the LTC6409 at fTEST, ensuring that the measured gain is the open loop gain of the amplifier. As long as this condition is met, GBW is a parameter that depends only on the internal design and compensation of the amplifier and is a suitable metric to specify the inherent speed capability of the amplifier. f–3dB, on the other hand, is a parameter of more practical interest in different applications and is by definition the frequency at which the gain is 3dB lower than its low frequency value. The value of f–3dB depends on the speed of the amplifier as well as the feedback factor. Since the LTC6409 is designed to be stable in a differential signal gain of 1 (where RI = RF or b = 1/2), the maximum f–3dB is obtained and measured in this gain setting, as reported in the Electrical Characteristics table. In most amplifiers, the open loop gain response exhibits a conventional single-pole roll-off for most of the frequencies before crossover frequency and the GBW and f–3dB numbers are close to each other. However, the LTC6409 is intentionally compensated in such a way that its GBW is significantly larger than its f–3dB. This means that at lower frequencies (where the input signal frequencies typically lie, e.g. 100MHz) the amplifier’s gain and the thus the feedback loop gain is larger. This has the important advantage of further linearizing the amplifier and improving distortion at those frequencies. Looking at the Frequency Response vs Closed Loop Gain graph in the Typical Performance Characteristics section of this data sheet, one sees that for a closed loop gain (AV) of 1 (where RI = RF = 150Ω), f–3dB is about 2GHz. However, for AV = 400 (where RI = 25Ω and RF = 10kΩ), the gain at 100MHz is close to 40dB = 100V/V, implying a GBW value of 10GHz. Feedback Capacitors When the LTC6409 is configured in low differential gains, it is often advantageous to utilize a feedback capacitor (CF) in parallel with each feedback resistor (RF). The use of CF implements a pole-zero pair (in which the zero frequency is usually smaller than the pole frequency) and adds positive phase to the feedback loop gain around the amplifier. Therefore, if properly chosen, the addition of CF boosts the phase margin and improves the stability response of the feedback loop. For example, with RI = RF = 150Ω, it is recommended for most general applications to use CF = 1.3pF across each RF. This value has been selected to maximize f–3dB for the LTC6409 while keeping the peaking of the closed loop gain versus frequency response under a reasonable level (