LTNT

FEATURES s s LTC2411/LTC2411-1 24-Bit No Latency ∆ΣTM ADC with Differential Input and Reference in MSOP DESCRIPTIO The LTC®2411/LTC2411-1 are 2.7V to 5.5V micropower 24-bit differential ∆Σ analog-to-digital converters with an integrated oscillator, 2ppm INL and 0.29ppm RMS noise. They use delta-sigma technology and provide single cycle settling time for multiplexed applications. Through a single pin, the LTC2411 can be configured for better than 110dB differential mode rejection at 50Hz or 60Hz ± 2%, and the LTC2411-1 can provide better than 87dB input differential mode rejection over the range of 49Hz to 61.2Hz, or they can be driven by an external oscillator for a user-defined rejection frequency. The LTC2411 and LTC2411-1 are identical when driven by an external oscillator. The internal oscillator requires no external frequency setting components. The converters accept any external differential reference voltage from 0.1V to VCC for flexible ratiometric and remote sensing measurement configurations. The fullscale differential input range is from – 0.5VREF to 0.5VREF. The reference common mode voltage, VREFCM, and the input common mode voltage, VINCM, may be independently set anywhere within the GND to VCC range of the LTC2411/LTC2411-1. The DC common mode input rejection is better than 140dB. The LTC2411/LTC2411-1 communicate through a flexible 3-wire digital interface that is compatible with SPI and MICROWIRETM protocols. , LTC and LT are registered trademarks of Linear Technology Corporation. No Latency ∆Σ is a trademark of Linear Technology Corporation. MICROWIRE is a trademark of National Semiconductor Corporation. s s s s s s s s s 24-Bit ADC in an MS10 Package Low Supply Current (200µA in Conversion Mode and 4µA in Autosleep Mode) Differential Input and Differential Reference with GND to VCC Common Mode Range 2ppm INL, No Missing Codes 4ppm Full-Scale Error and 1ppm Offset 0.29ppm Noise No Latency: Digital Filter Settles in a Single Cycle. Each Conversion Is Accurate, Even After an Input Step Single Supply 2.7V to 5.5V Operation Internal Oscillator—No External Components Required 110dB Min, Pin Selectable 50Hz/60Hz Notch Filter (LTC2411) Simultaneous 50Hz/60Hz Rejection (LTC2411-1) APPLICATIO S s s s s s s s s s Direct Sensor Digitizer Weight Scales Direct Temperature Measurement Gas Analyzers Strain Gauge Transducers Instrumentation Data Acquisition Industrial Process Control 6-Digit DVMs TYPICAL APPLICATIO 2.7V TO 5.5V 1µF 1 VCC FO 10 LTC2411/ LTC2411-1 REFERENCE VOLTAGE 0.1V TO VCC ANALOG INPUT RANGE –0.5VREF TO 0.5VREF 2 3 4 5 6 REF + REF – IN + IN – GND 2411 TA01 VCC = INTERNAL OSC/50Hz REJECTION (LTC2411) = EXTERNAL CLOCK SOURCE = INTERNAL OSC/60Hz REJECTION (LTC2411) = SIMULTANEOUS 50Hz/60Hz REJECTION (LTC2411-1) BRIDGE IMPEDANCE 100Ω TO 10kΩ 4 5 SCK 9 3-WIRE SPI INTERFACE SDO CS 8 7 U U U VCC 1µF 2 IN 1 REF + VCC + 9 SCK 8 SDO 7 CS FO 10 2411 TA02 IN – 3 LTC2411/ LTC2411-1 3-WIRE SPI INTERFACE REF – GND 6 1 LTC2411/LTC2411-1 ABSOLUTE (Notes 1, 2) AXI U RATI GS PACKAGE/ORDER I FOR ATIO Supply Voltage (VCC) to GND .......................– 0.3V to 7V Analog Input Pins Voltage to GND .................................... – 0.3V to (VCC + 0.3V) Reference Input Pins Voltage to GND .................................... – 0.3V to (VCC + 0.3V) Digital Input Voltage to GND ........ – 0.3V to (VCC + 0.3V) Digital Output Voltage to GND ..... – 0.3V to (VCC + 0.3V) Operating Temperature Range LTC2411C ............................................... 0°C to 70°C LTC2411I ............................................ – 40°C to 85°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER TOP VIEW VCC REF + REF – IN + IN – 1 2 3 4 5 10 9 8 7 6 FO SCK SDO CS GND LTC2411CMS LTC2411IMS LTC2411-1CMS LTC2411-1IMS MS10 PART MARKING LTNS LTNT LTWV LTNN MS10 PACKAGE 10-LEAD PLASTIC MSOP TJMAX = 125°C, θJA = 120°C/W Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS PARAMETER Resolution (No Missing Codes) Integral Nonlinearity CONDITIONS The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. (Notes 3, 4) MIN q q q TYP 1 2 6 5 20 MAX UNITS Bits ppm of VREF ppm of VREF ppm of VREF µV nV/°C 0.1V ≤ VREF ≤ VCC, – 0.5 • VREF ≤ VIN ≤ 0.5 • VREF (Note 5) 4.5V ≤ VCC ≤ 5.5V, REF + = 2.5V, REF– = GND, VINCM = 1.25V (Note 6) 5V ≤ VCC ≤ 5.5V, REF + = 5V, REF – = GND, VINCM = 2.5V (Note 6) REF + = 2.5V, REF – = GND, VINCM = 1.25V (Note 6) 2.5V ≤ REF + ≤ VCC, REF – = GND, GND ≤ IN + = IN – ≤ VCC (Note 14) 2.5V ≤ REF + ≤ VCC, REF – = GND, GND ≤ IN + = IN – ≤ VCC 2.5V ≤ REF + ≤ VCC, REF – = GND, IN + = 0.75REF +, IN – = 0.25 • REF + 2.5V ≤ REF + ≤ VCC, REF – = GND, IN + = 0.75REF +, IN – = 0.25 • REF + 2.5V ≤ REF + ≤ VCC, REF – = GND, IN + = 0.25 • REF +, IN – = 0.75 • REF + 2.5V ≤ REF + ≤ VCC, REF – = GND, IN + = 0.25 • REF+, IN – = 0.75 • REF + 4.5V ≤ VCC ≤ 5.5V, REF + = 2.5V, REF – = GND, VINCM = 1.25V 5V ≤ VCC ≤ 5.5V, REF + = 5V, REF– = GND, VINCM = 2.5V REF + = 2.5V, REF – = GND, VINCM = 1.25V 5V ≤ VCC ≤ 5.5V, REF + = 5V, VREF – = GND, GND ≤ IN – = IN + ≤ 5V, (Note 13) 24 14 20 Offset Error Offset Error Drift Positive Full-Scale Error Positive Full-Scale Error Drift Negative Full-Scale Error Negative Full-Scale Error Drift Total Unadjusted Error q 4 0.04 12 ppm of VREF ppm of VREF/°C q 4 0.04 3 3 6 1.45 12 ppm of VREF ppm of VREF/°C ppm of VREF ppm of VREF ppm of VREF µVRMS Output Noise 2 U W U U WW W LTC2411/LTC2411-1 CO VERTER CHARACTERISTICS PARAMETER Input Common Mode Rejection DC Input Common Mode Rejection 60Hz ± 2% (LTC2411) Input Common Mode Rejection 50Hz ± 2% (LTC2411) Input Common Mode Rejection 49Hz to 61.2Hz (LTC2411-1) Input Normal Mode Rejection 60Hz ± 2% (LTC2411) Input Normal Mode Rejection 50Hz ± 2% (LTC2411) Input Normal Mode Rejection 49Hz to 61.2Hz (LTC2411-1) Reference Common Mode Rejection DC Power Supply Rejection, DC Power Supply Rejection, 60Hz ± 2% (LTC2411) CONDITIONS GND ≤ 2.5V ≤ REF + ≤ V The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. (Notes 3, 4) MIN q q q q q q q q Power Supply Rejection, 50Hz ± 2% REF + = 2.5V, REF – = GND, IN – = IN + = GND, (Note 8) (LTC2411) Power Supply Rejection, 49Hz to 61.2Hz (LTC2411-1) REF + = 2.5V, REF – = GND, IN – = IN + = GND, (Note 15) A ALOG I PUT A D REFERE CE SYMBOL IN + IN – VIN REF + REF – VREF CS (IN +) CS CS (IN –) (REF +) (IN +) (IN –) (REF +) PARAMETER Absolute/Common Mode IN + Absolute/Common Mode IN – Voltage Voltage The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. (Note 3) CONDITIONS q q q q q q Input Differential Voltage Range (IN + – IN –) Absolute/Common Mode REF + Voltage Absolute/Common Mode REF – Voltage Reference Differential Voltage Range (REF + – REF –) IN + Sampling Capacitance IN – Sampling Capacitance REF + Sampling Capacitance REF – Sampling Capacitance IN + IN – DC Leakage Current DC Leakage Current CS = VCC CS = VCC CS = VCC = 5.5V, IN + = GND = 5.5V, IN – = GND = 5.5V, REF + = 5V CS (REF –) IDC_LEAK IDC_LEAK IDC_LEAK REF + DC Leakage Current REF – DC Leakage Current IDC_LEAK (REF –) U U U U TYP 140 MAX UNITS dB dB dB dB – CC, REF = GND, – = IN + ≤ 5V IN 130 140 140 140 110 110 87 130 2.5V ≤ REF+ ≤ VCC, REF – = GND, GND ≤ IN – = IN + ≤ 5V, (Note 7) 2.5V ≤ REF + ≤ VCC, REF – = GND, GND ≤ IN – = IN + ≤ 5V, (Note 8) 2.5V < REF + < VCC, REF – = GND, GND < IN– = IN+ < VCC (Note 15) (Note 7) (Note 8) (Note 15) 2.5V ≤ REF+ ≤ VCC, GND ≤ REF – ≤ 2.5V, VREF = 2.5V, IN – = IN + = GND REF + = 2.5V, REF – = GND, IN – = IN + = GND REF + = 2.5V, REF – = GND, IN – = IN + = GND, (Note 7) 140 140 dB dB dB 140 110 120 120 120 dB dB dB dB dB U MIN GND – 0.3V GND – 0.3V – VREF/2 0.1 GND 0.1 TYP MAX VCC + 0.3V VCC + 0.3V VREF/2 VCC VCC – 0.1V VCC UNITS V V V V V V pF pF pF pF 6 6 6 6 q q q q –10 –10 –10 –10 1 1 1 1 10 10 10 10 nA nA nA nA CS = VCC = 5.5V, REF – = GND 3 LTC2411/LTC2411-1 DIGITAL I PUTS A D DIGITAL OUTPUTS SYMBOL VIH VIL VIH VIL IIN IIN CIN CIN VOH VOL VOH VOL IOZ PARAMETER High Level Input Voltage CS, FO Low Level Input Voltage CS, FO High Level Input Voltage SCK Low Level Input Voltage SCK Digital Input Current CS, FO Digital Input Current SCK Digital Input Capacitance CS, FO Digital Input Capacitance SCK High Level Output Voltage SDO Low Level Output Voltage SDO High Level Output Voltage SCK Low Level Output Voltage SCK Hi-Z Output Leakage SDO (Note 9) IO = – 800µA IO = 1.6mA IO = – 800µA (Note 10) IO = 1.6mA (Note 10) q q q q q The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. (Note 3) CONDITIONS 2.7V ≤ VCC ≤ 5.5V 2.7V ≤ VCC ≤ 3.3V 4.5V ≤ VCC ≤ 5.5V 2.7V ≤ VCC ≤ 5.5V 2.7V ≤ VCC ≤ 5.5V (Note 9) 2.7V ≤ VCC ≤ 3.3V (Note 9) 4.5V ≤ VCC ≤ 5.5V (Note 9) 2.7V ≤ VCC ≤ 5.5V (Note 9) 0V ≤ VIN ≤ VCC 0V ≤ VIN ≤ VCC (Note 9) q q q q q q POWER REQUIRE E TS SYMBOL VCC ICC PARAMETER Supply Voltage Supply Current Conversion Mode Sleep Mode Sleep Mode The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. (Note 3) CONDITIONS q 4 UW U U MIN 2.5 2.0 TYP MAX UNITS V V 0.8 0.6 2.5 2.0 0.8 0.6 –10 –10 10 10 VCC – 0.5V 0.4 VCC – 0.5V 0.4 –10 10 10 10 V V V V V V µA µA pF pF V V V V µA MIN 2.7 TYP MAX 5.5 UNITS V µA µA µA CS = 0V (Note 12) CS = VCC (Note 12) CS = VCC, 2.7V ≤ VCC ≤ 3.3V (Note 12) q q 200 4 2 300 10 LTC2411/LTC2411-1 TI I G CHARACTERISTICS SYMBOL fEOSC tHEO tLEO tCONV PARAMETER External Oscillator Frequency Range External Oscillator High Period External Oscillator Low Period Conversion Time FO = 0V (LTC2411) FO = VCC (LTC2411) FO = 0V (LTC2411-1) External Oscillator (Note 11) Internal Oscillator (LTC2411) (Note 10) Internal Oscillator (LTC2411-1) (Note 10) External Oscillator (Notes 10, 11) (Note 10) (Note 9) (Note 9) (Note 9) Internal Oscillator (LTC2411) (Notes 10, 12) Internal Oscillator (LTC2411-1) (Notes 10, 12) External Oscillator (Notes 10, 11) q q q q q q q q q q The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. (Note 3) CONDITIONS q q q q q q q fISCK DISCK fESCK tLESCK tHESCK tDOUT_ISCK tDOUT_ESCK t1 t2 t3 t4 tKQMAX tKQMIN t5 t6 Note 1: Absolute Maximum Ratings are those values beyond which the life of the device may be impaired. Note 2: All voltage values are with respect to GND. Note 3: VCC = 2.7 to 5.5V unless otherwise specified. VREF = REF + – REF –, VREFCM = (REF + + REF –)/2; VIN = IN + – IN –, VINCM = (IN + + IN –)/2. Note 4: FO pin tied to GND or to VCC or to external conversion clock source with fEOSC = 153600Hz unless otherwise specified. Note 5: Guaranteed by design, not subject to test. Note 6: Integral nonlinearity is defined as the deviation of a code from a straight line passing through the actual endpoints of the transfer curve. The deviation is measured from the center of the quantization band. Note 7: FO = 0V (internal oscillator) or fEOSC = 153600Hz ± 2% (external oscillator). UW MIN 2.56 0.25 0.25 TYP MAX 2000 390 390 UNITS kHz µs µs ms ms ms ms kHz kHz kHz 130.86 133.53 136.20 157.03 160.23 163.44 143.78 146.71 149.64 20510/fEOSC (in kHz) 19.2 17.5 fEOSC/8 45 250 250 1.64 1.80 1.67 1.70 1.83 1.86 256/fEOSC (in kHz) 32/fESCK (in kHz) 0 0 0 50 220 15 50 50 200 200 200 55 2000 Internal SCK Frequency Internal SCK Duty Cycle External SCK Frequency Range External SCK Low Period External SCK High Period Internal SCK 32-Bit Data Output Time % kHz ns ns ms ms ms ms ns ns ns ns ns ns ns ns External SCK 32-Bit Data Output Time (Note 9) CS ↓ to SDO Low Z CS ↑ to SDO High Z CS ↓ to SCK ↓ CS ↓ to SCK ↑ SCK ↓ to SDO Valid SDO Hold After SCK ↓ SCK Set-Up Before CS ↓ SCK Hold After CS ↓ (Note 5) (Note 10) (Note 9) q q q q q q Note 8: FO = VCC (internal oscillator) or fEOSC = 128000Hz ± 2% (external oscillator). Note 9: The converter is in external SCK mode of operation such that the SCK pin is used as digital input. The frequency of the clock signal driving SCK during the data output is fESCK and is expressed in kHz. Note 10: The converter is in internal SCK mode of operation such that the SCK pin is used as digital output. In this mode of operation the SCK pin has a total equivalent load capacitance CLOAD = 20pF. Note 11: The external oscillator is connected to the FO pin. The external oscillator frequency, fEOSC, is expressed in kHz. Note 12: The converter uses the internal oscillator. FO = 0V or FO = VCC. Note 13: The output noise includes the contribution of the internal calibration operations. Note 14: Guaranteed by design and test correlation. Note 15: FO = 0V (internal oscillator) or fEOSC = 139800Hz ±2% (external oscillator). 5 LTC2411/LTC2411-1 TYPICAL PERFOR A CE CHARACTERISTICS Total Unadjusted Error (VCC = 5V, VREF = 5V) 3 2 TUE (ppm OF VREF) TA = 90°C 1 0 –1 –2 TA = – 45°C VCC = 5V REF + = 5V REF – = GND VINCM = 2.5V FO = GND 1 1.5 2 2.5 TUE (ppm OF VREF) TA = 25°C 0.5 0 1.5 1.0 TUE (ppm OF VREF) TA = 25°C TA = – 45°C –3 –2.5 –2 –1.5 –1 –0.5 0 0.5 VIN (V) Integral Nonlinearity (VCC = 5V, VREF = 5V) 3 2 INL (ppm OF VREF) 1 0 –1 –2 –3 –2.5 –2 –1.5 –1 –0.5 0 0.5 VIN (V) VCC = 5V REF + = 5V REF – = GND VINCM = 2.5V FO = GND 1.5 1.0 INL (ppm OF VREF) 0.5 0 TA = – 45°C INL (ppm OF VREF) TA = 90°C TA = – 45°C 1 Noise Histogram 16 10,000 CONSECUTIVE READINGS 14 VCC = 5V = 5V V 12 REF VIN = 0V = 2.5V V 10 INCM FO = GND TA = 25°C 8 6 4 2 0 –2.0 –1.5 0 0.5 –0.5 OUTPUT CODE (ppm OF VREF) –1.0 1 2411 G07 ADC READING (ppm OF VREF) NUMBER OF READINGS (%) 0 RMS NOISE (ppm OF VREF) GAUSSIAN DISTRIBUTION m = –0.647ppm σ = 0.287ppm 6 UW 2411 G01 Total Unadjusted Error (VCC = 5V, VREF = 2.5V) 10 8 6 4 2 0 –2 –4 –6 –8 Total Unadjusted Error (VCC = 2.7V, VREF = 2.5V) TA = 90°C –0.5 –1.0 VCC = 5V REF + = 2.5V REF – = GND VINCM = 2.5V FO = GND –0.75 TA = 90°C VCC = 2.7V REF + = 2.5V REF – = GND VINCM = 1.25V FO = GND – 0.75 TA = – 45°C TA = 25°C 0.75 1.25 2411 G03 –1.5 –1.25 –0.25 0.25 VIN (V) 0.75 1.25 2411 G02 –10 –1.25 0.25 – 0.25 VIN (V) Integral Nonlinearity (VCC = 5V, VREF = 2.5V) 10 8 Integral Nonlinearity (VCC = 2.7V, VREF = 2.5V) TA = 25°C 6 TA = 25°C 4 2 0 –2 –4 –6 –8 VCC = 2.7V REF + = 2.5V REF – = GND VINCM = 1.25V FO = GND – 0.75 TA = – 45°C TA = 25°C TA = 90°C –0.5 –1.0 VCC = 5V REF + = 2.5V REF – = GND VINCM = 2.5V FO = GND –0.75 TA = 90°C 1.5 2 2.5 –1.5 –1.25 –0.25 0.25 VIN (V) 0.75 1.25 2411 G01 –10 –1.25 0.25 – 0.25 VIN (V) 0.75 1.25 2411 G06 2411 G04 Long Term ADC Readings 1.0 0.5 VCC = 5V, VREF = 5V, VIN = 0V, VINCM = 2.5V, FO = GND, TA = 25°C, RMS NOISE = 0.29ppm 0.5 RMS Noise vs Input Differential Voltage 0.4 0.3 –0.5 –1.0 0.2 TA = 25°C VCC = 5V VREF = 5V VINCM = 2.5V FO = GND 2.5 –1.5 0.1 –2.0 0 5 10 15 20 25 30 35 40 45 50 55 60 TIME (HOURS) 2411 G08 0 –2.5 –2 –1.5 –1 –0.5 0 0.5 1 1.5 2 INPUT DIFFERENTIAL VOLTAGE (V) 2411 G09 LTC2411/LTC2411-1 TYPICAL PERFOR A CE CHARACTERISTICS RMS Noise vs VINCM 1.60 1.55 RMS NOISE (µV) VCC = 5V VIN = 0V REF + = 5V FO = GND – = GND REF TA = 25°C RMS NOISE (µV) 1.45 1.40 1.35 1.30 –1 0 1 3 2 VINCM (V) 4 5 6 1.45 1.40 1.35 1.30 –45 –30 –15 RMS NOISE (µV) 1.50 RMS Noise vs VREF 1.60 1.55 VCC = 5V REF – = GND VIN = 0V FO = GND TA = 25°C OFFSET ERROR (ppm OF VREF) OFFSET ERROR (ppm OF VREF) RMS NOISE (µV) 1.50 1.45 1.40 1.35 1.30 0 1 2 3 VREF (V) Offset Error vs VCC 0 0.8 OFFSET ERROR (ppm OF VREF) 0.6 0.4 0.2 0 –0.2 –0.4 –0.6 –0.8 –1.0 2.7 3.1 3.5 3.9 4.3 VCC (V) 4.7 5.1 5.5 REF + = 2.5V REF – = GND VIN = 0V VINCM = GND FO = GND TA = 25°C 0 0.8 OFFSET ERROR (ppm OF VREF) 0.6 0.4 0.2 0 –0.2 –0.4 –0.6 –0.8 –1.0 FULL-SCALE ERROR (ppm OF VREF) UW 2411 G10 RMS Noise vs Temperature 1.60 1.55 1.50 VCC = 5V VREF = 5V VIN = 0V VINCM = GND FO = GND 1.60 1.55 1.50 1.45 1.40 1.35 RMS Noise vs VCC REF + = 2.5V REF – = GND VIN = 0V FO = GND TA = 25°C 0 15 30 45 60 TEMPERATURE (°C) 75 90 1.30 2.7 3.1 3.5 3.9 4.3 VCC (V) 4.7 5.1 5.5 2411 G11 2411 G12 Offset Error vs VINCM 0 –0.1 –0.2 –0.3 –0.4 –0.5 –0.6 –0.7 –0.8 –0.9 4 5 2411 G13 Offset Error vs Temperature 0 –0.1 –0.2 –0.3 –0.4 –0.5 –0.6 –0.7 –0.8 –0.9 VCC = 5V VREF = 5V VIN = 0V VINCM = GND FO = GND VCC = 5V REF + = 5V REF – = GND VIN = 0V FO = GND TA = 25°C –1.0 –1 0 1 2 3 VINCM (V) 4 5 6 –1.0 –45 –30 –15 0 15 30 45 60 TEMPERATURE (°C) 75 90 2411 G14 2411 G15 Offset Error vs VREF VCC = 5V REF– = GND VIN = 0V VINCM = GND FO = GND TA = 25°C 3 2 1 0 –1 –2 + Full-Scale Error vs Temperature VCC = 5V REF + = 5V REF – = GND IN+ = 2.5V IN – = GND FO = GND 0 1 3 2 VREF (V) 4 5 2411 G17 –3 –45 –30 –15 0 15 30 45 60 TEMPERATURE (°C) 75 90 2411 G16 2411 G18 7 LTC2411/LTC2411-1 TYPICAL PERFOR A CE CHARACTERISTICS + Full-Scale Error vs Temperature 5 –FULL-SCALE ERROR (ppm OF VREF) FULL-SCALE ERROR (ppm OF VREF) 3 2 1 0 –1 –2 –3 –4 2 1 0 VCC = 5V REF + = 5V REF – = GND + –2 IN – = GND IN = 2.5V FO = GND –3 –45 –30 –15 0 15 30 45 60 TEMPERATURE (°C) –1 –FULL-SCALE ERROR (ppm OF VREF) 4 VCC = 2.7V REF + = 2.5V REF – = GND IN + = 1.25V IN – = GND FO = GND –5 –45 –30 –15 0 15 30 45 60 TEMPERATURE (°C) PSRR vs Frequency at VCC (LTC2411) 0 –20 –40 REJECTION (dB) –60 –80 –100 –120 –140 1 REJECTION (dB) REJECTION (dB) VCC = 4.1V DC REF + = 2.5V REF – = GND IN + = GND IN – = GND FO = GND TA = 25°C 10 10k 100k 1k 100 FREQUENCY AT VCC (Hz) PSRR vs Frequency at VCC (LTC2411-1) VCC = 4.1V DC REF + = 2.5V –20 REF – = GND IN + = GND –40 IN – = GND FO = GND –60 TA = 25°C –80 –100 –120 –140 1 10 10k 100k 1k 100 FREQUECY AT VCC (Hz) 1M 0 0 REJECTION (dB) REJECTION (dB) –80 –100 –120 –140 0 20 40 60 80 100 120 140 160 180 200 220 FREQUENCY AT VCC (Hz) 2411 G32 REJECTION (dB) 8 UW 75 2411 G19 2411 G22 –Full-Scale Error vs Temperature 3 5 4 3 2 1 0 –1 –2 –3 –4 –Full-Scale Error vs Temperature 90 75 90 –5 –45 –30 –15 VCC = 2.7V REF + = 2.5V REF – = GND IN + = GND IN – = 1.25V FO = GND 0 15 30 45 60 TEMPERATURE (°C) 75 90 2411 G20 2411 G21 PSRR vs Frequency at VCC (LTC2411) 0 –20 –40 –60 –80 VCC = 4.1V DC ±1.4V REF + = 2.5V REF – = GND IN + = GND IN – = GND FO = GND TA = 25°C 0 –20 –40 –60 –80 –100 –120 PSRR vs Frequency at VCC (LTC2411) VCC = 4.1V DC ±0.7VP-P REF + = 2.5V REF – = GND IN + = GND IN – = GND FO = GND TA = 25°C –100 –120 –140 1M 0 30 60 90 120 150 180 210 240 FREQUENCY AT VCC (Hz) 2411 G23 –140 7600 7650 7700 7750 FREQUENCY AT VCC (Hz) 7800 2411 G24 PSRR vs Frequency at VCC (LTC2411-1) VCC = 4.1V DC ± 1.4V REF + = 2.5V –20 REF – = GND IN + = GND –40 IN – = GND FO = GND –60 TA = 25°C 0 PSRR vs Frequency at VCC (LTC2411-1) VCC = 4.1V DC ± 0.7V REF + = 2.5V –20 REF – = GND IN + = GND –40 IN – = GND FO = GND –60 TA = 25°C –80 –100 –120 –140 6880 6930 6980 7030 FREQUENCY AT VCC (Hz) 7080 2411 G33 2411 G31 LTC2411/LTC2411-1 TYPICAL PERFOR A CE CHARACTERISTICS Conversion Current vs Temperature FO = GND 230 CS = GND SCK = NC 220 SDO = NC 210 200 190 180 170 160 –45 –30 –15 VCC = 3V VCC = 2.7V 0 15 30 45 60 TEMPERATURE (°C) 75 90 VCC = 5V 240 CONVERSION CURRENT (µA) SLEEP MODE CURRENT (µA) SUPPLY CURRENT (µA) VCC = 5.5V Offset Error vs Output Data Rate 40 20 VREF = 2.5V OFFSET ERROR (ppm OF VREF) 0 RESOLUTION (BITS) RESOLUTION (BITS) –20 VREF = 5V –40 –60 –80 VCC = 5V REF – = GND VINCM = 2.5V VIN = 0V FO = EXT OSC TA = 25°C 0 10 20 30 40 50 60 70 80 90 100 OUTPUT DATA RATE (READINGS/SEC) 2411 G28 –100 –120 PI FU CTIO S VCC (Pin 1): Positive Supply Voltage. Bypass to GND (Pin 6) with a 10µF tantalum capacitor in parallel with 0.1µF ceramic capacitor as close to the part as possible. REF + (Pin 2), REF – (Pin 3): Differential Reference Input. The voltage on these pins can have any value between GND and VCC as long as the reference positive input, REF +, is more positive than the reference negative input, REF –, by at least 0.1V. IN + (Pin 4), IN– (Pin 5): Differential Analog Input. The voltage on these pins can have any value between GND – 0.3V and VCC + 0.3V. Within these limits, the converter bipolar input range (VIN = IN+ – IN–) extends from – 0.5 • (VREF ) to 0.5 • (VREF ). Outside this input range, the converter produces unique overrange and underrange output codes. GND (Pin 6): Ground. Connect this pin to a ground plane through a low impedance connection. CS (Pin 7): Active LOW Digital Input. A LOW on this pin enables the SDO digital output and wakes up the ADC. Following each conversion the ADC automatically enters UW 2411 G25 Conversion Current vs Output Data Rate 650 REF + = VCC 600 REF – = GND IN + = GND 550 IN – = GND 500 TA = 25°C SCK = NC 450 SDO = NC CS = GND 400 FO = EXT OSC 350 300 250 200 150 0 10 20 30 40 50 60 70 80 90 100 OUTPUT DATA RATE (READINGS/SEC) 2411 G26 Sleep Mode Current vs Temperature 5 FO = GND CS = VCC SCK = NC SDO = NC VCC = 5.5V VCC = 5V 4 3 2 VCC = 5V VCC = 3V VCC = 2.7V VCC = 3V 1 0 –45 –30 –15 0 15 30 45 60 TEMPERATURE (°C) 75 90 2411 G27 Resolution (NOISERMS ≤ 1LSB) vs Output Data Rate 22 VREF = 5V 21 VREF = 2.5V 20 VCC = 5V REF – = GND VINCM = 2.5V 19 VIN = 0V FO = EXT OSC RES = LOG2(VREF/NOISERMS) TA = 25°C 18 0 10 20 30 40 50 60 70 80 90 100 OUTPUT DATA RATE (READINGS/SEC) 2411 G29 Resolution (INLMAX ≤ 1LSB) vs Output Data Rate 22 20 18 VREF = 5V 16 14 12 10 0 10 20 30 40 50 60 70 80 90 100 OUTPUT DATA RATE (READINGS/SEC) 2411 G30 VCC = 5V VREF = 2.5V REF – = GND VINCM = 2.5V VIN = 0V FO = EXT OSC RES = LOG2(VREF/INLMAX) TA = 25°C U U U 9 LTC2411/LTC2411-1 PI FU CTIO S the Sleep mode and remains in this low power state as long as CS is HIGH. A LOW-to-HIGH transition on CS during the Data Output transfer aborts the data transfer and starts a new conversion. SDO (Pin 8): Three-State Digital Output. During the Data Output period, this pin is used as the serial data output. When the chip select CS is HIGH (CS = VCC), the SDO pin is in a high impedance state. During the Conversion and Sleep periods, this pin is used as the conversion status output. The conversion status can be observed by pulling CS LOW. SCK (Pin 9): Bidirectional Digital Clock Pin. In Internal Serial Clock Operation mode, SCK is used as the digital output for the internal serial interface clock during the Data Output period. In External Serial Clock Operation mode, SCK is used as the digital input for the external serial interface clock during the Data Output period. A weak internal pull-up is automatically activated in Internal Serial Clock Operation mode. The Serial Clock Operation mode is determined by the logic level applied to the SCK pin at power up or during the most recent falling edge of CS. FO (Pin 10): Frequency Control Pin. Digital input that controls the ADC’s notch frequencies and conversion time. For the LTC2411, when the FO pin is connected to VCC (FO = VCC), the converter uses its internal oscillator and the digital filter first null is located at 50Hz. When the FO pin is connected to GND (FO = OV), the converter uses its internal oscillator and the digital filter first null is located at 60Hz. For the LTC2411-1, the converter provides simultaneous 50Hz/60Hz rejection with the FO pin connected to GND. When FO is driven by an external clock signal with a frequency fEOSC, the converters use this signal as their system clock and the digital filter first null is located at a frequency fEOSC/2560. FU CTIO AL BLOCK DIAGRA VCC GND IN + IN – + –∫ ∫ ∫ ∑ ADC SERIAL INTERFACE DECIMATING FIR REF REF – + –+ DAC TEST CIRCUITS SDO 1.69k CLOAD = 20pF 2411 TA03 Hi-Z TO VOH VOL TO VOH VOH TO Hi-Z 10 W U U U U U INTERNAL OSCILLATOR AUTOCALIBRATION AND CONTROL FO (INT/EXT) SDO SCK CS 2411 FD Figure 1 VCC 1.69k SDO CLOAD = 20pF 2411 TA04 Hi-Z TO VOL VOH TO VOL VOL TO Hi-Z LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO CONVERTER OPERATION Converter Operation Cycle The LTC2411/LTC2411-1 are low power, delta-sigma analog-to-digital converters with an easy-to-use 3-wire serial interface (see Figure 1). Their operation is made up of three states. The converter operating cycle begins with the conversion, followed by the low power sleep state and ends with the data output (see Figure 2). The 3-wire interface consists of serial data output (SDO), serial clock (SCK) and chip select (CS). Initially, the LTC2411/LTC2411-1 perform a conversion. Once the conversion is complete, the devices enter the sleep state. While in this sleep state, power consumption is reduced by an order of magnitude. The parts remain in the sleep state as long as CS is HIGH. The conversion result is held indefinitely in a static shift register while the converter is in the sleep state. Once CS is pulled LOW, the devices begin outputting the conversion result. There is no latency in the conversion result. The data output corresponds to the conversion just performed. This result is shifted out on the serial data out pin (SDO) under the control of the serial clock (SCK). Data is updated on the falling edge of SCK allowing the user to reliably latch data on the rising edge of SCK (see Figure 3). The data output state is concluded once 32 bits are read out of the ADC or when CS is brought HIGH. The devices automatically initiate a new conversion and the cycle repeats. CONVERT SLEEP FALSE CS = LOW AND SCK TRUE DATA OUTPUT 2411 F02 Figure 2. LTC2411/LTC2411-1 State Transition Diagram U Through timing control of the CS and SCK pins, the LTC2411/LTC2411-1 offer several flexible modes of operation (internal or external SCK and free-running conversion modes). These various modes do not require programming configuration registers; moreover, they do not disturb the cyclic operation described above. These modes of operation are described in detail in the Serial Interface Timing Modes section. Conversion Clock A major advantage the delta-sigma converter offers over conventional type converters is an on-chip digital filter (commonly implemented as a Sinc or Comb filter). For high resolution, low frequency applications, this filter is typically designed to reject line frequencies of 50 or 60Hz plus their harmonics. The filter rejection performance is directly related to the accuracy of the converter system clock. The LTC2411/LTC2411-1 incorporate a highly accurate on-chip oscillator. This eliminates the need for external frequency setting components such as crystals or oscillators. Clocked by the on-chip oscillator, the LTC2411 achieves a minimum of 110dB rejection at the line frequency (50Hz or 60Hz ± 2%) and the LTC2411-1 achieves a minimum of 87dB rejection over 49Hz to 61.2Hz. Ease of Use The LTC2411/LTC2411-1 data output has no latency, filter settling delay or redundant data associated with the conversion cycle. There is a one-to-one correspondence between the conversion and the output data. Therefore, multiplexing multiple analog voltages is easy. The LTC2411/LTC2411-1 perform offset and full-scale calibrations in every conversion cycle. This calibration is transparent to the user and has no effect on the cyclic operation described above. The advantage of continuous calibration is extreme stability of offset and full-scale readings with respect to time, supply voltage change and temperature drift. Power-Up Sequence The LTC2411/LTC2411-1 automatically enter an internal reset state when the power supply voltage VCC drops below approximately 1.9V. This feature guarantees the W UU 11 LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO integrity of the conversion result and of the serial interface mode selection. (See the 2-wire I/O sections in the Serial Interface Timing Modes section.) When the VCC voltage rises above this critical threshold, the converter creates an internal power-on-reset (POR) signal with a duration of approximately 1ms. The POR signal clears all internal registers. Following the POR signal, the LTC2411/LTC2411-1 start a normal conversion cycle and follow the succession of states described above. The first conversion result following POR is accurate within the specifications of the device if the power supply voltage is restored within the operating range (2.7V to 5.5V) before the end of the POR time interval. Reference Voltage Range The LTC2411/LTC2411-1 accept a truly differential external reference voltage. The absolute/common mode voltage specification for the REF + and REF – pins covers the entire range from GND to VCC. For correct converter operation, the REF + pin must always be more positive than the REF – pin. The LTC2411/LTC2411-1 can accept a differential reference voltage from 0.1V to VCC. The converter output noise is determined by the thermal noise of the front-end circuits, and, as such, its value in nanovolts is nearly constant with reference voltage. A decrease in reference voltage will not significantly improve the converter’s effective resolution. On the other hand, a reduced reference voltage will improve the converter’s overall INL performance. A reduced reference voltage will also improve the converter performance when operated with an external conversion clock (external FO signal) at substantially higher output data rates. Input Voltage Range The analog input is truly differential with an absolute/ common mode range for the IN+ and IN– input pins extending from GND – 0.3V to VCC + 0.3V. Outside these limits, the ESD protection devices begin to turn on and the errors due to input leakage current increase rapidly. Within these limits, the LTC2411/LTC2411-1 convert the bipolar differential input signal, VIN = IN + – IN –, from – FS = – 0.5 • VREF to +FS = 0.5 • VREF where VREF = REF+ – REF –. Outside this range the converter indicates 12 U the overrange or the underrange condition using distinct output codes. Input signals applied to IN+ and IN– pins may extend by 300mV below ground and above VCC. In order to limit any fault current, resistors of up to 5k may be added in series with the IN+ and IN– pins without affecting the performance of the device. In the physical layout, it is important to maintain the parasitic capacitance of the connection between these series resistors and the corresponding pins as low as possible; therefore, the resistors should be located as close as practical to the pins. In addition, series resistors will introduce a temperature dependent offset error due to the input leakage current. A 1nA input leakage current will develop a 1ppm offset error on a 5k resistor if VREF = 5V. This error has a very strong temperature dependency. Output Data Format The LTC2411/LTC2411-1 serial output data stream is 32 bits long. The first 3 bits represent status information indicating the sign and conversion state. The next 24 bits are the conversion result, MSB first. The remaining 5 bits are sub LSBs beyond the 24-bit level that may be included in averaging or discarded without loss of resolution. The third and fourth bits together are also used to indicate an underrange condition (the differential input voltage is below – FS) or an overrange condition (the differential input voltage is above + FS). Bit 31 (first output bit) is the end of conversion (EOC) indicator. This bit is available at the SDO pin during the conversion and sleep states whenever the CS pin is LOW. This bit is HIGH during the conversion and goes LOW when the conversion is complete. Bit 30 (second output bit) is a dummy bit (DMY) and is always LOW. Bit 29 (third output bit) is the conversion result sign indicator (SIG). If VIN is >0, this bit is HIGH. If VIN is 0.01µF) may be required in certain configurations for antialiasing or general input signal filtering. Such capacitors will average the input sampling charge and the external source resistance will see a quasi constant input differential impedance. For the LTC2411, when FO = LOW (internal oscillator and 60Hz notch), the typical differential input resistance is 5.4MΩ which will generate a gain error of approximately 0.093ppm for each ohm of source resistance driving IN+ or IN –. When FO = HIGH (internal oscillator and 50Hz notch), the typical differential input resistance is 6.5MΩ which will generate a gain error of approximately 0.077ppm for each ohm of source resistance driving IN+ or IN –. For the LTC2411-1, the typical differential input resistance is 6MΩ which will generate a gain error of approximately 0.084ppm for each ohm of source resistance driving IN+ or IN– (FO = LOW). When FO is driven by an external oscillator with a frequency fEOSC (external conversion clock operation), the typical differential input resistance is 0.83 • 1012/fEOSCΩ and each ohm of source resistance driving IN+ or IN – will result in 0.59 • 10–6 • fEOSCppm gain error. The effect of the source resistance on the two input pins is additive with respect to this gain error. The typical +FS and –FS errors as a function of the sum of the source resistance seen by IN+ and IN– for large values of CIN are shown in Figure 15. In addition to this gain error, an offset error term may also appear. The offset error is proportional with the mismatch between the source impedance driving the two input pins IN+ and IN– and with the difference between the input and 120 100 VCC = 5V REF + = 5V REF – = GND IN + = 3.75V IN – = 1.25V FO = GND TA = 25°C +FS ERROR (ppm OF VREF) 80 60 40 20 –FS ERROR (ppm OF VREF) CIN = 10µF CIN = 1µF CIN = 0.1µF CIN = 0.01µF 0 0 100 200 300 400 500 600 700 800 900 1000 RSOURCE (Ω) 2411 F15a Figure 15a. + FS Error vs RSOURCE at IN + or IN – (Large CIN) 24 U reference common mode voltages. While the input drive circuit nonzero source impedance combined with the converter average input current will not degrade the INL performance, indirect distortion may result from the modulation of the offset error by the common mode component of the input signal. Thus, when using large CIN capacitor values, it is advisable to carefully match the source impedance seen by the IN+ and IN– pins. For the LTC2411, when FO = LOW (internal oscillator and 60Hz notch), every 1Ω mismatch in source impedance transforms a full-scale common mode input signal into a differential mode input signal of 0.093ppm. When FO = HIGH (internal oscillator and 50Hz notch), every 1Ω mismatch in source impedance transforms a full-scale common mode input signal into a differential mode input signal of 0.077ppm. For the LTC2411-1, when internal oscillator is used (FO = LOW), every 1Ω mismatch in source impedance transforms a full-scale common mode input signal into a differential mode input signal of 0.084ppm. When FO is driven by an external oscillator with a frequency fEOSC, every 1Ω mismatch in source impedance transforms a full-scale common mode input signal into a differential mode input signal of 0.59 • 10–6 • fEOSCppm. Figure 16 shows the typical offset error due to input common mode voltage for various values of source resistance imbalance between the IN+ and IN– pins when large CIN values are used. If possible, it is desirable to operate with the input signal common mode voltage very close to the reference signal common mode voltage as is the case in the ratiometric measurement of a symmetric bridge. This configuration 0 CIN = 0.01µF –20 –40 CIN = 0.1µF –60 –80 –100 –120 VCC = 5V REF + = 5V REF – = GND IN + = 1.25V IN – = 3.75V FO = GND TA = 25°C CIN = 10µF CIN = 1µF 0 100 200 300 400 500 600 700 800 900 1000 RSOURCE (Ω) 2411 F15b W UU Figure 15b. – FS Error vs RSOURCE at IN + or IN – (Large CIN) LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO 50 40 OFFSET ERROR (ppm OF VREF) 30 20 10 0 –10 –20 –30 –40 –50 0 0.5 1 G FO = GND TA = 25°C RSOURCEIN – = 500Ω CIN = 10µF 1.5 2 2.5 3 VINCM (V) 3.5 4 4.5 5 B C D E F A VCC = 5V REF + = 5V REF – = GND IN + = IN – = VINCM A: ∆RIN = + 400Ω B: ∆RIN = + 200Ω C: ∆RIN = + 100Ω D: ∆RIN = 0Ω E: ∆RIN = – 100Ω F: ∆RIN = – 200Ω G: ∆RIN = – 400Ω 2411 F16 Figure 16. Offset Error vs Common Mode Voltage (VINCM = IN+ = IN–) and Input Source Resistance Imbalance (∆RIN = RSOURCEIN+ – RSOURCEIN–) for Large CIN Values (CIN ≥ 1µF) eliminates the offset error caused by mismatched source impedances. The magnitude of the dynamic input current depends upon the size of the very stable internal sampling capacitors and upon the accuracy of the converter sampling clock. The accuracy of the internal clock over the entire temperature and power supply range is typically better than 1%. Such a specification can also be easily achieved by an external clock. When relatively stable resistors (50ppm/°C) are used for the external source impedance seen by IN+ and IN–, the expected drift of the dynamic current, offset and gain errors will be insignificant (about 1% of their respective values over the entire temperature and voltage range). Even for the most stringent applications, a one-time calibration operation may be sufficient. In addition to the input sampling charge, the input ESD protection diodes have a temperature dependent leakage current. This current, nominally 1nA (±10nA max), results in a small offset shift. A 100Ω source resistance will create a 0.1µV typical and 1µV maximum offset voltage. Reference Current In a similar fashion, the LTC2411/LTC2411-1 sample the differential reference pins REF+ and REF– transfering small amount of charge to and from the external driving circuits thus producing a dynamic reference current. This current U does not change the converter offset, but it may degrade the gain and INL performance. The effect of this current can be analyzed in the same two distinct situations. For relatively small values of the external reference capacitors (CREF < 0.01µF), the voltage on the sampling capacitor settles almost completely and relatively large values for the source impedance result in only small errors. Such values for CREF will deteriorate the converter offset and gain performance without significant benefits of reference filtering and the user is advised to avoid them. Larger values of reference capacitors (CREF > 0.01µF) may be required as reference filters in certain configurations. Such capacitors will average the reference sampling charge and the external source resistance will see a quasi constant reference differential impedance. For the LTC2411, when FO = LOW (internal oscillator and 60Hz notch), the typical differential reference resistance is 3.9MΩ which will generate a gain error of approximately 0.13ppm for each ohm of source resistance driving REF+ or REF–. When FO = HIGH (internal oscillator and 50Hz notch), the typical differential reference resistance is 4.68MΩ which will generate a gain error of approximately 0.11ppm for each ohm of source resistance driving REF+ or REF –. For the LTC2411-1, when internal oscillator is used (FO = LOW), the typical differential reference resistance is 4.29MΩ which will generate a gain error of approximately 0.12ppm for each ohm of source resistance driving REF + or REF –. When FO is driven by an external oscillator with a frequency fEOSC (external conversion clock operation), the typical differential reference resistance is 0.60 • 1012/fEOSCΩ and each ohm of source resistance drving REF+ or REF– will result in 0.823 • 10–6 • fEOSCppm gain error. The effect of the source resistance on the two reference pins is additive with respect to this gain error. The typical FS errors for various combinations of source resistance seen by the REF+ and REF– pins and external capacitance CREF connected to these pins are shown in Figures 17 and 18. Typical – FS errors are similar to + FS errors with opposite polarity. In addition to this gain error, the converter INL performance is degraded by the reference source impedance. For LTC2411, when FO = LOW (internal oscillator and 60Hz notch), every 100Ω of source resistance driving REF+ or W UU 25 LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO 0 VCC = 5V REF + = 5V REF – = GND IN + = 3.75V IN – = 1.25V FO = GND TA = 25°C CREF = 0pF –30 CREF = 100pF –40 CREF = 0.001µF CREF = 0.01µF –50 1 10 100 1k RSOURCE (Ω) 10k 100k 2411 F17a +FS ERROR (ppm OF VREF) –20 –FS ERROR (ppm OF VREF) –10 Figure 17a. +FS Error vs RSOURCE at REF+ or REF– (Small CIN) 0 –20 +FS ERROR (ppm OF VREF) –40 –60 –80 VCC = 5V REF + = 5V REF – = GND IN + = 3.75V IN – = 1.25V FO = GND TA = 25°C 0 100 200 300 400 500 600 700 800 900 1000 RSOURCE (Ω) 2411 F18a CREF = 0.01µF CREF = 10µF CREF = 1µF – FS ERROR (ppm OF VREF) CREF = 0.1µF –100 –120 –140 –160 Figure 18a. +FS Error vs RSOURCE at REF+ or REF– (Large CIN) REF– translates into about 0.45ppm additional INL error. When FO = HIGH (internal oscillator and 50Hz notch), every 100Ω of source resistance driving REF+ or REF– translates into about 0.37ppm additional INL error. For the LTC2411-1, when FO = LOW, every 100Ω of source resistance driving REF+ or REF– translates into about 0.41ppm additional INL error. When FO is driven by an external oscillator with a frequency fEOSC, every 100Ω of source resistance driving REF+ or REF– translates into about 2.91 • 10–6 • fEOSCppm additional INL error. Figure 19 shows the typical INL error due to the source resistance driving the REF+ or REF– pins when large CREF values are used. The effect of the source resistance on the two reference pins is additive with respect to this INL error. In general, matching of source impedance for the REF+ and REF– pins does not help the gain or the INL error. The INL (ppm OF VREF) 26 U 50 VCC = 5V REF + = 5V REF – = GND 40 IN + = 1.25V IN – = 3.75V FO = GND 30 TA = 25°C CREF = 0pF 20 CREF = 100pF 10 CREF = 0.001µF CREF = 0.01µF 0 1 10 100 1k RSOURCE (Ω) 10k 100k 2411 F17b W UU Figure 17b. – FS Error vs RSOURCE at REF+ or REF– (Small CIN) 160 VCC = 5V + 140 REF – = 5V REF = GND + 120 IN – = 1.25V IN = 3.75V 100 FO = GND TA = 25°C 80 60 40 20 0 0 100 200 300 400 500 600 700 800 900 1000 RSOURCE (Ω) 2411 F18b CREF = 10µF CREF = 1µF CREF = 0.1µF CREF = 0.01µF Figure 18b. – FS Error vs RSOURCE at REF+ or REF– (Large CIN) 10 9 6 4 2 0 –2 –4 –6 –8 –10 –0.5–0.4– 0.3– 0.2– 0.1 0 0.1 0.2 0.3 0.4 0.5 VINDIF/VREFDIF VCC = 5V FO = GND REF + = 5V CREF = 10µF REF – = GND TA = 25°C 2411 F19 VINCM = 0.5 • (IN + + IN –) = 2.5V RSOURCE = 500Ω RSOURCE = 2k RSOURCE = 1k Figure 19. INL vs Differential Input Voltage (VIN = IN+ = IN–) and Reference Source Resistance (RSOURCE at REF + and REF –) for Large CREF Values (CREF ≥ 1µF) LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO user is thus advised to minimize the combined source impedance driving the REF+ and REF– pins rather than to try to match it. The magnitude of the dynamic reference current depends upon the size of the very stable internal sampling capacitors and upon the accuracy of the converter sampling clock. The accuracy of the internal clock over the entire temperature and power supply range is typical better than 1%. Such a specification can also be easily achieved by an external clock. When relatively stable resistors (50ppm/°C) are used for the external source impedance seen by REF+ and REF–, the expected drift of the dynamic current gain error will be insignificant (about 1% of its value over the entire temperature and voltage range). Even for the most stringent applications, a one-time calibration operation may be sufficient. In addition to the reference sampling charge, the reference pins ESD protection diodes have a temperature dependent leakage current. This leakage current, nominally 1nA (±10nA max), results in a small gain error. A 100Ω source resistance will create a 0.05µV typical and 0.5µV maximum full-scale error. Output Data Rate When using its internal oscillator, the LTC2411 can produce up to 7.5 readings per second with a notch frequency of 60Hz (FO = LOW) and 6.25 readings per second with a notch frequency of 50Hz (FO = HIGH) and the LTC2411-1 can produce up to 6.8 readings per second with FO = LOW. The actual output data rate will depend upon the length of the sleep and data output phases which are controlled by the user and which can be made insignificantly short. When operated with an external conversion clock (FO connected to an external oscillator), the LTC2411/LTC24111 output data rate can be increased as desired. The duration of the conversion phase is 20510/fEOSC. If fEOSC = 153600Hz, the converter behaves as if the internal oscillator is used and the notch is set at 60Hz. There is no significant difference in the LTC2411/LTC2411-1 performance between these two operation modes. An increase in fEOSC over the nominal 153600Hz will translate into a proportional increase in the maximum output data rate. This substantial advantage is nevertheless U accompanied by three potential effects, which must be carefully considered. First, a change in fEOSC will result in a proportional change in the internal notch position and in a reduction of the converter differential mode rejection at the power line frequency. In many applications, the subsequent performance degradation can be substantially reduced by relying upon the LTC2411/LTC2411-1’s exceptional common mode rejection and by carefully eliminating common mode to differential mode conversion sources in the input circuit. The user should avoid single-ended input filters and should maintain a very high degree of matching and symmetry in the circuits driving the IN+ and IN– pins. Second, the increase in clock frequency will increase proportionally the amount of sampling charge transferred through the input and the reference pins. If large external input and/or reference capacitors (CIN, CREF) are used, the previous section provides formulae for evaluating the effect of the source resistance upon the converter performance for any value of fEOSC. If small external input and/ or reference capacitors (CIN, CREF) are used, the effect of the external source resistance upon the LTC2411/ LTC2411-1 typical performance can be inferred from Figures 13, 14 and 17 in which the horizontal axis is scaled by 153600/fEOSC. Third, an increase in the frequency of the external oscillator above 460800Hz (a more than 3× increase in the output data rate) will start to decrease the effectiveness of the internal autocalibration circuits. This will result in a progressive degradation in the converter accuracy and linearity. Typical measured performance curves for output data rates up to 100 readings per second are shown in Figures 20 to 27. In order to obtain the highest possible level of accuracy from this converter at output data rates above 20 readings per second, the user is advised to maximize the power supply voltage used and to limit the maximum ambient operating temperature. In certain circumstances, a reduction of the differential reference voltage may be beneficial. Input Bandwidth The combined effect of the internal sinc4 digital filter and of the analog and digital autocalibration circuits determines the LTC2411/LTC2411-1 input bandwidth. When W UU 27 LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO 120 80 40 0 –40 TA = 25°C –80 OFFSET ERROR (ppm OF VREF) +FS ERROR (ppm OF VREF) VCC = 5V VREF = 5V VINCM = 2.5V VIN = 0V FO = EXT OSC TA = 85°C –120 0 10 20 30 40 50 60 70 80 90 100 OUTPUT DATA RATE (READINGS/SEC) 2411 F20 Figure 20. Offset Error vs Output Data Rate and Temperature 100 TA = 25°C –FS ERROR (ppm OF VREF) 50 RESOLUTION (BITS) 0 TA = 85°C –50 VCC = 5V VREF = 5V –100 IN + = 1.25V IN – = 3.75V FO = EXT OSC –150 0 10 20 30 40 50 60 70 80 90 100 OUTPUT DATA RATE (READINGS/SEC) 2411 F22 Figure 22. – FS Error vs Output Data Rate and Temperature 22 OFFSET ERROR (ppm OF VREF) 20 RESOLUTION (BITS) 18 TA = 85°C 16 14 12 10 0 VCC = 5V VREF = 5V VINCM = 2.5V VIN = 0V FO = EXT OSC RES = LOG2(VREF/INLMAX) 10 20 30 40 50 60 70 80 90 100 OUTPUT DATA RATE (READINGS/SEC) 2411 F24 TA = 25°C Figure 24. Resolution (INLRMS ≤ 1LSB) vs Output Data Rate and Temperature 28 U 250 200 150 100 TA = 85°C 50 0 –50 0 10 20 30 40 50 60 70 80 90 100 OUTPUT DATA RATE (READINGS/SEC) 2411 F21 W UU VCC = 5V VREF = 5V IN + = 3.75V IN – = 1.25V FO = EXT OSC TA = 25°C Figure 21. + FS Error vs Output Data Rate and Temperature 22 21 TA = 85°C 20 19 18 17 16 0 VCC = 5V VREF = 5V VINCM = 2.5V VIN = 0V FO = EXT OSC RES = LOG2(VREF/NOISERMS) 10 20 30 40 50 60 70 80 90 100 OUTPUT DATA RATE (READINGS/SEC) 2411 F23 TA = 25°C Figure 23. Resolution (NoiseRMS ≤ 1LSB) vs Output Data Rate and Temperature 40 20 0 –20 VREF = 5V –40 –60 –80 VCC = 5V REF – = GND VINCM = 2.5V VIN = 0V FO = EXT OSC TA = 25°C 0 10 20 30 40 50 60 70 80 90 100 OUTPUT DATA RATE (READINGS/SEC) 2411 F25 VREF = 2.5V –100 –120 Figure 25. Offset Error vs Output Data Rate and Reference Voltage LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO 22 VREF = 5V 21 RESOLUTION (BITS) RESOLUTION (BITS) VREF = 2.5V 20 VCC = 5V REF – = GND VINCM = 2.5V 19 VIN = 0V FO = EXT OSC RES = LOG2(VREF/NOISERMS) TA = 25°C 18 0 10 20 30 40 50 60 70 80 90 100 OUTPUT DATA RATE (READINGS/SEC) 2411 F26 Figure 26. Resolution (NoiseRMS ≤ 1LSB) vs Output Data Rate and Reference Voltage INPUT SIGNAL ATTENUATION (dB) the internal oscillator is used, the 3dB input bandwidth of the LTC2411 is 3.63Hz for 60Hz notch frequency (FO = LOW) and 3.02Hz for 50Hz notch frequency (FO = HIGH). The 3dB input bandwidth for the LTC2411-1 is 3.30Hz (FO = LOW). If an external conversion clock generator of frequency fEOSC is connected to the FO pin, the 3dB input bandwidth is 0.236 • 10–6 • fEOSC. Due to the complex filtering and calibration algorithms utilized, the converter input bandwidth is not modeled very accurately by a first order filter with the pole located at the 3dB frequency. When the internal oscillator is used, the shape of the LTC2411/LTC2411-1 input bandwidth is shown in Figure 28. When an external oscillator of frequency fEOSC is used, the shape of the LTC2411/LTC2411-1 input bandwidth can be derived from Figure 28, FO = LOW curve of the LTC2411 in which the horizontal axis is scaled by fEOSC/153600. The conversion noise (1.45µVRMS typical for VREF = 5V) can be modeled as a white noise source connected to a noise free converter. The noise spectral density is 70nV/√Hz for an infinite bandwidth source and 126nV/√Hz for a single 0.5MHz pole source. From these numbers, it is clear that particular attention must be given to the design of external amplification circuits. Such circuits face the simultaneous requirements of very low bandwidth (just a few Hz) in order to reduce the output referred noise and relatively high bandwidth (at least 500kHz) necessary to drive the input switched-capacitor network. A possible U 22 20 18 VREF = 2.5V 16 14 12 10 0 VCC = 5V REF – = GND VINCM = 2.5V VIN = 0V FO = EXT OSC RES = LOG2(VREF/INLMAX) TA = 25°C 10 20 30 40 50 60 70 80 90 100 OUTPUT DATA RATE (READINGS/SEC) 2411 F27 W UU VREF = 5V Figure 27. Resolution (INLMAX ≤ 1LSB) vs Output Data Rate and Reference Voltage 0.0 –0.5 –1.0 –1.5 –2.0 –2.5 –3.0 –3.5 –4.0 –4.5 –5.0 –5.5 –6.0 0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 DIFFERENTIAL INPUT SIGNAL FREQUENCY (Hz) 2411 F28 FO = HIGH (LTC2411) FO = LOW (LTC2411-1) FO = LOW (LTC2411) Figure 28. Input Signal Bandwidth Using the Internal Oscillator solution is a high gain, low bandwidth amplifier stage followed by a high bandwidth unity-gain buffer. When external amplifiers are driving the LTC2411/ LTC2411-1, the ADC input referred system noise calculation can be simplified by Figure 29. The noise of an amplifier driving the LTC2411/LTC2411-1 input pin can be modeled as a band-limited white noise source. Its bandwidth can be approximated by the bandwidth of a single pole lowpass filter with a corner frequency fi. The amplifier noise spectral density is ni. From Figure 29, using fi as the x-axis selector, we can find on the y-axis the noise equivalent bandwidth freqi of the input driving amplifier. This bandwidth includes the band limiting effects of the ADC 29 LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO internal calibration and filtering. The noise of the driving amplifier referred to the converter input and including all these effects can be calculated as N = ni • √freqi. The total system noise (referred to the LTC2411/LTC2411-1 input) can now be obtained by summing as square root of sum of squares the three ADC input referred noise sources: the LTC2411/LTC2411-1 internal noise (1.45µV), the noise of the IN + driving amplifier and the noise of the IN – driving amplifier. If the FO pin is driven by an external oscillator of frequency fEOSC, Figure 29 can still be used for noise calculation if the x-axis is scaled by fEOSC/153600. For large values of the ratio fEOSC/153600, the Figure 29 plot accuracy begins to decrease, but in the same time the LTC2411/LTC2411-1 1000 INPUT REFERRED NOISE EQUIVALENT BANDWIDTH (Hz) 100 10 FO = LOW FO = HIGH 1 0.1 0.1 1 10 100 1k 10k 100k INPUT NOISE SOURCE SINGLE POLE EQUIVALENT BANDWIDTH (Hz) 1M 2411 G29 Figure 29. Input Referred Noise Equivalent Bandwidth of an Input Connected White Noise Source 0 INPUT NORMAL MODE REJECTION (dB) –20 –30 –40 –50 –60 –70 –80 –90 –100 –110 –120 0 fS 2fS 3fS 4fS 5fS 6fS 7fS 8fS 9fS 10fS11fS12fS DIFFERENTIAL INPUT SIGNAL FREQUENCY (Hz) 2411 F30 INPUT NORMAL MODE REJECTION (dB) –10 FO = HIGH Figure 30. Input Normal Mode Rejection, Internal Oscillator and 50Hz Notch (LTC2411) 30 U noise floor rises and the noise contribution of the driving amplifiers lose significance. Normal Mode Rejection and Antialiasing One of the advantages delta-sigma ADCs offer over conventional ADCs is on-chip digital filtering. Combined with a large oversampling ratio, the LTC2411/LTC2411-1 significantly simplifies antialiasing filter requirements. The sinc4 digital filter provides greater than 120dB normal mode rejection at all frequencies except DC and integer multiples of the modulator sampling frequency (fS). The LTC2411/LTC2411-1’s autocalibration circuits further simplify the antialiasing requirements by additional normal mode signal filtering both in the analog and digital domain. Independent of the operating mode, fS = 256 • fN = 2048 • fOUTMAX where fN is the notch frequency and fOUTMAX is the maximum output data rate. In the internal oscillator mode, for the LTC2411, FS = 12800Hz with a 50Hz notch setting and fS = 15360Hz with a 60Hz notch setting. For the LTC2411-1, fS = 13980Hz (FO = LOW). In the external oscillator mode, fS = fEOSC/10. The combined normal mode rejection performance is shown in Figure 30 for the internal oscillator with 50Hz notch setting (FO = HIGH) and in Figure 31 for the internal oscillator with FO = LOW and for the external oscillator mode. The regions of low rejection occurring at integer multiples of fS have a very narrow bandwidth. Magnified details of the normal mode rejection curves are shown in 0 –10 –20 –30 –40 –50 –60 –70 –80 –90 –100 –110 –120 0 fS 2fS 3fS 4fS 5fS 6fS 7fS 8fS 9fS 10fS DIFFERENTIAL INPUT SIGNAL FREQUENCY (Hz) 2411 F31 W UU FO = LOW OR FO = EXTERNAL OSCILLATOR, fEOSC = 10 • fS Figure 31. Input Normal Mode Rejection, Internal Oscillator and FO = LOW or External Oscillator LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO 0 INPUT NORMAL MODE REJECTION (dB) INPUT NORMAL MODE REJECTION (dB) 0 fN 2fN 3fN 4fN 5fN 6fN 7fN INPUT SIGNAL FREQUENCY (Hz) 8fN –10 –20 –30 –40 –50 –60 –70 –80 –90 –100 –110 –120 2411 F32 Figure 32. Input Normal Mode Rejection NORMAL MODE REJECTION (dB) Figure 32 (rejection near DC) and Figure 33 (rejection at fS = 256fN) where fN represents the notch frequency. These curves have been derived for the external oscillator mode but they can be used in all operating modes by appropriately selecting the fN value. The user can expect to achieve in practice this level of performance using the internal oscillator as it is demonstrated by Figures 34 to 36. Typical measured values of the normal mode rejection of the LTC2411 operating with an internal oscillator and a 60Hz notch setting are shown in Figure 34 superimposed over the theoretical calculated curve. Similarly, typical measured values of the normal mode rejection of the LTC2411 operating with an internal oscillator and a 50Hz notch setting are shown in Figure 35 superimposed over the theoretical calculated curve. As a result of these remarkable normal mode specifications, minimal (if any) antialias filtering is required in front of the LTC2411/LTC2411-1. If passive RC components are placed in front of the LTC2411/LTC2411-1, the input dynamic current should be considered (see Input Current section). In cases where large effective RC time constants are used, an external buffer amplifier may be required to minimize the effects of dynamic input current. Traditional high order delta-sigma modulators, while providing very good linearity and resolution, suffer from potential instabilities at large input signal levels. The proprietary architecture used for the LTC2411/LTC2411-1 third order modulator resolves this problem and guarantees a predictable stable behavior at input signal levels of NORMAL MODE REJECTION (dB) U 0 –10 –20 –30 –40 –50 –60 –70 –80 –90 –100 –110 –120 250fN 252fN 254fN 256fN 258fN 260fN 262fN INPUT SIGNAL FREQUENCY (Hz) 2411 F33 W UU Figure 33. Input Normal Mode Rejection 0 –20 –40 – 60 –80 –100 –120 MEASURED DATA CALCULATED DATA VCC = 5V VREF = 5V VINCM = 2.5V VIN(P-P) = 5V FO = GND TA = 25°C 0 15 30 45 60 75 90 105 120 135 150 165 180 195 210 225 240 INPUT FREQUENCY (Hz) 2411 F34 Figure 34. Input Normal Mode Rejection vs Input Frequency (LTC2411) 0 –20 –40 – 60 –80 –100 –120 MEASURED DATA CALCULATED DATA VCC = 5V VREF = 5V VINCM = 2.5V VIN(P-P) = 5V FO = 5V TA = 25°C 0 25 50 75 100 125 INPUT FREQUENCY (Hz) 150 175 200 2411 F35 Figure 35. Input Normal Mode Rejection vs Input Frequency (LTC2411) 31 LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO 0 MEASURED DATA CALCULATED DATA NORMAL MODE REJECTION (dB) –20 –40 – 60 –80 –100 –120 NORMAL MODE REJECTION (dB) VCC = 5V VREF = 5V REF – = GND VINCM = 2.5V VIN(P-P) = 5V FO = GND TA = 25°C 0 20 40 60 80 100 120 140 160 INPUT FREQUENCY (Hz) 180 Figure 36. Input Normal Mode Rejection vs Input Frequency (LTC2411-1) NORMAL MODE REJECTION (dB) up to 150% of full scale. In many industrial applications, it is not uncommon to have to measure microvolt level signals superimposed over volt level perturbations and LTC2411/LTC2411-1 are eminently suited for such tasks. When the perturbation is differential, the specification of interest is the normal mode rejection for large input signal levels. With a reference voltage VREF = 5V, the LTC2411/ LTC2411-1 have a full-scale differential input range of 5V peak-to-peak. Figures 37 and 38 show measurement results for the LTC2411 normal mode rejection ratio with a 7.5V peak-to-peak (150% of full scale) input signal superimposed over the more traditional normal mode rejection ratio results obtained with a 5V peak-to-peak (full scale) input signal and Figure 39 shows the corresponding measurement result for the LTC2411-1. It is clear that the LTC2411/LTC2411-1 rejection performance is maintained with no compromises in this extreme situation. When operating with large input signal levels, the user must observe that such signals do not violate the device absolute maximum ratings. BRIDGE APPLICATIONS Typical strain gauge based bridges deliver only 2mV/Volt of excitation. As the maximum reference voltage of the LTC2411/LTC2411-1 is 5V, remote sensing of applied excitation without additional circuitry requires that excitation be limited to 5V. This gives only 10mV full scale, which can be resolved to 1 part in 5000 without averaging. For many solid state sensors, this is comparable to the sensor. NORMAL MODE REJECTION (dB) 32 U 0 –20 –40 – 60 –80 –100 –120 200 220 0 15 30 45 60 75 90 105 120 135 150 165 180 195 210 225 240 INPUT FREQUENCY (Hz) 2411 F37 W UU VIN(P-P) = 5V VIN(P-P) = 7.5V (150% OF FULL SCALE) VCC = 5V VREF = 5V VINCM = 2.5V FO = GND TA = 25°C 2411 F36 Figure 37. Measured Input Normal Mode Rejection vs Input Frequency (LTC2411) 0 –20 –40 – 60 –80 –100 –120 VIN(P-P) = 5V VIN(P-P) = 7.5V (150% OF FULL SCALE) VCC = 5V VREF = 5V VINCM = 2.5V FO = 5 V TA = 25°C 0 25 50 75 100 125 INPUT FREQUENCY (Hz) 150 175 200 2411 F38 Figure 38. Measured Input Normal Mode Rejection vs Input Frequency (LTC2411) 0 –20 –40 – 60 –80 –100 –120 VIN(P-P) = 5V VIN(P-P) = 7.5V (150% OF FULL SCALE) VCC = 5V VREF = 5V REF – = GND VINCM = 2.5V FO = GND TA = 25°C 0 20 40 60 80 100 120 140 160 INPUT FREQUENCY (Hz) 180 200 220 2411 F39 Figure 39. Measured Input Normal Mode Rejection vs Input Frequency (LTC2411-1) LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO Averaging 64 samples however reduces the noise level by a factor of eight, bringing the resolving power to 1 part in 40000, comparable to better weighing systems. Hysteresis and creep effects in the load cells are typically much greater than this. Most applications that require strain measurements to this level of accuracy are measuring slowly changing phenomena, hence the time required to average a large number of readings is usually not an issue. For those systems that require accurate measurement of a small incremental change on a significant tare weight, the lack of history effects in the LTC2400 family is of great benefit. For those applications that cannot be fulfilled by the LTC2411/LTC2411-1 alone, compensating for error in external amplification can be done effectively due to the “no latency” feature of the LTC2411/LTC2411-1. No latency operation allows samples of the amplifier offset and gain to be interleaved with weighing measurements. The use of correlated double sampling allows suppression of 1/f noise, offset and thermocouple effects within the bridge. Correlated double sampling involves alternating the polarity of excitation and dealing with the reversal of input polarity mathematically. Alternatively, bridge excitation can be increased to as much as ±10V, if one of several precision attenuation techniques is used to produce a precision divide operation on the reference signal. R1 2 350Ω BRIDGE 3 4 REF + REF – IN + R2 R1 AND R2 CAN BE USED TO INCREASE TOLERABLE AC COMPONENT ON REF SIGNALS Figure 40. Simple Bridge Connection U Another option is the use of a reference within the 5V input range of the LTC2411/LTC2411-1 and developing excitation via fixed gain, or LTC1043 based voltage multiplication, along with remote feedback in the excitation amplifiers, as shown in Figures 45 and 46. Figure 40 shows an example of a simple bridge connection. Note that it is suitable for any bridge application where measurement speed is not of the utmost importance. For many applications where large vessels are weighed, the average weight over an extended period of time is of concern and short term weight is not readily determined due to movement of contents, or mechanical resonance. Often, large weighing applications involve load cells located at each load bearing point, the output of which can be summed passively prior to the signal processing circuitry, actively with amplification prior to the ADC, or can be digitized via multiple ADC channels and summed mathematically. The mathematical summation of the output of multiple LTC2411/LTC2411-1’s provide the benefit of a root square reduction in noise. The low power consumption of the LTC2411/LTC2411-1 make it attractive for multidrop communication schemes where the ADC is located within the load-cell housing. A direct connection to a load cell is perhaps best incorporated into the load-cell body, as minimizing the distance to + 1 VCC SDO SCK CS 8 9 7 LT1019 LTC2411/ LTC2411-1 5 IN – GND 6 FO 10 2411 F40 W UU 33 LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO the sensor largely eliminates the need for protection devices, RFI suppression and wiring. The LTC2411/ LTC2411-1 exhibit extremely low temperature dependent drift. As a result, exposure to external ambient temperature ranges does not compromise performance. The incorporation of any amplification considerably complicates thermal stability, as input offset voltages and currents, temperature coefficient of gain settling resistors all become factors. The circuit in Figure 41 shows an example of a simple amplification scheme. This example produces a differential output with a common mode voltage of 2.5V, as determined by the bridge. The use of a true three amplifier instrumentation amplifier is not necessary, as the LTC2411/ LTC2411-1 have common mode rejection far beyond that of most amplifiers. The LTC1051 is a dual autozero amplifier that can be used to produce a gain of 30 before its input referred noise dominates the LTC2411/LTC2411-1 noise. This example shows a gain of 34, that is determined by a feedback network built using a resistor array containing eight individual resistors. The resistors are organized to optimize temperature tracking in the presence of thermal gradients. The second LTC1051 buffers the low noise 5V 3 0.1µF 8 0.1µF + – U1A 2 350Ω BRIDGE 1 RN1 16 6 11 2 6 7 10 4 15 14 3 – U1B 7 5 + RN1 = 5k × 8 RESISTOR ARRAY U1A, U1B, U2A, U2B = 1/2 LTC1051 Figure 41. Using Autozero Amplifiers to Reduce Input Referred Noise 34 U input stage from the transient load steps produced during conversion. The gain stability and accuracy of this approach is very good, due to a statistical improvement in resistor matching due to individual error contribution being reduced. A gain of 34 may seem low, when compared to common practice in earlier generations of load-cell interfaces, however the accuracy of the LTC2411/LTC2411-1 changes the rationale. Achieving high gain accuracy and linearity at higher gains may prove difficult, while providing little benefit in terms of noise reduction. At a gain of 100, the gain error that could result from typical open-loop gain of 160dB is –1ppm, however, worst-case is at the minimum gain of 116dB, giving a gain error of –158ppm. Worst-case gain error at a gain of 34, is –54ppm. The use of the LTC1051A reduces the worstcase gain error to –33ppm. The advantage of gain higher than 34, then becomes dubious, as the input referred noise sees little improvement1 and gain accuracy is potentially compromised. 1Input referred noise for A W U U 0.048µVRMS. V = 34 is approximately 0.05µVRMS, whereas at a gain of 50, it would be 5VREF 1 2 0.1µF 5V 8 U2A 4 5 12 3 1 2 3 4 4 REF + REF – IN + VCC 1 – + SDO SCK CS 8 9 7 8 9 13 6 – U2B 7 5 LTC2411/ LTC2411-1 IN – GND 6 FO 10 5 + 2411 F41 LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO Note that this 4-amplifier topology has advantages over the typical integrated 3-amplifier instrumentation amplifier in that it does not have the high noise level common in the output stage that usually dominates when an instrumentation amplifier is used at low gain. If this amplifier is used at a gain of 10, the gain error is only 10ppm and input referred noise is reduced to 0.15µVRMS. The buffer stages can also be configured to provide gain of up to 50 with high gain stability and linearity. Figure 42 shows an example of a single amplifier used to produce single-ended gain. This topology is best used in applications where the gain setting resistor can be made to match the temperature coefficient of the strain gauges. If the bridge is composed of precision resistors, with only one or two variable elements, the reference arm of the bridge can be made to act in conjunction with the feedback resistor to determine the gain. If the feedback resistor is incorporated into the design of the load cell, using resistors which match the temperature coefficient of the loadcell elements, good results can be achieved without the need for resistors with a high degree of absolute accuracy. The common mode voltage in this case, is again a function of the bridge output. Differential gain as used with a 350Ω bridge is: R1 + R2 A V = 9.95 = R1 + 175Ω 350Ω BRIDGE 3 2 + 1µF R1 4.99k AV = 9.95 = R1 + R2 R1 + 175Ω Figure 42. Bridge Amplification Using a Single Amplifier U Common mode gain is half the differential gain. The maximum differential signal that can be used is 1/4 VREF, as opposed to 1/2 VREF in the 2-amplifier topology above. Remote Half Bridge Interface As opposed to full bridge applications, typical half bridge applications must contend with nonlinearity in the bridge output, as signal swing is often much greater. Applications include RTD’s, thermistors and other resistive elements that undergo significant changes over their span. For single variable element bridges, the nonlinearity of the half VS 2.7V TO 5.5V 1 2 R1 25.5k 0.1% 24 PLATINUM 100Ω RTD 1 3 3 REF + VCC REF – LTC2411/ LTC2411-1 4 + IN 5 IN – GND 6 2411 F43 W UU Figure 43. Remote Half Bridge Interface 5V 0.1µF 1 2 6 175Ω 1µF R2 46.4k 20k 5 3 REF + REF – IN + LTC2411/ LTC2411-1 IN – GND 6 VCC 10µF 5V 0.1µV + + – 7 LTC1050S8 4 + 20k 4 2411 F42 35 LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO bridge output can be eliminated completely; if the reference arm of the bridge is used as the reference to the ADC, as shown in Figure 43. The LTC2411/LTC2411-1 can accept inputs up to 1/2 VREF. Hence, the reference resistor R1 must be at least 2 × the highest value of the variable resistor. In the case of 100Ω platinum RTD’s, this would suggest a value of 800Ω for R1. Such a low value for R1 is not advisable due to self-heating effects. A value of 25.5k is shown for R1, reducing self-heating effects to acceptable levels for most sensors. The basic circuit shown in Figure 43 shows connections for a full 4-wire connection to the sensor, which may be located remotely. The differential input connections will reject induced or coupled 60Hz interference, however, the reference inputs do not have the same rejection. If 60Hz or other noise is present on the RTD, a low pass filter is recommended as shown in Figure 44. Note that you cannot place a large capacitor directly at the junction of R1 and R2, as it will store charge from the sampling process. A better approach is to produce a low pass filter decoupled from the input lines with a high value resistor (R3). The use of a third resistor in the half bridge, between the variable and fixed elements gives essentially the same R2 10k 0.1% R1 10k, 5% R3 10k 5% 24 PLATINUM 100Ω RTD 1 3 Figure 44. Remote Half Bridge Sensing with Noise Suppression on Reference 36 U result as the two resistor version, but has a few benefits. If, for example, a 25k reference resistor is used to set the excitation current with a 100Ω RTD, the negative reference input is sampling the same external node as the positive input, but may result in errors if used with a long cable. For short cable applications, the errors may be acceptably low. If instead the single 25k resistor is replaced with a 10k 5% and a 10k 0.1% reference resistor, the noise level introduced at the reference, at least at higher frequencies, will be reduced. A filter can be introduced into the network, in the form of one or more capacitors, or ferrite beads, as long as the sampling pulses are not translated into an error. The reference voltage is also reduced, but this is not undesirable, as it will decrease the value of the LSB, although, not the input referred noise level. The circuit shown in Figure 44 shows a more rigorous example of Figure 43, with increased noise suppression and more protection for remote applications. Figure 45 shows an example of gain in the excitation circuit and remote feedback from the bridge. The LTC1043s provide voltage multiplication, providing ±10V from a 5V reference with only 1ppm error. The amplifiers are used at unity-gain and, hence, introduce a very little error due to 5V 5V 1 2 560Ω 3 REF + REF – LTC2411/ LTC2411-1 10k 10k 4 5 IN + IN – GND 6 VCC W UU + 1µF LTC1050 – 2411 F44 LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO 15V 15V 7 Q1 2N3904 20Ω 6 + – 3 1µF LTC1150 2 4 33Ω 1k 350Ω 10V BRIDGE 0.1µF –15V –10V 33Ω 15V 7 Q2 2N3906 –15V 6 20Ω + – 3 LTC1150 2 4 –15V 0.1µF 1k Figure 45. LTC1043 Provides Precise 4 × Reference for Excitation Voltages U 15V U1 4 LTC1043 10V 200Ω 8 * 11 47µF 7 5V LT1236-5 10V W UU + + 0.1µF 12 14 17 0.1µF 1 VCC LTC2411/ LTC2411-1 2 REF + 3 REF – 4 5 U2 LTC1043 5 * 2 6 IN + IN – GND 6 5V 13 10µF + 3 15 18 *FLYING CAPACITORS ARE 1µF FILM (MKP OR EQUIVALENT) SEE LTC1043 DATA SHEET FOR DETAILS ON UNUSED HALF OF U1 5V U2 4 LTC1043 8 * 1µF FILM 200Ω –10V 17 –10V 11 7 12 14 13 2411 F45 37 LTC2411/LTC2411-1 APPLICATIO S I FOR ATIO gain error or due to offset voltages. A 1µV/°C offset voltage drift translates into 0.05ppm/°C gain error. Simpler alternatives, with the amplifiers providing gain using resistor arrays for feedback, can produce results that are similar to bridge sensing schemes via attenuators. Note that the amplifiers must have high open-loop gain or gain error will be a source of error. The fact that input offset voltage has relatively little effect on overall error may lead one to use low performance amplifiers for this application. Note that the gain of a device such as an LF156, (25V/mV over temperature) will produce a worst-case error of –180ppm at a noise gain of 3, such as would be encountered in an inverting gain of 2, to produce –10V from a 5V reference. The error associated with the 10V excitation would be –80ppm. Hence, overall reference error could be as high as 130ppm, the average of the two. Figure 47 shows a similar scheme to provide excitation using resistor arrays to produce precise gain. The circuit is configured to provide 10V and –5V excitation to the bridge, producing a common mode voltage at the input to 5V TO OTHER DEVICES Figure 46. Use a Differential Multiplexer to Expand Channel Capability 38 U the LTC2411/LTC2411-1 of 2.5V, maximizing the AC input range for applications where induced 60Hz could reach amplitudes up to 2VRMS. The circuits in Figures 45 and 47 could be used where multiple bridge circuits are involved and bridge output can be multiplexed onto a single LTC2411/LTC2411-1, via an inexpensive multiplexer such as the 74HC4052. Figure 46 shows the use of an LTC2411/LTC2411-1 with a differential multiplexer. This is an inexpensive multiplexer that will contribute some error due to leakage if used directly with the output from the bridge, or if resistors are inserted as a protection mechanism from overvoltage. Although the bridge output may be within the input range of the A/D and multiplexer in normal operation, some thought should be given to fault conditions that could result in full excitation voltage at the inputs to the multiplexer or ADC. The use of amplification prior to the multiplexer will largely eliminate errors associated with channel leakage developing error voltages in the source impedance. 5V 16 12 14 15 11 1 5 2 4 8 9 10 A0 A1 2411 F46 W UU + 47µF 2 3 REF + REF – 1 VCC 74HC4052 13 3 6 4 5 LTC2411/ LTC2411-1 IN + IN – GND 6 LTC2411/LTC2411-1 PACKAGE DESCRIPTIO 0.007 (0.18) 0.021 ± 0.006 (0.53 ± 0.015) * DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE ** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. U MS10 Package 10-Lead Plastic MSOP (Reference LTC DWG # 05-08-1661) 0.118 ± 0.004* (3.00 ± 0.102) 10 9 8 7 6 0.193 ± 0.006 (4.90 ± 0.15) 0.118 ± 0.004** (3.00 ± 0.102) 12345 0.043 (1.10) MAX 0° – 6° TYP SEATING PLANE 0.007 – 0.011 (0.17 – 0.27) 0.034 (0.86) REF 0.0197 (0.50) BSC 0.005 ± 0.002 (0.13 ± 0.05) MSOP (MS10) 1100 39 LTC2411/LTC2411-1 TYPICAL APPLICATIO U 15V Q1 2N3904 22Ω RN1 10k 10V 350Ω BRIDGE TWO ELEMENTS VARYING 1 2 RN1 10k 5V 3 4 1 VCC LTC2411/ LTC2411-1 2 REF + 3 REF – 4 –5V 8 RN1 10k 5 6 7 RN1 10k 5 IN + IN – GND 6 20Ω + 1 C1 0.1µF 1/2 LT1112 3 5V + 2 LT1236-5 C3 47µF C1 0.1µF – 33Ω ×2 Q2, Q3 2N3906 ×2 20Ω 7 C2 0.1µF 15V RN1 IS CADDOCK T914 10K-010-02 8 – + 6 1/2 LT1112 4 5 –15V –15V 2411 F47 Figure 47. Use Resistor Arrays to Provide Precise Matching in Excitation Amplifier RELATED PARTS PART NUMBER LT1019 LT1025 LTC1050 LT1236A-5 LT1460 LTC2400 LTC2401/LTC2402 LTC2404/LTC2408 LTC2410 LTC2413 LTC2415 LTC2420 LTC2424/LTC2428 DESCRIPTION Precision Bandgap Reference, 2.5V, 5V Micropower Thermocouple Cold Junction Compensator Precision Chopper Stabilized Op Amp Precision Bandgap Reference, 5V Micropower Series Reference 24-Bit, No Latency ∆Σ ADC in SO-8 1-/2-Channel, 24-Bit, No Latency ∆Σ ADC in MSOP 4-/8-Channel, 24-Bit, No Latency ∆Σ ADC 24-Bit, Fully Differential, No Latency ∆Σ ADC 24-Bit, Fully Differential, No Latency ∆Σ ADC 24-Bit, No Latency ∆Σ ADC with 15Hz Output Rate 20-Bit, No Latency ∆Σ ADC in SO-8 4-/8-Channel, 20-Bit, No Latency ∆Σ ADC COMMENTS 3ppm/°C Drift, 0.05% Max Initial Accuracy 80µA Supply Current, 0.5°C Initial Accuracy No External Components 5µV Offset, 1.6µVP-P Noise 0.05% Max Initial Accuracy, 5ppm/°C Drift 0.075% Max Initial Accuracy, 10ppm/°C Max Drift 0.3ppm Noise, 4ppm INL, 10ppm Total Unadjusted Error, 200µA 0.6ppm Noise, 4ppm INL, 10ppm Total Unadjusted Error, 200µA 0.3ppm Noise, 4ppm INL, 10ppm Total Unadjusted Error, 200µA 0.16ppm Noise, 2ppm INL, 10ppm Total Unadjusted Error, 200µA Simultaneous 50Hz and 60Hz Rejection, 800nVRMS Noise Pin Compatible with the LTC2410 1.2ppm Noise, 8ppm INL, Pin Compatible with LTC2400 1.2ppm Noise, Pin Compatible with LTC2404/LTC2408 40 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 q FAX: (408) 434-0507 q 24111f LT/TP 0601 2K • PRINTED IN USA www.linear.com © LINEAR TECHNOLOGY CORPORATION 2000