MAX15053

19-5240; Rev 0; 5/10 High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator General Description The MAX15053 high-efficiency, current-mode, synchronous step-down switching regulator with integrated power switches delivers up to 2A of output current. The device operates from 2.7V to 5.5V and provides an output voltage from 0.6V up to 94% of the input voltage, making the device ideal for distributed power systems, portable devices, and preregulation applications. The MAX15053 utilizes a current-mode control architecture with a high gain transconductance error amplifier. The current-mode control architecture facilitates easy compensation design and ensures cycle-by-cycle current limit with fast response to line and load transients. The MAX15053 offers selectable skip-mode functionality to reduce current consumption and achieve a higher efficiency at light output load. The low RDS(ON) integrated switches ensure high efficiency at heavy loads while minimizing critical inductances, making the layout design a much simpler task with respect to discrete solutions. Utilizing a simple layout and footprint assures first-pass success in new designs. The MAX15053 features a 1MHz, factory-trimmed, fixedfrequency PWM mode operation. The high switching frequency, along with the PWM current-mode architecture, allows for a compact, all-ceramic capacitor design. The MAX15053 offers a capacitor-programmable softstart reducing inrush current, startup into PREBIAS operations, and a PGOOD open-drain output that can be used as an interrupt and for power sequencing. The MAX15053 is available in a 9-bump (3 x 3 array), 1.65mm x 1.65mm WLP package and is specified over the -40NC to +85NC temperature range. (typ) Low-Side MOSFETs at 5V TION KIT EVALUA BLE ILA AVA Features S Internal 30mI (typ) RDS(ON) High-Side and 18mI S Continuous 2A Output Current Over Temperature S 96% Efficiency with 3.3V Output at 2A S 1% Output Voltage Accuracy Over Load, Line, and MAX15053 Temperature S Operates from 2.7V to 5.5V Supply S Cycle-by-Cycle Overcurrent Protection S Adjustable Output from 0.6V to Up to 0.94 x VIN S Programmable Soft-Start S Factory-Trimmed, 1MHz Switching Frequency S Stable with Low-ESR Ceramic Output Capacitors S Safe-Startup Into Prebiased Output S External Reference Input S Skip-Mode Functionality S Enable Input/Power-Good Output S Fully Protected Against Overcurrent and Overtemperature S Input Undervoltage Lockout Ordering Information PART TEMP RANGE PIN-PACKAGE MAX15053EWL+ -40°C to +85°C 9 WLP +Denotes a lead(Pb)-free/RoHS-compliant package. Applications Distributed Power Systems Preregulators for Linear Regulators Portable Devices Notebook Power Server Power IP Phones _______________________________________________________________ Maxim Integrated Products 1 For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator MAX15053 ABSOLUTE MAXIMUM RATINGS IN, PGOOD to GND ................................................-0.3V to +6V LX to GND ..................................................-0.3V to (VIN + 0.3V) LX to GND .......................................-1V to (VIN + 0.3V) for 50ns EN, COMP, FB, SS/REFIN, SKIP to GND ...-0.3V to (VIN + 0.3V) LX Current (Note 1) ................................................... -5A to +5A Output Short-Circuit Duration ....................................Continuous Continuous Power Dissipation (TA = +70NC) 9-Bump WLP Multilayer Board (derate 14.1mW/NC above TA = +70NC) ....................1127mW Junction-to-Case Thermal Resistance (BJC)(Note 2) ......26NC/W Junction-to-Ambient Thermal Resistance (BJA)(Note 2) ...........................................71NC/W Operating Temperature Range .......................... -40NC to +85NC Operating Junction Temperature (Note 3) ......................+105NC Storage Temperature Range............................ -65NC to +150NC Soldering Temperature (reflow) ......................................+260NC Note 1: LX has internal clamp diodes to GND and IN. Applications that forward bias these diodes should not exceed the IC’s package power dissipation limits. Note 2: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a fourlayer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial. Note 3: Limit the junction temperature to +105NC for continuous operation at maximum output current. ELECTRICAL CHARACTERISTICS (VIN = 5V, TA = -40NC to +85NC, unless otherwise noted, typical values are at TA = +25NC.) (Note 4) PARAMETER IN Voltage Range IN Shutdown Supply Current IN Supply Current VIN Undervoltage Lockout Threshold VIN Undervoltage Lockout Hysteresis ERROR AMPLIFIER Transconductance Voltage Gain FB Set-Point Accuracy FB Input Bias Current COMP to Current-Sense Transconductance COMP Clamp Low POWER SWITCHES LX On-Resistance, High-Side pMOS LX On-Resistance, Low-Side nMOS High-Side Switch Current-Limit Threshold Low-Side Switch Sink CurrentLimit Threshold Low-Side Switch Source CurrentLimit Threshold IHSCL 30 18 4 4 4 mI mI A A A gMV AVEA VFB IFB gMC VFB = 0.65V, VSS = 0.6V Over line, load, and temperature VFB = 0.6V 594 -500 18 0.94 1.5 90 600 606 +500 mS dB mV nA A/V V IIN SYMBOL VIN VEN = 0V VEN = 5V, VFB = 0.65V, no switching LX starts switching, VIN rising LX stops switching, VIN falling CONDITIONS MIN 2.7 0.2 1.56 2.6 200 TYP MAX 5.5 2 2.3 2.7 UNITS V FA mA V mV 2 ______________________________________________________________________________________ High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator ELECTRICAL CHARACTERISTICS (continued) (VIN = 5V, TA = -40NC to +85NC, unless otherwise noted, typical values are at TA = +25NC.) (Note 4) PARAMETER LX Leakage Current RMS LX Output Current OSCILLATOR Switching Frequency Maximum Duty Cycle Minimum Controllable On-Time Slope Compensation Ramp Valley Slope Compensation Ramp Amplitude ENABLE EN Input High Threshold Voltage EN Input Low Threshold Voltage EN Input Leakage Current SKIP Input Leakage Current SOFT-START, PREBIAS, REFIN Soft-Start Current SS/REFIN Discharge Resistance SS/REFIN Prebias Mode Stop Voltage External Reference Input Range HICCUP Number of Consecutive CurrentLimit Events to Hiccup Timeout POWER-GOOD OUTPUT PGOOD Threshold PGOOD Threshold Hysteresis PGOOD VOL PGOOD Leakage THERMAL SHUTDOWN Thermal Shutdown Threshold Thermal Shutdown Hysteresis Temperature falling 150 20 NC NC VFB rising VFB falling IPGOOD = 5mA, VFB = 0.5V VPGOOD = 5V, VFB = 0.65V 0.535 8 1024 0.555 28 20 0.013 60 0.575 Events Clock Cycles V mV mV FA ISS RSS VSS/REFIN = 0.45V, sourcing ISS/REFIN = 10mA, sinking VSS/REFIN rising 0 10 8.3 0.58 IN - 1.8 FA I V V VEN rising VEN falling VEN = 5V VSKIP = VEN = 5V 1.45 0.4 0.025 25 V V FA FA VSLOPE Extrapolated to 100% duty cycle fSW DMAX 850 94 1000 95.8 70 1.15 320 1150 kHz % ns V mV SYMBOL VEN = 0V 2 CONDITIONS MIN TYP MAX 10 UNITS FA A MAX15053 3 Note 4: Specifications are 100% production tested at TA = +25NC. Limits over the operating temperature range are guaranteed by design and characterization. _______________________________________________________________________________________ High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator MAX15053 Typical Operating Characteristics (VIN = 5V, VOUT = 1.8V, ILOAD = 2A, Circuit of Figure 5, TA = +25NC, unless otherwise noted.) EFFICIENCY vs. OUTPUT CURRENT (PWM MODE) MAX15053 toc01 EFFICIENCY vs. OUTPUT CURRENT (PWM MODE) MAX15053 toc02 95 90 EFFICIENCY (%) 85 80 75 70 65 60 55 50 0 400 800 1200 1600 2000 95 90 EFFICIENCY (%) 85 80 75 70 65 60 55 50 95 90 EFFICIENCY (%) 85 80 75 70 65 60 VOUT = 3.3V VOUT = 3.3V VOUT = 1.8V VOUT = 1.2V VOUT = 2.5V VOUT = 1.8V VOUT = 1.2V VOUT = 2.5V VOUT = 1.5V VOUT = 1.2V VOUT = 1.5V VOUT = 2.5V VOUT = 1.5V VOUT = 1.8V VIN = 5V 2400 VIN = 3.3V 0 400 800 1200 1600 2000 2400 55 0 50 VIN = 5V 200 400 600 800 1000 1200 1400 1600 1800 2000 OUTPUT CURRENT (mA) OUTPUT CURRENT (mA) OUTPUT CURRENT (mA) EFFICIENCY vs. OUTPUT CURRENT (SKIP MODE) MAX15053 toc04 SWITCHING FREQUENCY vs. INPUT VOLTAGE 1080 SWITCHING FREQUENCY (kHz) 1060 1040 1020 1000 980 960 940 920 900 MAX15053 toc05 100 95 90 EFFICIENCY (%) 85 80 75 70 65 60 55 0 50 1100 VOUT = 2.5V VOUT = 1.5V VOUT = 1.2V VOUT = 1.8V VIN = 3.3V 200 400 600 800 1000 1200 1400 1600 1800 2000 2.7 3.2 3.7 4.2 4.7 5.2 OUTPUT CURRENT (mA) INPUT VOLTAGE (V) OUTPUT VOLTAGE vs. SUPPLY VOLTAGE MAX15053 toc06 OUTPUT VOLTAGE vs. OUTPUT CURRENT 1.89 1.87 OUTPUT VOLTAGE (V) 1.85 1.83 1.81 1.79 1.77 MAX15053 toc07 1.89 1.87 OUTPUT VOLTAGE (V) 1.85 1.83 1.81 1.79 1.77 1.75 2.7 VOUT = 3.3V VOUT = 5V IOUT = 0.5A 3.2 3.7 4.2 4.7 5.2 1.75 0 0.5 1.0 1.5 2.0 2.5 3.0 SUPPLY VOLTAGE (V) OUTPUT CURRENT (A) 4 ______________________________________________________________________________________ MAX15053 toc03 100 100 EFFICIENCY vs. OUTPUT CURRENT (SKIP MODE) 100 High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator Typical Operating Characteristics (continued) (VIN = 5V, VOUT = 1.8V, ILOAD = 2A, Circuit of Figure 5, TA = +25NC, unless otherwise noted.) SWITCHING WAVEFORMS (IOUT = 2A) MAX15053 toc9a MAX15053 LOAD-TRANSIENT RESPONSE MAX15053 toc08 VOUT 100mV/div AC-COUPLED VOUT 50mV/div AC-COUPLED ILX 1A/div IOUT 1A/div 0A 0A VLX 5V/div PWM MODE 40µs/div 400ns/div VIN = 5V SWITCHING WAVEFORMS MAX15053 toc09b SWITCHING WAVEFORM IN SKIP MODE (IOUT = 10mA) MAX15053 toc10 VOUT 50mV/div AC-COUPLED ILX 1A/div 0A VLX 5V/div VOUT 50mV/div AC-COUPLED ILX 1A/div VLX 5V/div VIN = 3.3V 400ns/div 10µs/div INPUT AND OUTPUT RIPPLE VOLTAGE WAVEFORM (IOUT = 2A) MAX15053 toc11 SHUTDOWN WAVEFORM MAX15053 toc12 ENABLE 5V/div INPUT 20mV/div AC-COUPLED OUTPUT 100mV/div AC-COUPLED VOUT 1V/div ILX 1A/div PGOOD 5V/div 400ns/div 10µs/div _______________________________________________________________________________________ 5 High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator MAX15053 Typical Operating Characteristics (continued) (VIN = 5V, VOUT = 1.8V, ILOAD = 2A, Circuit of Figure 5, TA = +25NC, unless otherwise noted.) SOFT-START WAVEFORMS (PWM) (IOUT = 2A) SOFT-START WAVEFORMS (SKIP MODE) (IOUT = 2A) MAX15053 toc13b MAX15053 toc13a ENABLE 5V/div VOUT 1V/div ILX 1A/div ENABLE 5V/div VOUT 1V/div ILX 1A/div PGOOD 5V/div PGOOD 5V/div 200µs/div 200µs/div QUIESCENT CURRENT vs. INPUT VOLTAGE 90 QUIESCENT CURRENT (nA) 80 70 60 50 40 30 20 10 0 2.7 3.2 3.7 4.2 4.7 5.2 SHORT-CIRCUIT HICCUP MODE MAX15053 toc15 MAX15053 toc14 100 VEN = 0V IIN 500mA/div VOUT 1V/div IOUT 5A/div 200µs/div INPUT VOLTAGE (V) RMS INPUT CURRENT vs. INPUT VOLTAGE 90 RMS INPUT CURRENT (mA) 80 70 60 50 40 30 20 10 0 2.7 MAX15053 toc16 FB VOLTAGE vs. TEMPERATURE MAX15053 toc17 100 606 604 FEEDBACK VOLTAGE (V) 602 600 598 596 SHORT CIRCUIT ON OUTPUT 3.2 3.7 4.2 4.7 5.2 INPUT VOLTAGE (V) NO LOAD 594 -40 -20 0 20 40 60 80 AMBIENT TEMPERATURE (°C) 6 ______________________________________________________________________________________ High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator Typical Operating Characteristics (continued) (VIN = 5V, VOUT = 1.8V, ILOAD = 2A, Circuit of Figure 5, TA = +25NC, unless otherwise noted.) SOFT-START WAVEFORMS (EXTERNAL REFIN) (PWM MODE) MAX15053 toc18a MAX15053 SOFT-START WAVEFORMS (EXTERNAL REFIN) (SKIP MODE) MAX15053 toc18b NO LOAD SS/REFIN 500mV/div VOUT 1V/div NO LOAD SS/REFIN 500mV/div VOUT 1V/div ILX 1A/div PGOOD 5V/div ILX 1A/div PGOOD 5V/div 200µs/div 200µs/div STARTING INTO A PREBIASED OUTPUT (IOUT = 2A) STARTING INTO A PREBIASED OUTPUT (NO LOAD) MAX15053 toc20a MAX15053 toc19 ENABLE 5V/div VOUT 1V/div ENABLE 5V/div VOUT 1V/div ILX 1A/div PGOOD 5V/div 200µs/div ILX 1A/div PGOOD 5V/div 200µs/div PWM MODE PWM MODE STARTING INTO A PREBIASED OUTPUT MAX15053 toc20b STARTING INTO A PREBIASED OUTPUT HIGHER THAN SET OUTPUT MAX15053 toc21 ENABLE 5V/div VOUT 1V/div 1.8V VOUT 500mV/div ILX 1A/div PGOOD 5V/div IL 1A/div SS/REFIN 500mV/div 10I LOAD AT OUT 200µs/div 400µs/div _______________________________________________________________________________________ 7 High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator MAX15053 Typical Operating Characteristics (continued) (VIN = 5V, VOUT = 1.8V, ILOAD = 2A, Circuit of Figure 5, TA = +25NC, unless otherwise noted.) CASE TEMPERATURE vs. AMBIENT TEMPERATURE MAX15053 toc22 INPUT CURRENT IN SKIP MODE vs. OUTPUT VOLTAGE 4.5 4.0 INPUT CURRENT (mA) 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 80 NO LOAD CASE TEMPERATURE (°C) 60 40 20 0 -20 -40 -40 -20 0 20 40 60 80 AMBIENT TEMPERATURE (°C) VCC = 5.0V VCC = 3.3V 1.2 1.7 2.2 2.7 3.2 OUTPUT VOLTAGE (V) 8 ______________________________________________________________________________________ MAX15053 toc23 100 5.0 High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator Pin Configuration TOP VIEW (BUMPS ON BOTTOM) MAX15053 GND A1 COMP B1 FB C1 LX A2 SKIP B2 SS/REFIN C2 IN A3 EN B3 PGOOD C3 MAX15053 WLP Pin Description BUMP A1 A2 A3 B1 B2 B3 C1 NAME GND LX IN COMP SKIP EN FB FUNCTION Analog Ground/Low-Side Switch Source Terminal. Connect to the PCB copper plane at one point near the input bypass capacitor return terminal. Inductor Connection. Connect LX to the switched side of the inductor. LX is high impedance when the IC is in shutdown mode. Input Power Supply. Input supply range is from 2.7V to 5.5V. Bypass with a minimum 10FF ceramic capacitor to GND. See Figures 5 and 6. Voltage Error-Amplifier Output. Connect the necessary compensation network from COMP to GND. See the Closing the Loop: Designing the Compensation Circuitry section. Skip-Mode Input. Connect to EN to select skip mode or leave unconnected for normal operation. Enable Input. EN is a digital input that turns the regulator on and off. Drive EN high to turn on the regulator. Connect to IN for always-on operation. Feedback Input. Connect FB to the center tap of an external resistor-divider from the output to GND to set the output voltage from 0.6V up to 94% of VIN. Soft-Start/External Voltage Reference Input. Connect a capacitor from SS/REFIN to GND to set the startup time. See the Setting the Soft-Start Time section for details on setting the soft-start time. Apply a voltage reference from 0V to VIN - 1.5V to drive soft-start externally. Open-Drain Power-Good Output. PGOOD goes high when FB is above 555mV and pulls low if FB is below 527mV. C2 SS/REFIN C3 PGOOD _______________________________________________________________________________________ 9 High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator MAX15053 Block Diagram SKIP EN IN BIAS GENERATOR EN LOGIC, IN UVLO THERMAL SHDN SHDN SKIP-MODE LOGIC SKPM HIGH-SIDE CURRENT LIMIT VOLTAGE REFERENCE LX CURRENT-SENSE AMPLIFIER LX IN IN STRONG PREBIASED FORCED START 0.58V SS/REFIN SS/REFIN BUFFER 0.6V SKPM CONTROL LOGIC CK IN LX GND 10µA PWM COMPARATOR ERROR AMPLIFIER LOW-SIDE SOURCE-SINK CURRENT LIMIT AND ZEROCROSSING COMPARATOR SINK SOURCE FB COMP RAMP OSCILLATOR RAMP GEN CK C ZX SKPM MAX15053 PGOOD POWER-GOOD COMPARATOR 0.555V RISING, 0.527V FALLING 10 _____________________________________________________________________________________ High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator Detailed Description The MAX15053 high-efficiency, current-mode switching regulator can deliver up to 2A of output current. The MAX15053 provides output voltages from 0.6V to 0.94 x VIN from 2.7V to 5.5V input supplies, making the device ideal for on-board point-of-load applications. The MAX15053 delivers current-mode control architecture using a high gain transconductance error amplifier. The current-mode control architecture facilitates easy compensation design and ensures cycle-by-cycle current limit with fast response to line and load transients. The MAX15053 features a 1MHz fixed switching frequency, allowing for all-ceramic capacitor designs and fast transient responses. The high operating frequency minimizes the size of external components. The MAX15053 is available in a 1.65mm x 1.65mm (3 x 3 array) x 0.5mm pitch WLP package. The MAX15053 offers a selectable skip-mode functionality to reduce current consumption and achieve a higher efficiency at light output loads. The low RDS(ON) integrated switches (30mI high-side and 18mI low-side, typ) ensure high efficiency at heavy loads while minimizing critical inductances, making the layout design a much simpler task with respect to discrete solutions. Utilizing a simple layout and footprint assures first-pass success in new designs. The MAX15053 features 1MHz Q15%, factory-trimmed, fixed-frequency PWM mode operation. The MAX15053 also offers capacitor-programmable, soft-start reducing inrush current, startup into PREBIAS operation, and a PGOOD open-drain output for sequencing with other devices. The controller logic block is the central processor that determines the duty cycle of the high-side MOSFET under different line, load, and temperature conditions. Under normal operation, where the current-limit and temperature protection are not triggered, the controller logic block takes the output from the PWM comparator and generates the driver signals for both high-side and low-side MOSFETs. The control logic block controls the break-before-make logic and all the necessary timing. limit is reached. The low-side MOSFET turns on for the remainder of the oscillation cycle. The MAX15053 can soft-start into a prebiased output without discharging the output capacitor. In safe prebiased startup, both low-side and high-side MOSFETs remain off to avoid discharging the prebiased output. PWM operation starts when the voltage on SS/REFIN crosses the voltage on FB. The MAX15053 can start into a prebiased voltage higher than the nominal set point without abruptly discharging the output. Forced PWM operation starts when the SS/REFIN voltage reaches 0.58V (typ), forcing the converter to start. In case of prebiased output, below or above the output nominal set point, if low-side sink current-limit threshold (set to the reduced value of -0.4A (typ) for the first 32 clock cycles and then set to -4A typ) is reached, the low-side switch turns off before the end of the clock period, and the high-side switch turns on until one of the following conditions is satisfied: • High-side source current hits the reduced high-side current limit (0.4A, typ); in this case, the high-side switch is turned off for the remaining time of the clock period. The clock period ends. Reduced high-side current limit is activated to recirculate the current into the high-side power switch rather than into the internal high-side body diode, which could be damaged. Low-side sink current limit is provided to protect the low-side switch from excessive reverse current during prebiased operation. MAX15053 Starting into a Prebiased Output • Controller Function–PWM Logic In skip mode operation, the prebias output needs to be lower than the set point. The MAX15053 features independent device enable control and power-good signal that allow for flexible power sequencing. Drive the enable input (EN) high to enable the regulator, or connect EN to IN for always-on operation. Power-good (PGOOD) is an open-drain output that asserts when VFB is above 555mV (typ), and deasserts low if VFB is below 527mV (typ). The MAX15053 utilizes a soft-start feature to slowly ramp up the regulated output voltage to reduce input inrush current during startup. Connect a capacitor from SS/REFIN to GND to set the startup time (see the Setting the Soft-Start Time section for capacitor selection details). Enable Input The high-side MOSFET turns on at the beginning of the oscillator cycle and turns off when the COMP voltage crosses the internal current-mode ramp waveform, which is the sum of the slope compensation ramp and the current-mode ramp derived from inductor current (current-sense block). The high-side MOSFET also turns off if the maximum duty cycle is 94%, or when the current Programmable Soft-Start (SS/REFIN) ______________________________________________________________________________________ 11 High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator MAX15053 A high-gain error amplifier provides accuracy for the voltage-feedback loop regulation. Connect the necessary compensation network between COMP and GND (see the Compensation Design Guidelines section). The error-amplifier transconductance is 1.5mS (typ). COMP clamp low is set to 0.94V (typ), just below the slope ramp compensation valley, helping COMP to rapidly return to the correct set point during load and line transients. The PWM comparator compares COMP voltage to the current-derived ramp waveform (LX current to COMP voltage transconductance value is 18A/V typ). To avoid instability due to subharmonic oscillations when the duty cycle is around 50% or higher, a slope compensation ramp is added to the current-derived ramp waveform. The compensation ramp slope (0.3V x 1MHz = 0.3V/Fs) is equivalent to half the inductor current downslope in the worst case (load 2A, current ripple 30% and maximum duty-cycle operation of 94%). The slope compensation ramp valley is set to 1.15V (typ). When the converter output is shorted or the device is overloaded, each high-side MOSFET current-limit event (4A typ) turns off the high-side MOSFET and turns on the low-side MOSFET. On each current-limit event a 3-bit counter is incremented. The counter is reset after three consecutive high-side MOSFETs turn on without reaching current limit. If the current-limit condition persists, the counter fills up reaching eight events. The control logic then discharges SS/REFIN, stops both high-side and low-side MOSFETs, and waits for a hiccup period (1024 clock cycles typ) before attempting a new softstart sequence. The hiccup mode is also enabled during soft-start time. The MAX15053 contains an internal thermal sensor that limits the total power dissipation to protect the device in the event of an extended thermal fault condition. When the die temperature exceeds +150NC (typ), the thermal sensor shuts down the device, turning off the DC-DC converter to allow the die to cool. After the die temperature falls by 20NC (typ), the device restarts, following the soft-start sequence. The MAX15053 operates in skip mode when SKIP is connected to EN. When in skip mode, LX output becomes high impedance when the inductor current falls below Error Amplifier PWM Comparator 200mA (typ). The inductor current does not become negative. If during a clock cycle the inductor current falls below the 200mA threshold (during off-time), the low side turns off. At the next clock cycle, if the output voltage is above set point, the PWM logic keeps both high-side and low-side MOSFETs off. If instead the output voltage is below the set point, the PWM logic drives the highside on for a minimum fixed on-time (300ns typ). In this way the system can skip cycles, reducing frequency of operations, and switches only as needed to service load at the cost of an increase in output voltage ripple (see the Skip Mode Frequency and Output Ripple section). In skip mode, power dissipation is reduced and efficiency is improved at light loads because power MOSFETs do not switch at every clock cycle. Applications Information The MAX15053 output voltage is adjustable from 0.6V up to 94% of VIN by connecting FB to the center tap of a resistor-divider between the output and GND (Figure 1). Choose R1 and R2 so that the DC errors due to the FB input bias current (Q500nA) do not affect the output voltage accuracy. With lower value resistors, the DC error is reduced, but the amount of power consumed in the resistor-divider increases. A typical value for R2 is 10kI, but values between 5kI and 50kI are acceptable. Once R2 is chosen, calculate R1 using: V  R1 = R2 ×  OUT − 1  VFB  where the feedback threshold voltage, VFB = 0.6V (typ). When regulating for an output of 0.6V in skip mode, short FB to OUT and keep R2 connected from FB to GND. A high-valued inductor results in reduced inductor ripple current, leading to a reduced output ripple voltage. However, a high-valued inductor results in either a larger physical size or a high series resistance (DCR) and a lower saturation current rating. Typically, choose an inductor value to produce a current ripple equal to 30% of load current. Choose the inductor with the following formula: V  VOUT L= × 1 − OUT  fSW × LIR × ILOAD  VIN  where fSW is the internally fixed 1MHz switching frequency, and LIR is the desired inductor current ratio (typically Setting the Output Voltage Overcurrent Protection and Hiccup Thermal-Shutdown Protection Inductor Selection Skip Mode Operation 12 _____________________________________________________________________________________ High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator MAX15053 FEEDBACK DIVIDER ERROR AMPLIFIER POWER MODULATOR SLOPE COMPENSATION RAMP VOUT OUTPUT FILTER AND LOAD VIN C FB R1 *CFF VFB R2 COMP VCOMP RC CC COMPARATOR gMC QHS PWM CONTROL LOGIC QLS IL L DCR IOUT VOUT ESR COUT RLOAD gMV ROUT VCOMP GMOD IOUT REF ROUT = 10AVEA(dB)/20/gMV *NOTE: CFF IS OPTIONAL AND DESIGNED TO EXTEND THE REGULATOR’S GAIN BANDWIDTH AND INCREASED PHASE MARGIN FOR SOME LOW-DUTY CYCLE APPLICATIONS. NOTE: THE GMOD STAGE SHOWN ABOVE MODELS THE AVERAGE CURRENT OF THE INDUCTOR, IL, INJECTED INTO THE OUTPUT LOAD, IOUT, e.g., IL = IOUT. THIS CAN BE USED TO SIMPLIFY/MODEL THE MODULATION/CONTROL/POWER STATE CIRCUITRY SHOWN WITHIN THE BOXED AREA. Figure 1. Peak Current-Mode Regulator Transfer Model set to 0.3). In addition, the peak inductor current, IL_PK, must always be below the minimum high-side currentlimit value, IHSCL, and the inductor saturation current rating, IL_SAT. Ensure that the following relationship is satisfied: 1 IL_PK = ILOAD + ∆IL < min IHSCL_, IL_SAT 2 ( ) fSW is the switching frequency (1MHz), and ILOAD is the output load. The impedance of the input capacitor at the switching frequency should be less than that of the input source so high-frequency switching currents do not pass through the input source, but are instead shunted through the input capacitor. The input capacitor must meet the ripple current requirement imposed by the switching currents. The RMS input ripple current is given by: V  OUT × (VIN − VOUT )  IRIPPLE =  ILOAD VIN     The input capacitor reduces the peak current drawn from the input power supply and reduces switching noise in the device. The total input capacitance must be equal to or greater than the value given by the following equation to keep the input ripple voltage within the specification and minimize the high-frequency ripple current being fed back to the input source: CIN = ILOAD V × OUT fSW × ∆VIN_RIPPLE VIN Input Capacitor Selection where IRIPPLE is the input RMS ripple current. The key selection parameters for the output capacitor are capacitance, ESR, ESL, and voltage rating. The parameters affect the overall stability, output ripple voltage, and transient response of the DC-DC converter. The output ripple occurs due to variations in the charge stored in the output capacitor, the voltage drop due to the capacitor’s ESR, and the voltage drop due to the Output Capacitor Selection where DVIN_RIPPLE is the maximum-allowed input ripple voltage across the input capacitors and is recommended to be less than 2% of the minimum input voltage, ______________________________________________________________________________________ 13 High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator MAX15053 capacitor’s ESL. Estimate the output-voltage ripple due to the output capacitance, ESR, and ESL as follows: ∆VOUT =  VOUT  VOUT   1 × 1 −   × R ESR_COUT + fSW × L  VIN   8 × fSW × C OUT  the inductor and output capacitor values. After a short time, the controller responds by regulating the output voltage back to the predetermined value. Use higher COUT values for applications that require light load operation or transition between heavy load and light load, triggering skip mode, causing output undershooting or overshooting. When applying the load, limit the output undershoot by sizing COUT according to the following formula: C OUT ≅ ∆ILOAD 3fCO x ∆VOUT For ceramic capacitors, ESR contribution is negligible: R ESR_OUT > 1 8 × fSW × C OUT Use these equations for initial output-capacitor selection. Determine final values by testing a prototype or an evaluation circuit. A smaller ripple current results in less output-voltage ripple. Since the inductor ripple current is a factor of the inductor value, the output-voltage ripple decreases with larger inductance. Use ceramic capacitors for low ESR and low ESL at the switching frequency of the converter. The ripple voltage due to ESL is negligible when using ceramic capacitors. Load-transient response also depends on the selected output capacitance. During a load transient, the output instantly changes by ESR x DILOAD. Before the controller can respond, the output deviates further, depending on where DILOAD is the total load change, fCO is the regulator unity-gain bandwidth (or zero crossover frequency), and DVOUT is the desired output undershooting. When removing the load and entering skip mode, the device cannot control output overshooting, since it has no sink current capability; see the Skip Mode Frequency and Output Ripple section to properly size COUT. In skip mode, the switching frequency (fSKIP) and output ripple voltage (VOUT-RIPPLE) shown in Figure 2 are calculated as follows: tON is a fixed time (300ns, typ); the peak inductor current reached is: V − VOUT I SKIP −LIMIT = IN × t ON L Skip Mode Frequency and Output Ripple IL ISKIP-LIMIT ILOAD tON tOFF1 tOFF2 = n × tCK VOUT VOUT-RIPPLE Figure 2. Skip Mode Waveform 14 _____________________________________________________________________________________ High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator tOFF1 is the time needed for inductor current to reach the zero-current crossing limit (~ 0A): t OFF1 = L × I SKIP −LIMIT VOUT the inductor’s pole frequency is shifted beyond the gain bandwidth of the regulator. System stability is provided with the addition of a simple series capacitor-resistor from COMP to GND. This pole-zero combination serves to tailor the desired response of the closed-loop system. The basic regulator loop consists of a power modulator (comprising the regulator’s pulse-width modulator, current sense and slope compensation ramps, control circuitry, MOSFETs, and inductor), the capacitive output filter and load, an output feedback divider, and a voltage-loop error amplifier with its associated compensation circuitry. See Figure 1. The average current through the inductor is expressed as: IL = G MOD × VCOMP where IL is the average inductor current and GMOD is the power modulator’s transconductance. For a buck converter: VOUT = R LOAD × IL where RLOAD is the equivalent load resistor value. Combining the above two relationships, the power modulator’s transfer function in terms of VOUT with respect to VCOMP is: VOUT VCOMP R ×I = LOAD L = RLOAD × G MOD IL GMOD MAX15053 During tON and tOFF1, the output capacitor stores a charge equal to (see Figure 2): 1 1 2 + L x (I SKIP −LIMIT − ILOAD ) x   VIN − VOUT VOUT   ∆Q OUT = 2 During tOFF2 (= n x tCK, number of clock cycles skipped), output capacitor loses this charge: t OFF2 = ∆Q OUT ⇒ ILOAD 1 1 2 L x (I SKIP − LIMIT − ILOAD ) x  +  VIN − VOUT VOUT   t OFF2 = 2 xILOAD Finally, frequency in skip mode is: fSKIP = 1 t ON + t OFF1 + t OFF2 Output ripple in skip mode is: VOUT −RIPPLE = VCOUT −RIPPLE + VESR−RIPPLE −I (I ) x t ON = SKIP −LIMIT LOAD C OUT + R ESR,COUT x (I SKIP −LIMIT − ILOAD )  L x I SKIP −LIMIT  VOUT −RIPPLE =  + RESR,COUT  C OUT x (VIN − VOUT )     x (I SKIP −LIMIT − ILOAD ) The peak current-mode controller’s modulator gain is attenuated by the equivalent divider ratio of the load resistance and the current-loop gain’s impedance. GMOD becomes: 1 G MOD (DC) = g MC ×  RLOAD  × K × (1 − D) − 0.5 1 +  fSW × L  S   where RLOAD = VOUT/IOUT(MAX), fSW is the switching frequency, L is the output inductance, D is the duty cycle (VOUT/VIN), and KS is a slope compensation factor calculated from the following equation: S V ×f × L × g MC K S = 1 + SLOPE = 1 + SLOPE SW SN (VIN − VOUT ) where: V S SLOPE = SLOPE = VSLOPE × fSW t SW SN = To limit output ripple in skip mode, size COUT based on the above formula. All the above calculations are applicable only in skip mode. The MAX15053 uses a fixed-frequency, peak-current-mode control scheme to provide easy compensation and fast transient response. The inductor peak current is monitored on a cycle-by-cycle basis and compared to the COMP voltage (output of the voltage error amplifier). The regulator’s duty cycle is modulated based on the inductor’s peak current value. This cycle-by-cycle control of the inductor current emulates a controlled current source. As a result, Compensation Design Guidelines (VIN − VOUT ) L × g MC 15 ______________________________________________________________________________________ High-Efficiency, 2A, Current-Mode Synchronous, Step-Down Switching Regulator MAX15053 1ST ASYMPTOTE R2 × (R1 + R2)-1 × 10AVEA(dB)/20 × gMC × RLOAD × {1 + RLOAD × [KS × (1 - D) - 0.5] × (L × fSW)-1}-1 GAIN 2ND ASYMPTOTE R2 × (R1 + R2)-1 × gMV × (2GCC)-1 × gMC × RLOAD × {1 + RLOAD × [KS × (1 - D) - 0.5] × (L × fSW)-1}-1 3RD ASYMPTOTE R2 × (R1 + R2)-1 × gMV × (2GCC)-1 × gMC × RLOAD × {1 + RLOAD × [KS × (1 - D) - 0.5] × (L × fSW)-1}-1 × (2GCOUT × {RLOAD-1 + [KS × (1 - D) - 0.5] × (L × fSW)-1}-1)-1 4TH ASYMPTOTE R2 × (R1 + R2)-1 × gMV × RC × gMC × RLOAD × {1 + RLOAD × [KS × (1 - D) - 0.5] × (L × fSW)-1}-1 × (2GCOUT × {RLOAD-1 + [KS × (1 - D) - 0.5] × (L × fSW)-1}-1)-1 3RD POLE (DBL) 2ND ZERO 0.5 × fSW (2GCOUTESR)-1 fCO FREQUENCY UNITY 1ST POLE [2GCC × (10AVEA(dB)/20 - gMV-1)]-1 2ND POLE fPMOD* 1ST ZERO (2GCCRC)-1 5TH ASYMPTOTE R2 × (R1 + R2)-1 × gMV × RC × gMC × RLOAD × {1 + RLOAD × [KS × (1 - D) - 0.5] × (L × fSW)-1}-1 × (2GCOUT × {RLOAD-1 + [KS × (1 - D) - 0.5] × (L × fSW)-1}-1)-1 × (0.5 × fSW)2 × (2Gf)-2 NOTE: ROUT = 10AVEA(dB)/20 × gMV-1 fPMOD = [2GCOUT × (ESR + {RLOAD-1 + [KS × (1 - D) - 0.5] × (L × fSW)-1}-1)]-1 WHICH FOR ESR C OUT × VOUT × I SS (IHSCL_ − IOUT ) × VFB and where the ESR is much smaller than the parallel combination of the equivalent load resistance and the current loop impedance, e.g.,: ESR