AN912

AN912 Designing LF Talkback for a Magnetic Base Station Author: Ruan Lourens Microchip Technology Inc. MAIN BUILDING BLOCKS Figure 1 shows the main building blocks that make up the LF Talkback system described in this document. The base station generates a strong magnetic field by setting up resonance in a serial resonant tank. The circulating energy in the resonant tank typically generates 300V peak-to-peak voltage across the transmitting antenna coil at 125 kHz. The transponder, whether active or passive, is magnetically coupled to the base station’s transmitting coil and the transponder’s magnetic loading has a small effect on the quality factor (Q) of the transmitter resonant tank. Talkback is accomplished by changing or modulating the magnetic loading and can be observed as small voltage changes across the base station's resonant transmitter coil. The difficulty is to detect a few mV of modulation on the 300V peak-to-peak carrier. INTRODUCTION This application note builds on application note AN232 Low Frequency Magnetic Transmitter Design (DS00232). It covers the design process to implement LF Talkback functionality. AN232 covers some of the magnetism basics and design principles to implement the drive circuitry. LF Talkback generally refers to the process in which a transponder can communicate back to a magnetic transmitter base station by loading the generated magnetic field. By measuring the small changes in the transmitter coil's voltage, used to generate the field, the communications’ data is extracted. LF Talkback is commonly used in RFID, automotive transponders, active transponders, and many other bidirectional LF communications topologies. This document will cover the different stages needed to implement a typical LF Talkback system and explain the process in choosing the different stage characteristics. It explains the various performance and cost tradeoffs made for the reference design and how it can be adapted to better suit the readers needs. FIGURE 1: Peak Detector DC Decouple Low Pass Filter Data Slicer VREF M Transponder Base Station Data Data  2004 Microchip Technology Inc. DS00912A-page 1 AN912 A high voltage peak detector is used to extract the basic envelope of the base station's resonant tank. The output of the peak detector will be 150 VDC with about 2V peak-to-peak of carrier ripple at 125 kHz and then about 2 mV of modulated signal. The modulation signal strength is mostly dependent on the distance between the transponder and the transmitter coil as the magnetic coupling decreases to the third power of the distance between the two devices. The next stage is a passive high-pass filter to decouple or block the high DC voltage. The DC extracted voltage is then fed into a low-pass filter, leaving the required modulating signal. The last stage is the data slicer that compares the modulating signal to some reference to extract the original signal sent by the transponder. LF Talkback receiver can be thought of as detecting and decoding an amplitude modulation (AM) signal that has a very low modulation index on a relatively large carrier. THE PEAK DETECTOR There are a number of aspects to consider in designing a peak detector for this application: 1. 2. The peak detector has to be able to operate at the high voltages of the resonant tank. Maintain a good tank Q or, in other words, it should not add unnecessary loading on the main resonant tank. If it does load the tank, it will result in a lower modulation voltage induced by the transponder. Reduce carrier ripple as far as possible. Maintain the modulation signal. Have a fast large swing dynamic response and be able to settle quickly after the field is turned on. Cost of the system. 3. 4. 5. 6. SYSTEM ASSUMPTIONS The LF Talkback system designed in this document is targeted for a LF base station that has the following characteristics and is based on the design as per AN232: • The LF Talkback signal is amplitude modulated at 200 µs multiples. This is also referred to as the basic pulse element period or TE. • The tank is driven by a 12V half-bridge driver. • The tank inductance is 162 µH and the resonant capacitor is 10 nF with a resonant frequency of 125 kHz. • The tank Q is 25. As a result, the tank or carrier voltage is 300V peak-to-peak or 150V 0-to-peak. • Transponder induced modulation of 2 mV in magnitude needs to be detected. To get an understanding of the impedances involved, lets consider the following: using Equation 1, the equivalent parallel resistance of the tank is 3.18 kΩ. The additional parallel impedance that a transponder represents to induce a 2 mV signal on the tank is in the order of 500 MΩ. What the LF Talkback system detects is the result of a 500 MΩ resistor being switched in and out in parallel with the tank at the data rate. Therefore, it is very important that the peak detector have a highimpedance at the data rate to maintain good sensitivity. Some of the peak detector requirements are conflicting and as a result, the designer has to find an acceptable compromise with the final system performance in mind. One can sacrifice a specific parameter and make up for it in a later stage where optimization of that aspect is easily accomplished. As an example to optimize requirement 3, one needs to increase the size of the capacitor C2 (Figure 2), but that will negatively affect requirements 2, 4 and 5 if a passive peak detector is used. An active peak detector could have solved the conflict, but at the 600V swing, one has little choice but to use a passive peak detector while maintaining a low-cost design. A relatively low capacitance value is chosen for C1 of 1 nF. This maintains the dynamic response requirement for settling quickly after the field is applied and does not load the tank unnecessarily. Capacitor C2 should have at least a 300 VDC peak rating and a high tolerance capacity is acceptable to save cost. An ultra fast diode is required in the peak detector with a 400V or better rating and low junction capacitance. A UF1005 diode was chosen, it has a 600V rating and 10 pF of junction capacitance. FIGURE 2: D1 C1 L1 C2 R1 HV-Env EQUATION 1: THE EFFECTIVE PARALLEL IMPEDANCE OF A RESONANT TANK = 2πLFCQ RPARALLEL L = Tank inductance in H = 162 µH Fc = Center frequency of tank = 125 kHz Q = Tank quality factor = 25 DS00912A-page 2  2004 Microchip Technology Inc. AN912 The envelope detector with only D1 and C2 has a greatly different response to increasing and decreasing voltage amplitudes of the resonant tank. The voltage designated by the signal HV_Env (Figure 2) rises quickly with increasing tank amplitudes because D1 has a low-impedance in forward conduction. The tank voltage decreases slower when the tank amplitude is lowered because C2 can only discharge through D1, which has a high-impedance in the reverse direction. The situation can be remedied to some extent by the introduction of R1 which helps to discharge C2, but the value of R1 should be high enough to maintain a good tank Q as per requirement 2 above. A 10 MΩ value for R1 works well, but note that R1 needs to be implemented as a series of two resistors. This is done to stay within the safe voltage range of 0805 resistors are used. The 125 kHz carrier ripple voltage, without R1, is about 2V peak-to-peak and is due to the junction capacitance and reverse leakage of D1. The addition of R1 has little effect on the ripple voltage, but does improve the detectors dynamic performance at the data rate. The carrier ripple voltage will be filtered out at a later stage where a more effective solution can be implemented. FIGURE 3: HV-Env LP Filter C R The system can be simplified as shown in Figure 3. The output of the peak detector can be simplified as the step response source with a 150V amplitude that also has the carrier and data signals superimposed on it as described earlier. The output response of the decoupling stage is given by Equation 2. This is also the input signal to the low-pass filter. EQUATION 2: V = 150e-t/τ τ = RC It is useful to think in terms of τ (RC time constant) because the voltage across the resistor reduces by a factor of 0.368 as every τ second elapses. The exponential decay curve, for the voltage across R, is shown in Figure 4 and indicates that the initial voltage decays rapidly, but settles out slower as the voltage is reduced across the resistor. The system must be allowed to settle for a long enough period so that the step response voltage has reduced to a voltage that is smaller than the modulation voltage. The required value for RC, or τ, can be calculated using Equation 3, based on the following assumptions: • The system needs to be able to start LF communications 200 µs after the resonant tank has stabilized. • The decoupler should settle to at least half the data modulation voltage. THE DC DECOUPLER CONFLICTS The HV_Env signal (Figure 2) consists of three main components: 1. 2. 3. A 150V DC signal, as a result of the peak detector. 2V peak-to-peak ripple voltage at the carrier frequency. The modulated data signal at a TE of 200 µs and a 2 mV peak-to-peak amplitude, highest fundamental harmonic content is at 2.5 kHz [1/(2*200 uS)], irrespective of the modulation scheme used (i.e., Manchester, PWM etc.). The aim of the decoupling stage is to reject the high DC voltage without adding unnecessary loading to the tank via the peak detector. It should also have a fast dynamic response and stabilize quickly after the tank is energized. The dynamic response of the LF Talkback system is the major design hurdle to overcome as far as the decoupling stage is concerned. The problem is aggravated when the transponder needs to communicate on the LF link soon after the base station communicated with the transponder. The base station typically uses On Off Keying (OOK) modulation to communicate to the transponder. This means the tank resonance is completely halted and then started up to transfer data via the magnetic link. The decoupling stage experiences large “step” responses as data is transmitted to the transponder. The tank can ramp up to its full resonant amplitude in 100 µs to 400 µs depending on the drive system used.  2004 Microchip Technology Inc. DS00912A-page 3 AN912 FIGURE 4: 160 140 120 100 Voltage 80 60 40 20 0 0 1 2 τ 3 4 5 EQUATION 3: EQUATION 4: tSETTLE ln(Vo/VSETTLE) tSETTLE = 200 µS VO = 150V VSETTLE = 1 mV FC = 1 2πτ τ= Using Equation 3, τ was calculated to be 16.78 µs. The question now is how will the data signal be affected by the decoupling stage? The decoupling stage, shown in Figure 3, is also a high-pass filter and it was calculated that the RC time constant needs to be 16.78 µs to satisfy the transient response requirement. The 3 dB cutoff frequency, for a τ, of 16.78 µs is calculated as 9.48 kHz using Equation 4. This means that the decoupling stage will only pass one quarter of the original data signal at 2.5 kHz, which is not desirable from a signal-to-noise ratio perspective. From a data signal conservation, or high-pass filter point-of-view, τ should be at least 64 µs. The conflicting τ requirement shows that a basic high-pass filter is not sufficient as a decoupler unless either dynamic response or data signal strength is sacrificed. DS00912A-page 4  2004 Microchip Technology Inc. AN912 AN IMPROVED DECOUPLER From the previous section, it is clear that a high-pass filter is needed with either a controllable τ or a nonlinear τ that is based on the voltage across the output of the decoupler. Both approaches will be covered and the latter solution is shown in Figure 5. FIGURE 6: HV-Env C R1 S1 R2 2.5V -2.5V FIGURE 5: 2.5V -2.5V LP Filter HV-Env C R LP Filter The addition of the two diodes, shown in Figure 5, results in a nonlinear τ with respect to voltage because it effectively lowers the R component of τ whenever the voltage is either above 3.1V or below -3.1V. In a practical circuit, the diodes will start conducting when the tank is turned on and the voltage, across the resistor R, is around 3 volts, after the tank has stabilized. Previously, τ was calculated with an initial voltage of 150 VDC, but if the calculation is repeated with an initial voltage of 3 VDC, then the required τ comes to 25 µs. The RC time constant is improved by a significant factor from 16.78 µs to 25 µs with the additional diodes, however, it is still not in the 64 µs ball park. The diodes have the additional advantage in that they protect the low-pass filter from the large positive and negative voltages that develop across the resistor during tank transient periods. The final part to solving the time constant problem is to add an additional resistor via a switch, as shown in Figure 6. The switch is closed to reduce τ from 64 µs to 25 µs during transient periods and opened while data is received via the LF Talkback link. The final part of the decoupling stage is to lower the output impedance by adding an active buffer in the form of an inverting amplifier that has an input resistance equal to R1. The use of an inverting amplifier has the additional advantages that it can add gain and a single order low-pass filter to the decoupler, as shown in Figure 7. The gain is equal to the ratio of R3/R1, and the low pass cutoff frequency is set by R3 and C2, as per Equation 4. The low pass cutoff frequency should be chosen at least two decades above the main lowpass filter, otherwise it will have an undesirable effect on the envelope response. For single ended 5V designs, the gain should be limited to about 10 dB to avoid amplifier saturation due to carrier ripple and data modulation. FIGURE 7: HV-Env C1 S1 R2 R1 C2 Output + R3  2004 Microchip Technology Inc. DS00912A-page 5 AN912 THE LOW PASS FILTER STAGE The output signal from the decoupling stage consists of the 125 kHz carrier ripple and the modulated data signal, if one ignores the dynamic response signal. The carrier ripple is about 300 mV peak-to-peak. The data is 4 mV peak-to-peak with 6 dB of gain of the decoupler and a cutoff frequency at about 10 kHz. The aim of the low-pass filter stage is to amplify the data signal at 2.5 kHz and to filter out the carrier ripple in the most effective manner. The three most common active filter topologies used are the Chebyshev, Butterworth and Bessel filters. The Chebyshev filter has the steepest transition from pass band to stop band, but has ripple in the pass band. The Butterworth filters have the flattest pass band response, but does not have such a steep transition as the Chebyshev. The Bessel filter has a linear phase response with a smooth transition from pass to stop band. It seems the Chebyshev filter would best be suited for this application, but the frequency response does not tell the whole story. The data signal is amplitude modulated and the tank has steep transient response dynamics. As a result, the filter should have a stable and flat transient response. The Chebyshev filter has a very sharp frequency cutoff response, but has the worst transient response of the three filter topologies. The Chebyshev filter also has an underdamped step response with overshoot and ringing. The Butterworth filter has a better transient response, but still some overshoot. The Bessel filter has the worst response from a frequency perspective, but has the best transient response as a result of its linear phase characteristics. There are of course other active filter topologies such as elliptical, state variable, biquad and more, but a Bessel filter has adequate performance for the application. The data signal, in this example, has maximum modulation frequency of 2.5 kHz or a TE of 200 µs. A Bessel filter, with a cutoff frequency of 1/(2.2TE) = 2.27 kHz, would be ideal from a noise rejection point of view, but a 2.5 kHz cutoff was chosen to minimize symbol overlap. The target is to design a filter with sufficient performance using a single operational amplifier in order to reduce the system cost. A dual operational amplifier can then be used because the decoupling stage also uses an amplifier. A third order Bessel filter can now be implemented with the remaining amplifier. The filter gain is the final aspect to specifying the Bessel filter. Using Microchip's FilterLab® program, one can get the response for a unity gain – 2.5 kHz, 3d order Bessel filter. At 125 kHz, the filter has 93 dB of attenuation and the input ripple amplitude is 300 mV peak-to-peak. Assuming the filter should have an output ripple of no more then 1 mV peak-to-peak with 12 dB of headroom for noise, coupled through the supply line, then one needs at least 62 dB of attenuation. This leaves 31 dB of allowable gain from the third order filter. For the design, a gain of 20 or 26 dB was chosen, leaving some additional headroom for ripple rejection. The 3d order low-pass Bessel filter is shown in Figure 8 and has a Fc = 2.5 kHz and 26 dB of gain. Please note that the circuit shown in Figure 8 has a fairly high output impedance at the data rate, but the output of the filter will be driving a high-impedance load, and this is therefore acceptable. FIGURE 8: 78.7k Input 3.92k 16.5k 10 nF 150 pF + Output 4.87k 10 nF DS00912A-page 6  2004 Microchip Technology Inc. AN912 DATA SLICER The data slicer is essentially a comparator with some input hysteresis voltage to reduce the influence of noise. The overall system gain of the decoupler and the low-pass filter, at 2.5 kHz, is about 29 dB or a factor of 28, and the system should be able to detect the 2 mV data signal. The headroom between the hysteresis and data signal was chosen to be about 9 dB or a factor of 2.8. This means that the minimum input voltage to overcome the data slicer hysteresis is about 700 µV. This translates to 20 mV of hysteresis for the data slicer. Most comparators have some deliberate hysteresis to improve noise stability and this amount should be extracted from the required hysteresis when calculating the amount of required feedback. Figure 9 shows a typical hysteresis circuit and Equation 5 can be used to calculate the amount of hysteresis for a single-ended circuit. AN EXAMPLE SYSTEM A complete circuit with layout, based on the foregoing design study, is shown in this section. The circuit diagram is shown in Figure 11. The top and bottom layout for the printed circuit board is shown in Figure 12. The PIC16F648A was chosen for the application, it has two comparators, a USART, EEPROM and 4k of Flash program memory. The PIC16F648A can be substituted with its smaller program memory equivalents, the PIC16F627A or PIC16F628A. The filter examples have been converted to operate from a single 5 VDC supply. The 2.5 VDC virtual ground is provided by the voltage divider consisting of R23 and R24, shown in Figure 11. The Reference voltage does not have to be actively buffered, it is lightly loaded. A 0.1 µF decoupling capacitor C10 is sufficient for noise reduction. A TC4422 FET driver, U1, drives the resonant tank consisting of L1 and C2. The tank generates a strong magnetic field and the voltage at the test pin TP1 can reach 320V peak-to-peak. The main antenna, L1, is an air-cored inductor with a 25 mm radius and 41 turns of 26-gage wire, and has a 162 µH inductance. The inductor L2 and capacitors C3 and C4 are not populated and are added to the printed circuit board to test alternative antennas. The peak detector consists of D1, C5, R1, and R2, and is connected to the decoupling stage via C6. The RC time constant of the decoupling tank is set by C6 and R4 to 177 µs, which is substantially longer than the minimum filter requirement of 64 µs. Resistor R3 is used to change the decoupler's time constant to 11 µs by changing RB7 from a highimpedance input to an output. The decoupler buffer, U2:A, has a gain of 6 dB and a low pass cutoff frequency at 9.8 kHz, set by R5 and C8. The R22 resistor is used to ensure the proper DC bias of the stage, but does not have a significant effect on the overall sensitivity. The output of the decoupler is connected to the input of the low pass Bessel filter and one of the PIC16F648's comparators. The remaining op amp, U2:B, is used for the Bessel filter. U2 is a dual MCP6002 op amp that has a GBWP of 1 MHz. The filter components should have better tolerances than the high voltage components and 1% resistors. The 5% NP0 capacitors are recommended. EQUATION 5: VHYST = R1 R2 VDD FIGURE 9: VREF Input R1 R2 + Output For example, if a comparator with 10 mV of offset and hysteresis is used, then an additional 10 mV of hysteresis should be added. The resistor R2 is calculated to be 5 MΩ for a VDD of 5 VDC and R1 = 10 kΩ.  2004 Microchip Technology Inc. DS00912A-page 7 AN912 The PIC16F648A has various comparator options. Figure 10 shows the topology that was chosen for this application. The main filter output signal “ENV_IN” is connected to comparator C1 via RA0. Resistor R10 was placed in series with the output of the filter to have 10 kΩ impedance. Together with the 4.99 MΩ resistor, R11 adds an additional 10 mV of hysteresis. The comparator has a combined offset and hysteresis of 10 mV, in the worst case, making for a total of 20 mV of hysteresis, in the worst case, and about 15 mV on average. It should be noted that the output of comparator C1 has to be inverted by setting bit C1INV, in the CMCON register. The output inversion is needed to result in positive feedback, via R11, as is shown in Figure 9. At first glance, it seems as if R10 can be removed and R11 changed to a 2.43 MΩ resistor, but the capacitor C12 will cause delay and that can lead to instability. There are additional aspects around the decoupler that need to be explained for the system as it is implemented. The port pin RB7 is essentially an open circuit when it is configured as an input and the input voltage is between VDD (5V) and ground. All the general purpose I/O pins have internal ESD protection diodes that become conductive when a pin voltage is forced outside the VDD to ground range. This has the effect that the RC time constant for the decoupling stage is reduced to 11 µs from 177 µs whenever the “BIAS” signal is about 0.6V above VDD, or below ground even if RB7 is configured as an input. The addition of R3 works well, but keep in mind, the stable DC voltage for signal “A”, shown in Figure 11, is 2.5 VDC and the signal “BIAS” is either 5V or ground. One can implement one of two approaches to correctly bias the signal at point “A”. The first solution is to toggle RB7 between high and low with a 50% duty cycle at 20 kHz or more. This is equivalent to connecting the “BIAS” signal to the desired 2.5 VDC. This is only done for a short period after the tank is turned on or off, to force the decoupler to stabilize faster than it would with just R4. The second approach is to force the signal “A” in the required direction. The voltage at “A” will go above VDD if the tank is turned on after it has been turned off for some time. The “BIAS” signal can be grounded during the turn-on transient period until the voltage at point “A” reaches the desired 2.5 VDC or VREF. By monitoring either of the comparator output signals, it is possible to detect when the voltage at point A goes through VREF. Pin RB7 can be turned into an input as soon as the cross over is detected resulting in a decoupler RC time constant of 177 µs. The filters introduce delay that cause some overshoot of the voltage at point “A”. The overshoot can be resolved by allowing some additional stabilizing time with R4, before LF communication is interpreted as data. FIGURE 10: Two common Reference Comparators with Outputs CM2:CM0 = 110 RA0/AN0 RA3/AN3/CMP1 A D VIN+ C1 + VIN- VIN+ C2 + VIN- C1VOUT RA1/AN1 RA2/AN2/VREF RA4/T0CKI/CMP2 A A C2VOUT Open Drain DS00912A-page 8  2004 Microchip Technology Inc. AN912 SYSTEM MODIFICATIONS The system can be modified to better suit the user's requirements. The first aspect is to change the Bessel filter for a different LF Talkback TE. The rule of thumb is to set the filter's 3 dB cutoff frequency to Fc = 1/(2*TE). The new values for the Bessel filter, with a 400 µs TE, is given in Table 1. INCREASED DATA SENSITIVITY Increasing the system’s sensitivity to the modulated data signal can increase the LF Talkback range. A solution has been partly described in the previous section; increase TE from 200 µs to 400 µs and then increase the gain by up to 18 dB. The component values for a system with a 400 µs TE, or a center frequency of 1.25 kHz, and a gain of 100, or a 14 dB increase, is described in Table 2 below. This approach decreases the dynamic range that may or may not be used depending on how well the transponder loads the resonant tank. TABLE 1: R6 = 3.57k R10 = 5.62k R7 = 15.0k C9 = C12 = 22 nF R8 = 71.5k C11 = 330 pF R9 = 4.42k In addition to changing the filter cutoff frequency for a TE of 400 µs, it is possible to increase the gain up to 18 dB and still maintain the carrier rejection chosen. It is also possible to increase C6 to a 4.7 nF capacitor, but please note that this will increase the transient response period. Increasing C6 will not have a dramatic influence on the overall system performance and it is not recommended. TABLE 2: R6 = 2.26k R10 = 5.62k R7 = 10.5k C9 = 33 nF R8 = 226k R9 = 4.42k C11 = 100 pF C12 = 22 nF LONGER TRANSIENT STABILIZING PERIOD The example system was designed with the requirement that LF Talkback communications should be able to start 200 µS after the resonant tank has stabilized. The tank itself takes 100 µS to 400 µs to stabilize sufficiently, depending on the drive mechanism. The example circuit should be able to start LF Talkback communications with 2 mV of data modulation after 350 µs to 450 µs, from when the tank is turned. The exact period depends on the residual charge in the peak detector from previous transmissions. The system can be simplified and improved if the system allows for a longer transient stabilizing period before LF Talkback communications are initiated. The peak detector capacitor, C5, can be increased proportionally to the longer stabilizing period, but not by more than a factor of about 3, otherwise it can influence data modulation sensitivity. R1 and R2 tank resistors should be reduced if C5 is increased, but not proportionally, it will effect sensitivity. The combined value for R1 and R2 should be no less then 4 MΩ. Capacitor C6 can also be increased, but it will not have a dramatic performance increase. The biggest advantage of a longer transient stabilizing period is that bias resistor R3 can be increased. Increasing the value of R3 will result in a slower change in the signal at point “A”, which means the tank can be controlled more accurately during the transient period. Another solution is to remove resistor R11 to get the maximum sensitivity from the comparator, but this will also increase noise in the data. Another quick solution is to increase the gain of the decoupler buffer by up to 10 dB and lower the decoupler cutoff frequency by about half the gain increase ratio. The existing design makes use of a 3d order Bessel filter. For improved noise reduction, increase the order of the filter and add more gain per stage. This would typically be done if a TE of 200 µs or 100 µs, is desired with more sensitivity than can be reliably obtained with the example system. DRIVE SYSTEM The example circuit uses a half-bridge driver based on the TC range of FET drivers from Microchip. To increase the transient response period of the tank, start the tank in Full-bridge mode until the desired tank amplitude is reached and the tank oscillation is maintained in Half-bridge mode. This method is described in AN232 Low Frequency Magnetic Transmitter Design. CONCLUSION This LF Talkback Design application note can be used to implement a cost-effective system to be used in RFID, passive keyless entry and other bidirectional transponder based technologies. The example circuit can be used as a basis for further hardware and firmware development to suit the user's requirements.  2004 Microchip Technology Inc. DS00912A-page 9 FIGURE 11: AN912 DS00912A-page 10 HIGH-VOLTAGE SECTION NOTES: Unless otherwise specified; Resistance values are in ohms. Resistors are 1% tolerance. Capacitance values are in uF. SMT resistors are size 1206 and 1/8W. APPENDIX A: +5V C1 0.1 uF 3 VDD L1 D1 2.2 nF A 500V A C2 A L2 DO5022P R2 4.99M 10 nF 400V P3476-ND C3 .200LS C4 .200LS C5 1.0 nF 500V 1412PH-ND R1 4.99M A UF1005 10-00189 TP1 C6 TP2 6 U1 TC4422 PWM 1 VDD Device names and numbers shown here are for reference only and may differ from the actual number. Items labeled with A are unpopulated. Items labeled with B are socketed and populated. IN OUT 5 GND GND 4 LF BASE STATION SCHEMATICS 2 C8 100 pF TP7 COARSE_ENV_OUT +5V R8 78.7K TP6 C11 150 pF R11 4.99M R6 R7 16.5K C9 +5V 10 nF VREF 5 +IN OUT 3.92K 6 -IN 7 R9 4.87K U2:B MCP6002/SN C10 0.1 uF VREF R24 49.9K C12 10 nF TP5 R10 5.11K ENV_IN ENV_OUT R5 162K R3 A C7 0.1 uF COARSE_ENV_IN TP4 OUT 3 +IN U2:A VSS MCP6002/SN 4 1 BIAS 4.99K 8 2 -IN VDD TP3 R4 80.6K R22 4.99M VREF VREF R23 49.9K  2004 Microchip Technology Inc. FIGURE 12: +5V VREF ENV_OUT COARSE_ENV_OUT MCLR 5 VSS RBO/INT RB1 RB6 RB5 RB4 RF_IN 10 RB6 11 RB2 RB3 PIC16F648A/P 6 7 8 9 R20 4.7K RESET SW2 1 2 C19 1 uF MOM-NO 3 470 MCLR 4 R21 1 2 MOM-NO SW1 4 3 R18 470 RB0 RBO RX TX RB5 RB6 BIAS +5V +5V 12 RB5 C22 20 pF BIAS C24 0.1 uF RB7 13 VDD 14 6 7 8 9 +5V RB0 RX TX PWM J1 U5 MAX232CPE 4 MCLR OSC2 OSC2 OSC2 15 OSC1 OSC1 Y1 20.0 MHz 311-1153-1-ND 1 uF C21 3 RA4/TO OSC1 16 2 RA3 RA0 ENV_IN 1 RA2 RA1 18 17 COARSE_ENV_IN B U3 C20 1 uF C23 20 pF 2 14 7 4 13 5 DE-9S (FEM) 8 4 5 3 1 2 V+ R14 TX VCC 16 T1OUT T2OUT R1IN R2IN C2+ C2- 15 GND  2004 Microchip Technology Inc. A J3 1 R19 4.7K 2 3 4 5 6 +5V 1 A1 C16 0.1 uF 6.8" WIRE VR1 78L05 3 IN OUT GND 2 R12 270 1 10 AF_+VCC 11 AF_GND 12 R13 D2 POWER GRN RF_IN 1K NC AF_+VCC 13 TP 14 DATA_OUT 15 AF_+VCC RR8 433.92 MHz 1K 11 T1IN 10 T2IN R15 RX 12 R1OUT 1K 9 R2OUT LF BASE STATION (Continued) 1 C1+ 3 C1- C18 1 uF 6 V- C17 1 uF U4 RF_+VCC 2 1 3 RF_GND DATA_IN R16 270 ANTENNA 7 RF_GND D3 RED D4 GRN R17 270 +5V +5V C15 10 uF 6.3V +12V +5V J2 4P-DIN 4 2 C14 47 uF 10V P11180-ND 3 RB5 RB6 1 5 6 7 C13 560 uF 25V P11220-ND DS00912A-page 11 POWER DYNAMIC MDC-034 AN912 AN912 FIGURE 13: BOTTOM SIDE 05-01 XXXX REV. A BOTTOM SIDE FIGURE 14: TOP SIDE 05-01 XXXX REV. A TOP SIDE DS00912A-page 12  2004 Microchip Technology Inc. AN912 FIGURE 15: TOP MASK D2 J2 R12 U5 R14 C17 J1 R15 C13 C21 C20 C22 C23 C15 C16 R13 D3 R16 D4 R17 U4 A1 RB0 U1 HIGH VOLTAGE SECTION D1 D5 C14 VR1 C2 POWER FILTER R11 R10 C18 C19 RS232 C24 L1 TP7 TP6 TP5 TP1 C3 C4 C5 D1 R1 C6 TP2 R2 C10 R22 L2 RESET R20 R21 R19 R18 U3 R9 C11 R8 R7 R6 R5 TP4 C8 TP3 C12 C7 R23 R24 U2 C9 J3 R4 R3  2004 Microchip Technology Inc. RB0 RB1 RB2 RB5 RB6 RB7 05-01 XXXX REV. A TOP MASK DS00912A-page 13 AN912 NOTES: DS00912A-page 14  2004 Microchip Technology Inc. Note the following details of the code protection feature on Microchip devices: • • Microchip products meet the specification contained in their particular Microchip Data Sheet. Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions. There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip's Data Sheets. Most likely, the person doing so is engaged in theft of intellectual property. Microchip is willing to work with the customer who is concerned about the integrity of their code. Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not mean that we are guaranteeing the product as “unbreakable.” • • • Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act. Information contained in this publication regarding device applications and the like is intended through suggestion only and may be superseded by updates. It is your responsibility to ensure that your application meets with your specifications. No representation or warranty is given and no liability is assumed by Microchip Technology Incorporated with respect to the accuracy or use of such information, or infringement of patents or other intellectual property rights arising from such use or otherwise. Use of Microchip’s products as critical components in life support systems is not authorized except with express written approval by Microchip. No licenses are conveyed, implicitly or otherwise, under any intellectual property rights. Trademarks The Microchip name and logo, the Microchip logo, Accuron, dsPIC, KEELOQ, MPLAB, PIC, PICmicro, PICSTART, PRO MATE and PowerSmart are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. AmpLab, FilterLab, microID, MXDEV, MXLAB, PICMASTER, SEEVAL, SmartShunt and The Embedded Control Solutions Company are registered trademarks of Microchip Technology Incorporated in the U.S.A. Application Maestro, dsPICDEM, dsPICDEM.net, dsPICworks, ECAN, ECONOMONITOR, FanSense, FlexROM, fuzzyLAB, In-Circuit Serial Programming, ICSP, ICEPIC, microPort, Migratable Memory, MPASM, MPLIB, MPLINK, MPSIM, PICkit, PICDEM, PICDEM.net, PICtail, PowerCal, PowerInfo, PowerMate, PowerTool, rfLAB, rfPIC, Select Mode, SmartSensor, SmartTel and Total Endurance are trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. Serialized Quick Turn Programming (SQTP) is a service mark of Microchip Technology Incorporated in the U.S.A. All other trademarks mentioned herein are property of their respective companies. © 2004, Microchip Technology Incorporated, Printed in the U.S.A., All Rights Reserved. Printed on recycled paper. Microchip received ISO/TS-16949:2002 quality system certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona and Mountain View, California in October 2003. The Company’s quality system processes and procedures are for its PICmicro® 8-bit MCUs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified.  2004 Microchip Technology Inc. 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De Biesbosch 14 NL-5152 SC Drunen, Netherlands Tel: 31-416-690399 Fax: 31-416-690340 ASIA/PACIFIC Australia Suite 22, 41 Rawson Street Epping 2121, NSW Australia Tel: 61-2-9868-6733 Fax: 61-2-9868-6755 United Kingdom 505 Eskdale Road Winnersh Triangle Wokingham Berkshire, England RG41 5TU Tel: 44-118-921-5869 Fax: 44-118-921-5820 01/26/04 DS00912A-page 16  2004 Microchip Technology Inc.