TC1043

TC1043 Linear Building Block – Low Power Voltage Reference with Dual Op Amp, Dual Comparator, and Shutdown Mode Features • Combines Two Op Amps, Two Comparators and a Voltage Reference in a Single Package • Optimized for Single-Supply Operation • Small Package: 16-Pin QSOP • Ultra Low Input Bias Current: Less than 100pA • Low Quiescent Current: Operating 16µA (Typ.) Shutdown 6µA (Typ.) • Rail-to-Rail Inputs and Outputs • Operates Down to VDD = 1.8V • Reference and One Comparator Remain Active in Shutdown to Provide Supervisory Functions General Description The TC1043 is a mixed-function device combining two general purpose op amps, two general purpose comparators, and a voltage reference in a single 16-Pin package. This increased integration allows the user to replace two or three packages, saving space, lowering supply current, and increasing system performance. A shutdown input, SHDN, disables the op amps and one of the comparators, placing their outputs in a high-impedance state. The reference and one comparator stay active in shutdown mode. Standby power consumption is typically 6µA. Both the op amps and comparators have rail-to-rail inputs and outputs which allows operation from low supply voltages with large input and output signal swings. Packaged in a 16-Pin QSOP, the TC1043 is ideal for applications requiring high integration, small size and low power. Applications • • • • Power Management Circuits Battery Operated Equipment Consumer Products Replacements for Discrete Components Device Selection Table Part Number TC1043CEQR Package 16-Pin QSOP Temperature Range -40°C to +85°C Package Type 16-Pin QSOP A1IN+ 1 A1IN– 2 TC1043_EQR A2IN+ 3 A2IN– 4 C1OUT 5 C2OUT 6 SHDN 7 VSS 8 16 VDD 15 A1OUT 14 A2OUT 13 C1IN+ 12 C1IN– 11 C2IN+ 10 C2IN– 9 VREF 2002 Microchip Technology Inc. DS21347B-page 1 © TC1043 1.0 ELECTRICAL CHARACTERISTICS *Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only and functional operation of the device at these or any other conditions above those indicated in the operation sections of the specifications is not implied. Exposure to Absolute Maximum Rating conditions for extended periods my affect device reliability. ABSOLUTE MAXIMUM RATINGS* Supply Voltage ......................................................6.0V Voltage on Any Pin ........... (VSS – 0.3V) to (VDD +0.3V) Junction Temperature....................................... +150°C Operating Temperature Range............. -40°C to +85°C Storage Temperature Range .............. -55°C to +150°C TC1043 ELECTRICAL SPECIFICATIONS Electrical Characteristics: Typical values apply at 25°C and VDD = 3V. Minimum and maximum values apply for TA = -40° to + 85°C, and VDD = 1.8V to 5.5V, unless otherwise specified. Symbol VDD IQ ISHDN Parameter Supply Voltage Supply Current Operating Supply Current, Shutdown Min. 1.8 — — Typ — 16 6 Max 5.5 30 10 Units V µA µA All outputs unloaded, SHDN = VDD CMPTR2 and VREF Outputs unloaded, SHDN = VSS Test Conditions Shutdown Input VIH VIL ISI Op Amps TSEL TDESEL ROUT(SD) COUT(SD) AVOL VICMR VOS IB VOS(DRIFT) GBWP Select Time Deselect Time Output Resistance in Shutdown Output Capacitance in Shutdown Large Signal Voltage Gain Common Mode Input Voltage Range Input Offset Voltage Input Bias Current Input Offset Voltage Drift Gain-Bandwidth Product -100 — — — — 20 — — VSS - 0.2 15 100 — — 100 — ±100 ±0.3 50 ±4 90 — — — 6 — VDD + 0.2 ±500 ±1.5 100 — — µsec nsec MΩ pF V/mV V µV mV pA µV/°C kHz VDD = 3V, VCM = 1.5V, TA = 25°C, TA = -40°C to 85°C TA = 25°C, VCM = VDD to VSS VDD = 3V, VCM = 1.5V VDD = 1.8V to 5.5V VO = VDD to VSS CL = 100pF RL = 1MΩ to GND Gain = 1 VIN = VSS to VDD RL = 10kΩ TA = 25°C, VDD = 5V VCM = VDD to VSS TA = 25°C, VCM = VSS VDD = VDD to VSS IN+ = VDD, IN- = VSS Output Shorted to VSS VDD = 1.8V, Gain = 1 IN+ = VSS, IN- = VDD Output Shorted to VDD VDD = 1.8V, Gain = 1 0.1Hz to 10Hz (VOUT from SHDN = VIH) RL = 10kΩ to VSS (VOUT from SHDN = VIL) RL = 10kΩ to VSS SHDN = VSS SHDN = VSS RL = 10kΩ, VDD = 5V Input High Threshold Input Low Threshold Shutdown Input Current 80% VDD — — — — — — 20% VDD ±100 V V nA SR Slew Rate — 35 — mV/ µsec V dB dB VOUT CMRR PSRR Output Signal Swing Common Mode Rejection Ratio Power Supply Rejection Ratio VSS + 0.05 70 80 — — — VSS - 0.05 — — ISRC Output Source Current 3 — — mA ISINK En en Output Sink Current Input Noise Voltage Input Noise Voltage Density 4 — — — 10 125 — — — mA µVPP nV/√Hz 1kHz © DS21347B-page 2 2002 Microchip Technology Inc. TC1043 TC1043 ELECTRICAL SPECIFICATIONS (CONTINUED) Electrical Characteristics: Typical values apply at 25°C and VDD = 3V. Minimum and maximum values apply for TA = - 40° to + 85°C, and VDD = 1.8V to 5.5V, unless otherwise specified. Symbol Comparators ROUT(SD) COUT(SD) TSEL TDESEL VICMR VOS IB VOH VOL CMRR PSRR Output Resistance in Shutdown Output Capacitance in Shutdown Select Time (For Valid Output) Deselect Time Common Mode Input Voltage Range Input Offset Voltage 20 — — — VSS – 0.2 –5 –5 — VDD – 0.3 — 66 60 — — 20 500 — — — 5 — — VDD + 0.2 +5 +5 ±100 — 0.3 — — MΩ pF µsec nsec V mV VDD = 3V, VCM = 1.5V TA = 25°C TA = -40°C to 85°C TA = 25°C, IN+, IN- = VDD to VSS RL = 10kΩ to VSS RL = 10kΩ to VDD TA = 25°C, VDD = 5V VCM = VDD to VSS TA = 25°C, VCM = 1.2V VDD = 1.8V to 5V IN+ = VDD, IN- = VSS Output Shorted to VSS VDD = 1.8V IN+ = VSS, IN- = VDD, Output Shorted to VDD VDD = 1.8V 100mV Overdrive, CL = 100pF 10mV Overdrive, CL = 100pF SHDN = VSS SHDN = VSS VOUT from SHDN = VIH RL =10kΩ to VSS VOUT from SHDN = VIL RL =10kΩ to VSS Parameter Min. Typ Max Units Test Conditions Input Bias Current Output High Voltage Output Low Voltage Common Mode Rejection Ratio Power Supply Rejection Ratio — — — — — pA V V dB dB ISRC Output Source Current 1 — — mA ISINK tPD1 tPD2 VREF IREF(SOURCE) IREF(SINK) CL(REF) NVREF Output Sink Current Response Time Response Time Reference Voltage Source Current Sink Current Load Capacitance Voltage Noise Noise Density 2 — — 1.176 50 50 — — — — 4 6 1.200 — — — 20 1.0 — — — 1.224 — — 100 — — mA µsec µsec V µA µA pF Voltage Reference µVRMS 100Hz to 100kHz µV/√Hz 1kHz 2002 Microchip Technology Inc. DS21347B-page 3 © TC1043 2.0 PIN DESCRIPTION The descriptions of the pins are listed in Table 2-1. TABLE 2-1: Pin Number 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 PIN FUNCTION TABLE Symbol A1IN+ A1INA2IN+ A2INC1OUT C2OUT SHDN VSS VREF C2INC2IN+ C1INC1IN+ A2OUT A1OUT VDD Op Amp Non-Inverting Input Op Amp Inverting Input Op Amp Non-Inverting Input Op Amp Inverting Input Comparator Output Comparator Output Shutdown Input Negative Power Supply Voltage Reference Output Comparator Inverting Input Comparator Non-Inverting Input Comparator Inverting Input Comparator Non-Inverting Input Op Amp Output Op Amp Output Positive Power Supply Description © DS21347B-page 4 2002 Microchip Technology Inc. TC1043 3.0 DETAILED DESCRIPTION 3.4 Shutdown Input The TC1043 is one of a series of very low power, linear building block products targeted at low voltage, single supply applications. The TC1043 minimum operating voltage is 1.8V and typical supply current is only 20µA (fully enabled). It combines two comparators, two op amps and a voltage reference in a single package. A shutdown mode is incorporated for easy adaptation to system power management schemes. During shutdown, all but one comparator and the voltage reference are disabled (i.e. powered down with their respective outputs at high impedance). The “still awake” comparator and voltage reference can be used as a wake-up timer, power supply monitor, LDO controller or other continuous duty circuit function. SHDN at VIL disables both op amps and one comparator. The SHDN input cannot be allowed to float. When not used, connect it to VDD. The disabled comparator’s output and the two disabled op amp outputs are in a high impedance state when shutdown is active. The disabled comparator’s inputs and the two disabled op amp inputs can be driven from rail-to-rail by an external voltage when the TC1043 is in shutdown. No latch-up will occur when the device is driven to its enabled state when SHDN is set to VIH. 3.1 Comparators The TC1043 contains two comparators. The comparators input range extends beyond both supply voltages by 200mV and the outputs will swing to within several millivolts of the supplies, depending on the load current being driven. The comparators exhibit a propagation delay and supply current which are largely independent of supply voltage. The low input bias current and offset voltage make them suitable for high impedance precision applications. Comparator CMPTR1 is disabled during shutdown and has a high impedance output. Comparator CMPTR2 remains active. 3.2 Operational Amplifiers The TC1043 contains two rail-to-rail op amps. The amplifiers’ input range extends beyond both supplies by 200mV and the outputs will swing to within several millivolts of the supplies depending on the load current being driven. The amplifier design is such that large signal gain, slew rate and bandwidth are largely independent of supply voltage. The low input bias current and offset voltage of the TC1043 make it suitable for precision applications. Both op amps are disabled during shutdown and have high output impedance. 3.3 Voltage Reference A 2.0% tolerance, internally biased, 1.20V bandgap voltage reference is included in the TC1043. It has a push-pull output capable of sourcing and sinking at least 50µA. The voltage reference remains fully enabled during shutdown. 2002 Microchip Technology Inc. DS21347B-page 5 © TC1043 4.0 TYPICAL APPLICATIONS 4.3 Dual LDO with Shutdown The TC1043 lends itself to a wide variety of applications, particularly in battery powered systems. It typically finds application in power management, processor supervisory, and interface circuitry. Figure 4-3 shows a portion of a TC1043 configured as a dual low dropout regulator with shutdown. AMP1 and AMP2 are independent error amplifiers that use VR as a reference. Resistors RA1, RB1, RA2 and RB2 set the feedback around the amplifiers and therefore determine the output voltage settings (please see equation in the figure). RA1, RB1, RA2 and RB2 can have large ohmic values (i.e. 100’s of kΩ) to minimize supply current. Using the 2N2222 output transistors as shown, these regulators exhibit low dropout operation. For example, with VOUT = 3.0V, the typical dropout voltage is only 50mV at an output current of 50mA. The unused comparators can be used in conjunction with this circuit as power-on reset or low voltage detectors for a complete LDO solution at a very low installed cost. 4.1 Wake-Up Timer Many microcontrollers have a low power “sleep” mode that significantly reduces their supply current. Typically, the microcontroller is placed in this mode via a software instruction, and returns to a fully enabled state upon reception of an external signal (“wake-up”). The wakeup signal is usually supplied by a hardware timer. Most system applications demand that this timer have a long duration (typically seconds or minutes), and consume as little supply current as possible. The circuit shown in Figure 4-1 is a wake-up timer made from comparator CMPTR2. (CMPTR2 is used because the wake-up timer must operate when SHDN is active.) Capacitor C1 charges through R1 until a voltage equal to VR is reached, at which point the WAKEUP is driven active. Upon wake-up, the microcontroller resets the timer by forcing a logic low on a dedicated, open drain I/O port pin. This discharges C1 through R4 (the value of R4 is chosen to limit the maximum current sunk by the I/O port pin). With a 3V supply, the circuit as shown consumes typically 6µA and furnishes a nominal timer duration of 25 seconds. 4.4 External Hysteresis Hysteresis can be set externally with two resistors using positive feedback techniques (see Figure 4-3). The design procedure for setting external comparator hysteresis is as follows: 1. Choose the feedback resistor RC. Since the input bias current of the comparator is at most 100pA, the current through RC can be set to 100nA (i.e. 1000 times the input bias current) and retain excellent accuracy. The current through RC at the comparator’s trip point is VR / RC where VR is a stable reference voltage. 4.2 Precision Battery Monitor Figure 4-2 is a precision battery low/battery dead monitoring circuit. Typically, the battery low output warns the user that a battery dead condition is imminent. Battery dead typically initiates a forced shutdown to prevent operation at low internal supply voltages (which can cause unstable system operation). The circuit of Figure 4-2 uses a single TC1043 (one op amp is unused) and only six external resistors. AMP 1 is a simple buffer, while CMPTR1 and CMPTR2 provide precision voltage detection using VR as a reference. Resistors R2 and R4 set the detection threshold for BATTLOW, while resistors R1 and R3 set the detection threshold for BATTFAIL. The component values shown assert BATTLOW at 2.2V (typical) and BATTFAIL at 2.0V (typical). Total current consumed by this circuit is typically 22µA at 3V. Resistors R5 and R6 provide hysteresis for comparators CMPTR1 and CMPTR2 respectively. FIGURE 4-1: R4 WAKE-UP TIMER Microcontroller I/O* VDD R1 5M C1 10µF VDD – + CMPTR2 WAKE-UP VR TC1043 *Open Drain Port Pin 2. 3. Determine the hysteresis voltage (VHY) between the upper and lower thresholds. Calculate RA as follows: © DS21347B-page 6 2002 Microchip Technology Inc. TC1043 EQUATION 4-1:         V HY R A = R C ---------V DD 4. 5.   Choose the rising threshold voltage for VSRC (VTHR). Calculate RB as follows: the voltage required at the input to raise the voltage at node A higher than the voltage at node B, and is set by the resistive divider R4 and R10 and the impedance network composed of R1, R2 and R3. When the one shot has been triggered, the output of CMPTR2 is high, causing the reference voltage at the non-inverting input of CMPTR1 to go to VDD. This prevents any additional input pulses from disturbing the circuit until the output pulse has timed out. The value of the timing capacitor C1 must be small enough to allow CMPTR1 to discharge C1 to a diode voltage before the feedback signal from CMPTR2 (through R10) switches CMPTR1 to its high state and allows C1 to start an exponential charge through R5. Proper circuit action depends upon rapidly discharging C1 through the voltage set by R6, R9 and D2 to a final voltage of a small diode drop. Two propagation delays after the voltage on C1 drops below the level on the non-inverting input of CMPTR2, the output of CMPTR1 switches to the positive rail and begins to charge C1 through R5. The time delay which sets the output pulse width results from C1 charging to the reference voltage set by R6, R9 and D2, plus four comparator propagation delays. When the voltage across C1 charges beyond the reference, the output pulse returns to ground and the input is again ready to accept a trigger signal. EQUATION 4-2: 1 R B = ---------------------------------------------------------V THR 1 1 -------------------- – ------ – ------VR × RA RA RC 6. Verify the threshold voltages with these formulas: VSRC rising:     EQUATION 4-3: 1 1 1 V TH R = ( V R ) ( R A ) ------ + ------- + ------RA RB RC VSRC falling:       EQUATION 4-4: ( RA × VD D) V THF = V THR – ----------------------------RC 4.7 Oscillators and Pulse Width Modulators 4.5 32.768kHz ‘Time Of Day Clock’ Crystal Controlled Oscillator A very stable oscillator driver can be designed by using a crystal resonator as the feedback element. Figure 45 shows a typical application circuit using this technique to develop a clock driver for a Time-Of-Day (TOD) clock chip. The value of R A and RB determines the DC voltage level at which the comparator trips; in this case one-half of VDD. The RC time constant of RC and CA should be set several times greater than the crystal oscillator’s period, which will ensure a 50% duty cycle by maintaining a DC voltage at the inverting comparator input equal to the absolute average of the output signal. Microchip’s linear building block comparators adapt well to oscillator applications for low frequencies (less than 100kHz). Figure 4-7 shows a symmetrical square wave generator using a minimum number of components. The output is set by the RC time constant of R4 and C1, and the total hysteresis of the loop is set by R1, R2 and R3. The maximum frequency of the oscillator is limited only by the large signal propagation delay of the comparator in addition to any capacitive loading at the output which degrades the slew rate. To analyze this circuit, assume that the output is initially high. For this to occur, the voltage at the inverting input must be less than the voltage at the non-inverting input. Therefore, capacitor C1 is discharged. The voltage at the non-inverting input (VH) is: 4.6 Non-Retriggerable One Shot Multivibrator EQUATION 4-5: R2 ( V DD ) V H = -------------------------------------------[ R2 + ( R1 || R3 ) ] where, if R1 = R2 = R3, then: Using two comparators, a non-retriggerable, one shot multi-vibrator can be designed using the circuit configuration of Figure 4-6. A key feature of this design is that the pulse width is independent of the magnitude of the supply voltage because the charging voltage and the intercept voltage are a fixed percentage of V DD. In addition, this one shot is capable of pulse width with as much as a 99% duty cycle and exhibits input lockout to ensure that the circuit will not re-trigger before the output pulse has completely timed out. The trigger level is EQUATION 4-6: 2 ( V DD ) V H = ------------------3 2002 Microchip Technology Inc. DS21347B-page 7 © TC1043 Capacitor C1 will charge up through R4. When the voltage at the comparator’s inverting input is equal to VH, the comparator output will switch. With the output at ground potential, the value at the non-inverting input terminal (V L) is reduced by the hysteresis network to a value given by: 4.9 Supervisory Audio Tone (SAT) Filter for Cellular EQUATION 4-7: V DD V L = ---------3 Using the same resistors as before, capacitor C1 must now discharge through R4 toward ground. The output will return to a high state when the voltage across the capacitor has discharged to a value equal to VL. The period of oscillation will be twice the time it takes for the RC circuit to charge up to one-half its final value. The period can be calculated from: Supervisory Audio Tones (SAT) provide a reliable transmission path between cellular subscriber units and base stations. The SAT tone functions much like the current/voltage used in land line telephone systems to indicate that a phone is off the hook. The SAT tone may be one of three frequencies: 5970, 6000 or 6030Hz. A loss of SAT implies that channel conditions are impaired and if SAT is interrupted for more than 5 seconds a cellular call is terminated. Figure 4-10 shows a high Q (30) second order SAT detection bandpass filter using Microchip’s CMOS op amp architecture. This circuit nulls all frequencies except the three SAT tones of interest. EQUATION 4-8: 1 ---------------- = 2 ( 0.694 ) ( R4 ) ( C1 ) FREQ The frequency stability of this circuit should only be a function of the external component tolerances. Figure 4-8 shows the circuit for a pulse width modulator circuit. It is essentially the same as in Figure 4-7 with the addition of an input control voltage. When the input control voltage is equal to one-half V DD, operation is basically the same as described for the free-running oscillator. If the input control voltage is moved above or below one-half VDD, the duty cycle of the output square wave will be altered. This is because the addition of the control voltage at the input has now altered the trip points. The equations for these trip points are shown in Figure 4-8 (see VH and V L). Pulse width sensitivity to the input voltage variations can be increased by reducing the value of R6 from 10kΩ and conversely, sensitivity will be reduced by increasing the value of R6. The values of R1 and C1 can be varied to produce the desired center frequency. 4.8 Voice Band Receive Filter The majority of spectral energy for human voices is found to be in a 2.7kHz frequency band from 300Hz to 3kHz. To properly recover a voice signal in applications such as radios, cellular phones, and voice pagers, a low power bandpass filter that is matched to the human voice spectrum can be implemented using MIcrochip’s CMOS op amps. Figure 4-9 shows a unity gain multipole Butterworth filter with ripple less than 0.15dB in the human voice band. The lower 3dB cut-off frequency is 70Hz (single order response), while the upper cut-off frequency is 3.5kHz (fourth order response). © DS21347B-page 8 2002 Microchip Technology Inc. TC1043 FIGURE 4-2: PRECISION BATTERY MONITOR To System DC/DC Converter R4, 470k, 1% R5, 7.5M VDD R2, 330k, 1% VDD + AMP1 + CMPTR1 – BATTLOW – 3V ALKALINE TC1043 R1, 270k, 1% VR VDD – CMPTR2 BATTFAIL + R6, 7.5M R3, 470k, 1% FIGURE 4-3: DUAL LOW DROPOUT REGULATOR VIN TC1043 VDD VDD SHDN + AMP1 – 2N2222 VOUT1 VR RA1 C1, 1µF + AMP2 – 2N2222 VOUT2 RA2 C2, 1µF RB1 RB2 VOUT = VR x (RA + RB)/RB 2002 Microchip Technology Inc. DS21347B-page 9 © TC1043 FIGURE 4-4: COMPARATOR EXTERNAL HYSTERESIS CONFIGURATION RC VDD VSRC RA + – RB VR VOUT Comparator RB 150k RA 150k FIGURE 4-5: 32.768 KHZ “TIME-OFDAY” CLOCK OSCILLATOR 32.768 kHz VDD TC1043 VDD TC1043 Comparator VOUT RC + – CA 100 pF 1M TPER = 30.52 µsec FIGURE 4-6: NON-RETRIGGERABLE MULTI-VIBRATOR VDD R3 1M R1 In 100k R2 100k In t0 GND R4 1M A – CMPTR1 R5 10M R6 562k C C1 100 pF R7 1M TC1025 – CMPTR2 Out Out VDD GND + B D1 R10 61.9k + R8 C VDD GND R9 243k 10M TC1043 D2 FIGURE 4-7: SQUARE WAVE GENERATOR VDD R1 100k TC1043 R4 VDD – C1 Comparator R2 (VDD) R2 + (R1||R3) (VDD) (R2||R3) R1 + (R2||R3) 1 2(0.694)(R4)(C1) + VH = VL = R2 100k R3 100k FREQ = © DS21347B-page 10 2002 Microchip Technology Inc. TC1043 FIGURE 4-8: PULSE WIDTH MODULATOR VDD R1 100k TC1043 VC R6 10k R4 VH = VDD – C1 + VL = VDD (R1R2R6 + R2R3R6) + VC (R1R2R3) R1R2R6 + R1R3R6 + R2R3R6 + R1R2R3 VDD (R2R3R6) + VC (R1R2R3) R1R2R6 + R1R3R6 + R2R3R6 + R1R2R3 1 2 (0.694) (R4) (C1) FREQ = R2 100k R3 100k Comparator For Square Wave Generation, Select R1 = R2 = R3 V VC = DD 2 FIGURE 4-9: MULTI-POLE BUTTERWORTH VOICE BAND RECEIVE FILTER TC1043 Gain = 0 dB Fch = 3.5kHz -24 dB/Octave Fcl = 70Hz +6 dB/Octave Passband Ripple < 0.15 dB 0.1 µF 22.6k – 22.6k VDD /2 VDD + VOUT 750 pF 6800 pF VIN 21.0k 21.0k 21.0k VDD + 2400 pF 470 pF – Two (2) TC1043 Op Amps 2002 Microchip Technology Inc. DS21347B-page 11 © TC1043 FIGURE 4-10: SECOND ORDER SAT BANDPASS FILTER Gain = 0 dB Q = 30 Q = FC BW (3 dB) 48.7k TC1043 .036 µF FC = 6kHz VIN 24.3k .036 µF – + 11.2 VDD/2 VDD/2 VDD VOUT TC1043 Op Amp © DS21347B-page 12 2002 Microchip Technology Inc. TC1043 5.0 Note: TYPICAL CHARACTERISTICS The graphs and tables provided following this note are a statistical summary based on a limited number of samples and are provided for informational purposes only. The performance characteristics listed herein are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified operating range (e.g., outside specified power supply range) and therefore outside the warranted range. Comparator Propagation Delay vs. Supply Voltage 7 DELAY TO RISING EDGE (µsec) 6 Overdrive = 10mV DELAY TO FALLING EDGE (µsec) TA = 25°C CL = 100pF Comparator Propagation Delay vs. Supply Voltage 7 6 DELAY TO RISING EDGE (µsec) TA = 25°C CL = 100pF Comparator Propagation Delay vs. Temperature 7 Overdrive = 100mV 6 Overdrive = 10mV 5 4 5 4 Overdrive = 100mV Overdrive = 50mV 5 VDD = 5V VDD = 4V 3 2 1.5 2 2.5 3 Overdrive = 50mV 3 2 4 VDD = 2V VDD = 3V 3 1.5 2 2.5 3 3.5 4 4.5 5 5.5 -40 25 SUPPLY VOLTAGE (V) 3.5 4 4.5 5 5.5 85 SUPPLY VOLTAGE (V) TEMPERATURE (°C) Comparator Propagation Delay vs. Temperature 7 DELAY TO FALLING EDGE (µsec) Overdrive = 100mV 6 VDD - VOUT (V) VDD = 5V Comparator Output Swing vs. Output Source Current 2.5 2.0 VDD = 3V VDD = 1.8V TA = 25°C Comparator Output Swing vs. Output Sink Current 2.5 2.0 VOUT - VSS (V) TA = 25°C 5 VDD = 4V VDD = 3V VDD = 2V 1.5 1.0 1.5 1.0 VDD = 3V VDD = 1.8V VDD = 5.5V 4 .5 0 -40 25 TEMPERATURE (°C) 85 0 1 VDD = 5.5V .5 0 3 3 2 4 ISOURCE (mA) 5 6 0 1 2 3 4 5 6 ISINK (mA) 60 50 40 1.240 TA = -40°C REFERENCE VOLTAGE (V) TA = 25°C TA = 85°C 1.220 1.200 1.180 1.160 VDD = 1.8V VDD = 1.8V VDD = 3V VDD = 5.5V SUPPLY AND REFERENCE VOLTAGES (V) Comparator Output Short-Circuit Current vs. Supply Voltage OUTPUT SHORT-CIRCUIT CURRENT (mA) Reference Voltage vs. Load Current Line Transient Response of VREF 4 VDD 3 Sinking 30 20 Sinking 10 Sourcing 0 0 TA = -4 C 0° Sourcing 2 VREF TA = 25°C TA = 85°C 6 VDD = 5.5V VDD = 3V 1 1.140 0 2 4 0 0 100 200 TIME (µsec) 300 400 3 1 2 4 5 SUPPLY VOLTAGE (V) 6 8 10 LOAD CURRENT (mA) 2002 Microchip Technology Inc. DS21347B-page 13 © TC1043 TYPICAL CHARACTERISTICS (CONTINUED) Op-Amp DC Open-Loop Gain vs. Supply Voltage 140 DC OPEN-LOOP GAIN (dB) 120 100 2000 80 1500 60 40 20 0 0.0 1000 500 0 -40°C 3000 OUTPUT CURRENT (mA) 2500 Op-Amp DC Open-Loop Gain vs. Temperature 50 45 40 35 30 25 20 15 10 5 25°C TEMPERATURE (°C) 85°C Op-Amp Short-Circuit Current vs. Supply Voltage ISINK 1.0 2.0 3.0 4.0 5.0 6.0 0 0.0 1.0 SUPPLY VOLTAGE (V) 2.0 3.0 4.0 5 .0 SUPPLY VOLTAGE (V) 6.0 0 OUTPUT CURRENT (mA) -5 -10 RLOAD (kΩ) 1000 10% Overshoot V V INPUT VOLTAGE (mV) Op-Amp Short-Circuit Current vs. Supply Voltage Op-Amp Load Resistance vs. Load Capacitance = 1.5V Op-Amp Small-Signal Transient Response 100 50 0 100 Region of Marginal Stability -15 -20 -25 -30 -35 0.0 ISRC OUTPUT VOLTAGE (mV) 10 Region of Stable Operation 100 50 0 10 20 30 40 50 60 70 80 90 TIME (µsec) 1 1.0 2.0 3.0 4.0 5.0 SUPPLY VOLTAGE (V) 6.0 0 250 500 750 1000 1250 1500 1750 2000 INPUT VOLTAGE (mV) Large-Signal Transient Response 6 4 2 PSRR (dB) 0 6 4 2 0 10 20 30 40 50 60 70 80 90 TIME (µsec) Op-Amp Power Supply Rejection Ratio (PSRR) vs. Frequency 0 -10 -20 -30 -40 -50 -60 -70 100 1K 10K 100K FREQUENCY (Hz) VDD = 3V VCM = 1.5V VIN = 100mVPP © DS21347B-page 14 OUTPUT VOLTAGE (mV) 2002 Microchip Technology Inc. TC1043 TYPICAL CHARACTERISTICS (CONTINUED) Reference Voltage vs. Supply Voltage 1.25 REFERENCE VOLTAGE (V) Supply Current vs. Supply Voltage 20 18 16 TA = 25°C 14 12 10 TA = -40°C TA = 85°C 1.20 1.15 1.10 1.05 1 4 2 3 SUPPLY VOLTAGE (V) 5 SUPPLY CURRENT (µA) 8 0 1 2 3 4 5 SUPPLY VOLTAGE (V) 6 2002 Microchip Technology Inc. DS21347B-page 15 © TC1043 6.0 6.1 PACKAGING INFORMATION Package Marking Information Package marking information not available at this time. 6.2 Taping Information Component Taping Orientation for 16-Pin QSOP (Narrow) Devices User Direction of Feed PIN 1 W P Standard Reel Component Orientation for TR Suffix Device Carrier Tape, Reel Size, Number of Components Per Reel and Reel Size Package Carrier Width (W) Pitch (P) Part Per Full Reel Reel Size 16-Pin QSOP (N) 12 mm 8 mm 2500 13 in 6.3 Package Dimensions 16-Pin QSOP (Narrow) PIN 1 .157 (3.99) .150 (3.81) .244 (6.20) .227 (5.79) .196 (4.98) .189 (4.80) .010 (0.25) .004 (0.10) .069 (1.75) .053 (1.35) 8° MAX. .050 (1.27) .015 (0.40) Dimensions: inches (mm) .010 (0.25) .007 (0.19) (0.635) .011 (0.30) BSC .007 (0.20) © DS21347B-page 16 2002 Microchip Technology Inc. TC1043 SALES AND SUPPORT Data Sheets Products supported by a preliminary Data Sheet may have an errata sheet describing minor operational differences and recommended workarounds. To determine if an errata sheet exists for a particular device, please contact one of the following: 1. 2. 3. Your local Microchip sales office The Microchip Corporate Literature Center U.S. FAX: (480) 792-7277 The Microchip Worldwide Site (www.microchip.com) Please specify which device, revision of silicon and Data Sheet (include Literature #) you are using. New Customer Notification System Register on our web site (www.microchip.com/cn) to receive the most current information on our products. 2002 Microchip Technology Inc. DS21347B-page 17 © TC1043 NOTES: © DS21347B-page 18 2002 Microchip Technology Inc. TC1043 Information contained in this publication regarding device applications and the like is intended through suggestion only and may be superseded by updates. It is your responsibility to ensure that your application meets with your specifications. No representation or warranty is given and no liability is assumed by Microchip Technology Incorporated with respect to the accuracy or use of such information, or infringement of patents or other intellectual property rights arising from such use or otherwise. Use of Microchip’s products as critical components in life support systems is not authorized except with express written approval by Microchip. No licenses are conveyed, implicitly or otherwise, under any intellectual property rights. Trademarks The Microchip name and logo, the Microchip logo, FilterLab, KEELOQ, microID, MPLAB, PIC, PICmicro, PICMASTER, PICSTART, PRO MATE, SEEVAL and The Embedded Control Solutions Company are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. dsPIC, ECONOMONITOR, FanSense, FlexROM, fuzzyLAB, In-Circuit Serial Programming, ICSP, ICEPIC, microPort, Migratable Memory, MPASM, MPLIB, MPLINK, MPSIM, MXDEV, PICC, PICDEM, PICDEM.net, rfPIC, Select Mode and Total Endurance are trademarks of Microchip Technology Incorporated in the U.S.A. Serialized Quick Turn Programming (SQTP) is a service mark of Microchip Technology Incorporated in the U.S.A. All other trademarks mentioned herein are property of their respective companies. © 2002, Microchip Technology Incorporated, Printed in the U.S.A., All Rights Reserved. Printed on recycled paper. Microchip received QS-9000 quality system certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona in July 1999 and Mountain View, California in March 2002. The Company’s quality system processes and procedures are QS-9000 compliant for its PICmicro ® 8-bit MCUs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, non-volatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001 certified.  2002 Microchip Technology Inc. DS21347B - page 19 WORLDWIDE SALES AND SERVICE AMERICAS Corporate Office 2355 West Chandler Blvd. Chandler, AZ 85224-6199 Tel: 480-792-7200 Fax: 480-792-7277 Technical Support: 480-792-7627 Web Address: http://www.microchip.com ASIA/PACIFIC Australia Microchip Technology Australia Pty Ltd Suite 22, 41 Rawson Street Epping 2121, NSW Australia Tel: 61-2-9868-6733 Fax: 61-2-9868-6755 Japan Microchip Technology Japan K.K. Benex S-1 6F 3-18-20, Shinyokohama Kohoku-Ku, Yokohama-shi Kanagawa, 222-0033, Japan Tel: 81-45-471- 6166 Fax: 81-45-471-6122 Rocky Mountain 2355 West Chandler Blvd. Chandler, AZ 85224-6199 Tel: 480-792-7966 Fax: 480-792-7456 China - Beijing Microchip Technology Consulting (Shanghai) Co., Ltd., Beijing Liaison Office Unit 915 Bei Hai Wan Tai Bldg. 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