LX1681_05

LX1681/1682 TM ® Voltage-Mode PWM Controllers P RODUCTION D ATA S HEET DESCRIPTION KEY FEATURES W WW . Microsemi . C OM The LX1681/1682 are monolithic, pulse-width modulator controller ICs. They are designed to implement a flexible, low cost buck (step-down) regulator supply with minimal external components. The LX1681 is a non-synchronous controller; the LX1682 has a synchronous driver for higher efficiency. The output voltage is adjustable by means of a resistor divider to set the voltage between 1.25V and 4.5V. Short-circuit current limiting can be implemented without expensive current sense resistors. Current is sensed using the voltage drop across the RDS(ON) of the MOSFET — sensing is delayed for 1µs to eliminate MOSFET ringing errors. Hiccup-mode fault protection reduces average power to the power elements during short-circuit conditions. Switching frequency is fixed at 200kHz for optimal cost and space. Under-voltage lockout and soft-start for optimal start-up performance. Pulling the soft-start pin to ground can disable the LX1681/82. Small 8-pin SOIC packaging reduces board space. Optimized for 5V-to-3.3V or 5V-to-2.5V conversion, the LX1681/82 can also be used for converting 12V to 5V, 3.3V or other voltages with high efficiency, eliminating the need for bulky heat sinks. Fixed 200kHz Switching Frequency Constant Frequency Voltage-Mode Control Requires NO External Compensation Hiccup-Mode Over-Current Protection High Efficiency Output Voltage Set By Resistor Divider Under-Voltage Lockout Soft-Start And Enable Synchronous Rectification (LX1682) Non-Synchronous Rectification (LX1681) Small, 8-pin Surface Mount Package APPLICATIONS 5V to 3.3V Or Less Buck Regulators FPGA Supplies Microprocessor Chipset Supplies (e.g. Camino, Whitney, etc.) © Rambus RIMM™ Supplies Hard Disk Drives Computer Add-on Cards IMPORTANT: For the most current data, consult MICROSEMI’s website: http://www.microsemi.com PRODUCT HIGHLIGHT V BO OST 12V V IN 5V C3 1µF C1 1500µF x3 C3 1µF V BOO ST 12V V IN 5V C1 V FB C SS 0.1µF V C2 CS V FB C SS V C2 CS 1500µF x3 SS SS L X1681 N.C. GND V C1 TDRV R SET D2 Q1 IRL3103S L1 5µH C2 1500µF x3 R1 0.1µF V OU T LX1682 G ND BDRV V C1 TDRV R SET Q1 IRL3103S L1 5µH C2 1500µF x3 R1 V OU T R2 MBR2545 R2 Q2 IRL3103S LX1681/1682 LX1681/1682 LX1681 Non-Synchronous Controller LX1682 Synchronous Controller PACKAGE ORDER INFO TA (°C) 0 to 70 OUTPUT Non-Synchronous Synchronous DM Plastic SOIC 8-PIN LX1681CDM LX1682CDM RoHS Compliant / Pb-free Transition D/C: 0440 Note: Available in Tape & Reel. Append the letters “TR” to the part number. (i.e. LX1681CDM-TR) Copyright © 2000 Rev. 1.1b,2005-03-09 Microsemi Linfinity Microelectronics Division 11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570 Page 1 LX1681/1682 TM ® Voltage-Mode PWM Controllers P RODUCTION D ATA S HEET ABSOLUTE MAXIMUM RATINGS (NOTE 1) Supply Voltage (VC1).................................................................................................... 18V Supply Voltage (VCC) ..................................................................................................... 7V Output Drive Peak Current Source (500ns) ................................................................. 1.0A Output Drive Peak Current Sink (500ns) ..................................................................... 1.0A Input Voltage (SS/ENABLE Pin) ........................................................................ -0.3 to 6V Operating Junction Temperature................................................................................ 150°C Storage Temperature................................................................................. -65°C to +150°C RoHS / Pb-free Peak Package Solder Reflow Temp (40 second max. exposure) ........... 260°C (+0, -5) Note: Exceeding these ratings could cause damage to the device. All voltages are with respect to Ground. Currents are positive into, negative out of specified terminal. PACKAGE PIN OUT W WW . Microsemi . C OM V FB SS N.C. GND 1 2 3 4 8 7 6 5 V CC CS V C1 TDRV LX1681 DM P ACKAGE (Top View) V FB SS GND BDRV 1 2 3 4 8 7 6 5 V CC CS V C1 TDRV THERMAL DATA LX1682 DM P ACKAGE (Top View) DM PACKAGE 165°C/W RoHS / Pb-free 100% Matte Tin Lead Finish THERMAL RESISTANCE-JUNCTION TO AMBIENT, θJA Junction Temperature Calculation: TJ = TA + (PD x θJA). The θJA numbers are guidelines for the thermal performance of the device/pc-board system. All of the above assume no ambient airflow. FUNCTIONAL PIN DESCRIPTION PIN NAME VFB DESCRIPTION Voltage Feedback. A 1.25V reference is connected to a resistor divider to set desired output voltage. Soft-Start And Hiccup Capacitor Pin. During start up the voltage of this pin controls the output voltage. An internal 20kΩ resistor and the external capacitor set the time constant for soft-startup. Soft-start does not begin until the supply voltage exceeds the UVLO threshold. When over-current occurs, this capacitor is used for timing hiccup. The PWM can be disabled by pulling the SS pin below 0.3V Ground for IC. Gate Drive For Upper MOSFET. SS GND TDRV BDRV VC1 CS VCC PACKAGE DATA PACKAGE DATA Gate Drive For Lower MOSFET. Separate Supply For MOSFET Gate Drive. Connect to 12V. Over-Current Set. Connect resistor between CS pin and the source of the upper MOSFET to set current-limit point. IC Supply Voltage (nominal 5V) And High Side Drain Sense Voltage. Copyright © 2000 Rev. 1.1b,2005-03-09 Microsemi Linfinity Microelectronics Division 11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570 Page 2 LX1681/1682 TM ® Voltage-Mode PWM Controllers P RODUCTION D ATA S HEET ELECTRICAL CHARACTERISTICS W WW . Microsemi . C OM Unless otherwise specified, the following specifications apply over the operating ambient temperature 0°C ≤ TA ≤ 70°C except where otherwise noted. Test conditions: VCC=5V, VC1=12V, T=25°C Parameter REFERENCE Symbol Test Conditions LX1681/1682 Min Typ Max Units Reference Voltage VFB VOUT=VFB, TA=25°C VOUT=VFB, 0°C < TA < 70°C 1.237 1.231 1.25 1.262 1.269 V V OSCILLATOR Frequency Ramp Amplitude ERROR AMPLIFIER FOSC VRAMP 170 190 1.25 230 kHz VPP Input Resistance CURRENT SENSE RIN VOUT=VFB 20 kΩ Current Set VTRIP Current Sense Delayed OUTPUT DRIVERS ISET VCS = VCC –0.4V Reference to VCC 40 40 45 45 1.1 µA µA µsec TCSD Drive Rise Time, Fall Time Drive High Drive Low UVLO AND SOFT-START (SS) TRF VDH VDL CL=3000pF ISOURCE=20mA, VC1=12V ISINK=20mA, VC1=12V 10 50 11 0.1 0.2 Ns V V VCC5 Start-Up Threshold Hysteresis SS Resistor SS Output Enable Hiccup Duty Cycle SUPPLY CURRENT VST VC1 > 4.0V 4.0 4.25 0.10 4.5 V V kΩ RSS VEN DCHIC CSS = 0.1µF, FREQ=100Hz 0.25 20 0.3 10 0.35 V % ELECTRICALS ELECTRICALS VCC12 Dynamic Supply Current Static Supply Current 12CV 5V ICD IVC1 IVCC Out Freq = 200kHz, CL=3000pF, Synch., VSS > 0.3V VSS < 0.3V VSS > 0.3V 24 5 10 28 7 12 mA mA mA Copyright © 2000 Rev. 1.1b,2005-03-09 Microsemi Linfinity Microelectronics Division 11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570 Page 3 LX1681/1682 TM ® Voltage-Mode PWM Controllers P RODUCTION D ATA S HEET THEORY OF OPERATION W WW . Microsemi . C OM GENERAL DESCRIPTION OVER-CURRENT PROTECTION (OCP) and HICCUP The LX1681/82 are voltage-mode pulse-width modulation controller integrated circuits. The internal oscillator and ramp generator frequency is fixed at 200kHz. The devices have internal compensation, so that no external compensation is required. POWER UP and INITIALIZATION At power up, the LX1681/82 monitors the supply voltage to both the +5V and the +12V pins (there is no special requirement for the sequence of the two supplies). Before both supplies reach their under-voltage lock-out (UVLO) thresholds, the soft-start (SS) pin is held low to prevent soft-start from beginning; the oscillator control is disabled and the top MOSFET is kept OFF. SOFT-START Once the supplies are above the UVLO threshold, the soft-start capacitor begins to be charged up by the reference through a 20k internal resistor. The capacitor voltage at the SS pin rises as a simple RC circuit. The SS pin is connected to the amplifier's noninverting input that controls the output voltage. The output voltage will follow the SS pin voltage if sufficient charging current is provided to the output capacitor. The simple RC soft-start allows the output to rise faster at the beginning and slower at the end of the soft-start interval. Thus, the required charging current into the output capacitor is less at the end of the soft-start interval so decreasing the possibility of an over-current. A comparator monitors the SS pin voltage and indicates the end of soft-start when SS pin voltage reaches 95% of VREF. The LX1681/1682 family uses the RDS(ON) of the upper MOSFET, together with a resistor (RSET) to set the actual current limit point. The comparator senses the current 1µs after the top MOSFET is switched on. Experiments have shown that the MOSFET drain voltage will ring for 200-500ns after the gate is turned on. In order to reduce inaccuracies due to ringing, a 1µs delay after gate turn-on is built into the current sense comparator. The comparator draws a current (ISET), whose magnitude is 45µA. The set resistor is selected to set the current limit for the application. When the sensed voltage across the RDS(ON) plus the set resistor exceeds the 400mV V TRIP threshold, the OCP comparator outputs a signal to reset the PWM latch and to start hiccup mode. The soft-start capacitor (CSS) is discharged slowly (10 times slower than when being charged up by RSS). When the voltage on the SS/ENABLE pin reaches a 0.3V threshold, hiccup finishes and the circuit soft-starts again. During hiccup, the top MOSFET is OFF and the bottom MOSFET remains ON. Hiccup is disabled during the soft-start interval, allowing the circuit to start up with the maximum current. If the rise speed of the output voltage is too fast, the required charging current to the output capacitor may be higher than the limit-current. In this case, the peak MOSFET current is regulated to the limit-current by the current-sense comparator. If the MOSFET current still reaches its limit after the soft-start finishes, the hiccup is triggered again. The hiccup ensures the average heat generation on both MOSFET’s and the average current to be much less than that in normal operation, if the output has a short circuit. Over-current protection can also be implemented using a sense resistor, instead of using the RDS(ON) of the upper MOSFET, for greater set-point accuracy. See Application Information section. OSCILLATOR FREQUENCY An internal oscillator sets the switching frequency at 200 kHz. DESCRIPTION DESCRIPTION Copyright © 2000 Rev. 1.1b,2005-03-09 Microsemi Linfinity Microelectronics Division 11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570 Page 4 LX1681/1682 TM ® Voltage-Mode PWM Controllers P RODUCTION D ATA S HEET APPLICATION INFORMATION OUTPUT INDUCTOR OUTPUT CAPACITOR (continued) W WW . Microsemi . C OM The output inductor should be selected to meet the requirements of the output voltage ripple in steady-state operation and the inductor current slew-rate during transient. The peak-to-peak output voltage ripple is: VRIPPLE = ESR × IRIPPLE where IRIPPLE = (VIN - VOUT ) × VOUT fSW × L VIN IRIPPLE is the inductor ripple current, L is the output inductor value and ESR is the Effective Series Resistance of the output capacitor. IRIPPLE should typically be in the range of 20% to 40% of the maximum output current. Higher inductance results in lower output voltage ripple, allowing slightly higher ESR to satisfy the transient specification. Higher inductance also slows the inductor current slew rate in response to the load-current step change, ∆I, resulting in more output-capacitor voltage droop. The inductorcurrent rise and fall times are: Electrolytic capacitors can be used for the output capacitor, but are less stable with age than tantalum capacitors. As they age, their ESR degrades, reducing the system performance and increasing the risk of failure. It is recommended that multiple parallel capacitors be used, so that, as ESR increases with age, overall performance will still meet the processor’s requirements. There is frequently strong pressure to use the least expensive components possible, however, this could lead to degraded longterm reliability, especially in the case of filter capacitors. Linfinity’s demonstration boards use Sanyo MV-GX filter capacitors, which are aluminum electrolytic, and have demonstrated reliability. The Oscon series from Sanyo generally provides the very best performance in terms of long term ESR stability and general reliability, but at a substantial cost penalty. The MV-GX series provides excellent ESR performance at a reasonable cost. Beware of off-brand, very low-cost filter capacitors, which have been shown to degrade in both ESR and general electrolytic characteristics over time. INPUT CAPACITOR TRISE = and L × ∆I (VIN − VOUT ) The input capacitor and the input inductor are to filter the pulsating current generated by the buck converter to reduce interference to other circuits connected to the same 5V rail. In addition, the input capacitor provides local de-coupling the buck converter. The capacitor should be rated to handle the RMS current requirement. The RMS current is: TFALL = L × ∆I VOUT IRMS = IL d (1 − d ) where IL is the inductor current and the d is the duty cycle. The maximum value, when d = 50%, IRMS = 0.5IL . For 5V input and output in the range of 2 to 3V, the required RMS current is very close to 0.5IL. SOFT-START CAPACITOR When using electrolytic capacitors, the capacitor voltage droop is usually negligible, due to the large capacitance. OUTPUT CAPACITOR The output capacitor is sized to meet ripple and transient performance specifications. Effective Series Resistance (ESR) is a critical parameter. When a step load current occurs, the output voltage will have a step that equals the product of the ESR and the current step, ∆I. In an advanced microprocessor power supply, the output capacitor is usually selected for ESR instead of capacitance or RMS current capability. A capacitor that satisfies the ESR requirement usually has a larger capacitance and current capability than strictly needed. The allowed ESR can be found by: The value of the soft-start capacitor determines how fast the output voltage rises and how large the inductor current is required to charge the output capacitor. The output voltage will follow the voltage at SS pin if the required inductor current does not exceed the maximum current in the inductor. The SS pin voltage can be expressed as: APPLICATION APPLICATION VSS = VSET (1 − e −t / RSSCSS ) where VSET is the reference voltage. RSS and CSS are soft start resistor and capacitor. The required inductor current for the output capacitor to follow the SS-pin voltage equals the required capacitor current plus the load current. The soft-start capacitor should be selected so that the overall inductor current does not exceed it maximum. ESR × (IRIPPLE + ∆I ) < VEX where IRIPPLE is the inductor ripple current, ∆I is the maximum load current step change, and VEX is the allowed output voltage excursion in the transient. Copyright © 2000 Rev. 1.1b,2005-03-09 Microsemi Linfinity Microelectronics Division 11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570 Page 5 LX1681/1682 TM ® Voltage-Mode PWM Controllers P RODUCTION D ATA S HEET APPLICATION INFORMATION SOFT-START CAPACITOR (continued) OUTPUT ENABLE W WW . Microsemi . C OM The capacitor current to follow the SS-pin voltage is: ICOUT = COUT dV COUT −(t / RSSCSS ) = ×e dt CSS The LX1681/82 FET driver outputs are driven to ground by pulling the soft-start pin below 0.3V. PROGRAMMING THE OUTPUT VOLTAGE where COUT is the output capacitance. The typical value of CSS should be in the range of 0.1 to 0.2µF. During the soft-start interval the load current from a microprocessor is negligible; therefore, the capacitor current is approximately the required inductor current. OVER-CURRENT PROTECTION The output voltage is sensed by the feedback pin (VFB ) which has a 1.25V reference. The output voltage can be set to any voltage above 1.25V (and lower than the input voltage) by means of a resistor divider (see Product Highlight). VOUT = VREF (1 + R1 ) R2 Current limiting occurs at current level ICL , when the voltage detected by the current sense comparator is greater than the current sense comparator threshold, VTRIP (400mV). Note: Keep R1 and R2 close to 100(order of magnitude). FET SELECTION ICL × RDS (ON ) + ISET × RSET = VTRIP So, RSET = RSET = VTRIP − ICL × RDS (ON ) ISET 400mV − ICL × RDS (ON ) 45µA Example: For 10A current limit, using IRL3303 MOSFET (26mΩ RDS(ON) ): To insure reliable operation, the operating junction temperature of the FET switches must be kept below certain limits. The Intel specification states that 115°C maximum junction temperature should be maintained with an ambient of 50°C. This is achieved by properly derating the part, and by adequate heat sinking. One of the most critical parameters for FET selection is the RDS(ON) resistance. This parameter directly contributes to the power dissipation of the FET devices, and thus impacts heat sink design, mechanical layout, and reliability. In general, the larger the current handling capability of the FET, the lower the RDS(ON) will be, since more die area is available. This table gives selection of suitable FETs from International Rectifier. RDS(ON) ID @ Max. BreakDevice @10V(mΩ) TC=100ºC down Voltage IRL3803 6 83 30 IRL22203N 7 71 30 IRL3103 14 40 30 IRL3102 13 56 20 IRL3303 26 24 30 IRL2703 40 17 30 All devices in TO-220 package. For surface mount devices (TO-263 / D 2 -Pak), add 'S' to part number, e.g. IRL3103S. TABLE 1 - FET Selection Guide RSET = 0.4 − 10 × 0.026 = 3.1kΩ 45 ×10 −6 Current Sensing Using Sense Resistor The method of current sensing using the RDS(ON) of the upper MOSFET is economical, but can have a large tolerance, since the RDS(ON) can vary with temperature, etc. A more accurate alternative is to use an external sense resistor (RSENSE ). Since one input to the current sense comparator is the supply voltage to the IC (VCC - pin 8), the sense resistor could be a PCB trace (for construction details, see Application Note AN-10 or LX1668 data sheet). The overcurrent trip point is calculated as in the equations above, replacing RDS(ON) with RSENSE . Example: For 10A current limit, using a 5µ sense resistor: Heat Dissipated In Upper MOSFET The heat dissipated in the top MOSFET will be: APPLICATION APPLICATION PD = ( I 2 × RDS (ON ) × Duty Cycle ) + (0.5 × I × VIN × tSW × fS ) Where t SW is switching transition line for body diode (~100ns) and fS is the switching frequency. For the IRL3102 (13µ RDS(ON) ), converting 5V to 2.0V at 15A will result in typical heat dissipation of 1.92W. RSET = RSET = VTRIP − ( ICL × RSENSE ) ISET 0.4 − 10 × 0.005 = 7.8kΩ 45 × 10 −6 Copyright © 2000 Rev. 1.1b,2005-03-09 Microsemi Linfinity Microelectronics Division 11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570 Page 6 LX1681/1682 TM ® Voltage-Mode PWM Controllers P RODUCTION D ATA S HEET APPLICATION INFORMATION FET SELECTION (continued) Synchronous Rectification – Lower MOSFET 5V Input W WW . Microsemi . C OM The lower pass element can be either a MOSFET or a Schottky diode. The use of a MOSFET (synchronous rectification) will result in higher efficiency, but at higher cost than using a Schottky diode (non-synchronous). Power dissipated in the bottom MOSFET will be: PD = I 2 × RDS (ON ) × [1 − Duty Cycle ] = 3.51W [IRL3303 or 1.76W for the IRL3102] Output L X168x Non-Synchronous Operation - Schottky Diode A typical Schottky diode, with a forward drop of 0.6V will dissipate 0.6 * 15 * [1 – 2/5] = 5.4W (compared to the 1.8 to 3.5W dissipated by a MOSFET under the same conditions). This power loss becomes much more significant at lower duty cycles. The use of a dual Schottky diode in a single TO-220 package (e.g. the MBR2535) helps improve thermal dissipation. Operation From A Single Power Supply GND The LX1681/1682 needs a secondary supply voltage (VC1) to provide sufficient drive to the upper MOSFET. In many applications with a 5V (VCC) and a 12V (VC1) supply are present. In situations where only 5V is present, VC1 can be generated using a bootstrap (charge pump) circuit, as shown in Figure 4 (Typical Applications section). The capacitor (C4) is alternatively charged up from VCC via the Schottky diode (D2), and then boosted up when the FET is turned on. This scheme provides a VC1 voltage equal to 2 * VCC - VDS (D2), or approximately 9.5V with VCC = 5V. This voltage will provide sufficient gate drive to the external MOSFET in order to get a low RDS(ON) . Note that using the bootstrap circuit in synchronous rectification mode is likely to result in faster turn-on than in non-synchronous mode. LAYOUT GUIDELINES - THERMAL DESIGN FIGURE 2 — Enabling Linear Regulator General Notes As always, be sure to provide local capacitive decoupling close to the chip. Be sure use ground plane construction for all highfrequency work. Use low ESR capacitors where justified, but be alert for damping and ringing problems. High-frequency designs demand careful routing and layout, and may require several iterations to achieve desired performance levels. Power Traces A great deal of time and effort were spent optimizing the thermal design of the demonstration boards. Any user who intends to implement an embedded motherboard would be well advised to carefully read and follow these guidelines. If the FET switches have been carefully selected, external heatsinking is generally not required. However, this means that copper trace on the PC board must now be used. This is a potential trouble spot; as much copper area as possible must be dedicated to heatsinking the FET switches, and the diode as well if a non-synchronous solution is used. In our VRM module, heatsink area was taken from internal ground and VCC planes which were actually split and connected with VIAS to the power device tabs. The TO-220 and TO-263 cases are well suited for this application, and are the preferred packages. Remember to remove any conformal coating from all exposed PC traces which are involved in heatsinking. To reduce power losses due to ohmic resistance, careful consideration should be given to the layout of traces that carry high currents. The main paths to consider are: Input power from 5V supply to drain of top MOSFET. Trace between top MOSFET and lower MOSFET or Schottky diode. Trace between lower MOSFET or Schottky diode and ground. Trace between source of top MOSFET and inductor and load. All of these traces should be made as wide and thick as possible, in order to minimize resistance and hence power losses. It is also recommended that, whenever possible, the ground, input and output power signals should be on separate planes (PCB layers). See Figure 2 – bold traces are power traces. Layout Assistance APPLICATION APPLICATION Please contact Linfinity’s Applications Engineers for assistance with any layout or component selection issues. A Gerber file with layout for the most popular devices is available upon request. Evaluation boards are also available upon request. Please check Linfinity's web site for further application notes. Copyright © 2000 Rev. 1.1b,2005-03-09 Microsemi Linfinity Microelectronics Division 11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570 Page 7 LX1681/1682 TM ® Voltage-Mode PWM Controllers P RODUCTION D ATA S HEET TYPICAL APPLICATION V BOO ST 12V W WW . Microsemi . C OM V IN 5V C3 C1 V FB C SS 0.1µF V C2 CS R SENSE SS L X1681 N.C. GND V C1 TDRV R SET V OUT Q1 L1 C2 R1 R2 D2 V IN 5V C1 V FB C SS 0.1µF V C2 CS D2 V O UT SS L X1682 GND BDRV V C1 TDRV C4 Q1 L1 C2 R1 APPLICATION APPLICATION R SET Q2 R2 Copyright © 2000 Rev. 1.1b,2005-03-09 Microsemi Linfinity Microelectronics Division 11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570 Page 8 LX1681/1682 TM ® Voltage-Mode PWM Controllers P RODUCTION D ATA S HEET BLOCK DIAGRAM R SET I SE T W WW . Microsemi . C OM 7 CS +12V C S Com p I RES ET PW M V IN (5V) R2 C IN R1 6 V C1 V CC + V T R IP 320k I S ET R S Q Q 5 TDRV L 1 V FB 20k Am plifier/ Com pensation V CO RE ESR C O UT Error Com p Set + + V RE SE T 4 BDRV V REF R SS Hiccup Hiccup Ramp Oscillator UVLO 3 GND +5V UVLO 8 V CC 2 SS/ENABLE C SS PHYSICAL DIMENSIONS DM A 8-Pin Plastic SOIC Dim A B C D F G J K L M P *LC Note: 1. Dimensions do not include mold flash or protrusions; these shall not exceed 0.155mm(.006”) on any side. Lead dimension shall not include solder coverage. P B G L C D K J F M MILLIMETERS MIN MAX 4.83 5.00 3.81 3.94 1.35 1.75 0.33 0.51 0.77 1.27 BSC 0.19 0.25 0.13 0.25 4.80 5.21 8° 5.79 6.20 0.10 INCHES MIN MAX 0.190 0.197 0.150 0.155 0.053 0.069 0.013 0.020 0.030 0.050 BSC 0.007 0.010 0.005 0.010 0.189 0.205 8° 0.228 0.244 0.004 BLOCK DIAGRAM BLOCK DIAGRAM *Lead Coplanarity Copyright © 2000 Rev. 1.1b,2005-03-09 Microsemi Linfinity Microelectronics Division 11861 Western Avenue, Garden Grove, CA. 92841, 714-898-8121, Fax: 714-893-2570 Page 9