MP24875
200kHz, 55V Input, 1.5A
High Power LED Driver
The Future of Analog IC Technology
DESCRIPTION
FEATURES
The MP24875 is a fixed frequency step-down
switching regulator to deliver a constant current of
up to 1.5A to high power LEDs. It integrates a
high-side, high voltage power MOSFET. The wide
4.5V to 55V input range accommodates a variety
of step-down applications, making it ideal for
automotive, industry and general lighting
applications. Peak current mode control and
200mV reference are applied for fast loop
response, easy compensation and accurate LED
current regulation. The switching frequency is
programmable with an external resistor up to
200kHz, which can prevent EMI (Electromagnetic
Interference) noise problems. The thermal shut
down provides reliable, fault tolerant operations. A
12µA quiescent current in shutdown mode allows
its use in battery-powered applications.
•
•
•
The MP24875 is available in SOIC8 with exposed
pad package.
“MPS” and “The Future of Analog IC Technology” are Registered Trademarks of
Monolithic Power Systems, Inc.
•
•
•
•
•
•
Wide 4.5V to 55V Operating Input Range
220mΩ Internal Power MOSFET
Up to 200kHz Programmable Switching
Frequency
130µA Quiescent Current
Ceramic Capacitor Stable
Internal Soft-Start
Up to 97.5% Efficiency
200mV reference voltage for high efficiency
Available SOIC8 with Exposed Pad
Package
APPLICATIONS
•
•
•
High Power LED Driver
Automotive, and General Lighting
Constant Source
TYPICAL APPLICATION
Efficiency vs. Input Voltage
CCOM2
10 WLED@330mA
100
CONTROL
RFB
VIN
EN
BST
MP24875
RFREQ
CFREQ
COMP
FREQ
LED-
CBST
L1
LED+
SW
D1
FB
VIN
8V to 55V
CIN
GND
COUT
EFFICIENCY(%)
RCOM
CCOM1
98
96
94
92
90
30
35
40
45
50
55
60
INPUT VOLTAGE(V)
MP24875 Rev. 1.0
9/21/2010
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MP24875 – 1.5A, 200kHz, 55V HIGH-POWER LEDS DRIVER
ORDERING INFORMATION
Part Number*
Package
Top Marking
Free Air Temperature (TA)
MP24875DN
SOIC8E
MP24875DN
–40°C to +85°C
*For Tape & Reel, add suffix –Z (eg. MP24875 DN–Z);
For RoHS compliant packaging, add suffix –LF (eg. MP24875 DN–LF–Z)
PACKAGE REFERENCE
TOP VIEW
SW
1
8
BST
EN
2
7
VIN
COMP
3
6
FREQ
FB
4
5
GND
EXPOSED PAD
ON BACKSIDE
CONNECT TO GND
SOIC8E
ABSOLUTE MAXIMUM RATINGS (1)
Thermal Resistance
Supply Voltage (VIN)......................-0.3V to +60V
Switch Voltage (VSW).............-0.5V to VIN + 0.5V
BST to SW ......................................-0.3V to +5V
All Other Pins ..................................-0.3V to +5V
(2)
Continuous Power Dissipation (TA = +25°C)
……………………………………………....2.5W
Junction Temperature ...............................150°C
Lead Temperature ....................................260°C
Storage Temperature............... -65°C to +150°C
SOIC8 (Exposed Pad) ............ 50 ...... 10... °C/W
Recommended Operating Conditions
(3)
Supply Voltage VIN ...........................4.5V to 55V
Operating Junc Temp(TJ)…….-40°C to +125°C
MP24875 Rev. 1.0
9/21/2010
(4)
θJA
θJC
Notes:
1) Exceeding these ratings may damage the device.
2) The maximum allowable power dissipation is a function of the
maximum junction temperature TJ(MAX), the junction-toambient thermal resistance θJA, and the ambient temperature
TA. The maximum allowable continuous power dissipation at
any ambient temperature is calculated by PD(MAX)=(TJ(MAX)TA)/ θJA. Exceeding the maximum allowable power dissipation
will cause excessive die temperature, and the regulator will go
into thermal shutdown. Internal thermal shutdown circuitry
protects the device from permanent damage.
3) The device is not guaranteed to function outside of its
operating conditions.
4) Measured on JESD51-7, 4-layer PCB.
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MP24875 – 1.5A, 200kHz, 55V HIGH-POWER LEDS DRIVER
ELECTRICAL CHARACTERISTICS
VIN = 12V, VEN = 2.5V, VCOMP = 1.4V, TA= +25°C, unless otherwise noted.
Specifications over temperature are guaranteed by design and characterization.
Parameter
Symbol
Condition
Min
Typ
Max
Units
Feedback Voltage
Upper Switch On Resistance (5)
Upper Switch Leakage
Current Limit
Error Amp Voltage Gain
Error Amp Transconductance
Error Amp Min Source current
Error Amp Min Sink current
VIN UVLO Threshold
VIN UVLO Hysteresis
Soft-Start Time (5)
Oscillator Frequency
Minimum Switch On Time (5)
Shutdown Supply Current
Quiescent Supply Current
Thermal Shutdown
Minimum Off Time (5)
Minimum On Time (5)
EN Up Threshold
EN Threshold Hysteresis
VFB
RDS(ON)
4.5V < VIN < 55V
VBST – VSW = 5V
VEN = 0V, VSW = 0V
188
200
220
0.1
2.5
400
350
10
-10
3.0
0.4
0.18
200
100
12
130
150
100
100
1.5
300
212
mV
mΩ
µA
A
V/V
µA/V
µA
µA
V
V
ms
kHz
ns
µA
µA
°C
ns
ns
V
mV
2
AVEA
GEA
fs
ICOMP = ±3µA
VFB = 175mV
VFB = 225mV
VIN rising
2.7
20mV < VFB < 190mV
RFREQ = 495kΩ
150
VEN < 0.3V
No load, VFB = 240mV
Hysteresis = 20°C
1.3
3.3
250
25
1.7
Note:
5) Guaranteed by design.
MP24875 Rev. 1.0
9/21/2010
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MP24875 – 1.5A, 200kHz, 55V HIGH-POWER LEDS DRIVER
PIN FUNCTIONS
SOIC8
Pin #
Name
1
SW
2
EN
3
COMP
4
FB
5
6
7
8
Description
Switch Node. This is the output from the high-side switch. A low VF Schottky rectifier to
ground is required. The rectifier must be close to the SW pins to reduce switching spikes.
Enable Input. Pulling this pin below the specified threshold shuts the chip down. Pulling it up
above the specified threshold or leaving it floating enables the chip.
Compensation. This node is the output of the GM error amplifier. Control loop frequency
compensation is applied to this pin.
Feedback. This is the input to the error amplifier. An external current sensing resistor is
connected in series with the LEDs to GND. The feedback voltage is connected to this pin and
is compared to the internal +200mV reference to set the regulation current.
GND,
Ground. It should be connected as close as possible to the output capacitor avoiding the high
Exposed
current switch paths. Connect exposed pad to GND plane for optimal thermal performance.
pad
Switching Frequency Program Input. Connect a resistor from this pin to ground to set the
FREQ
switching frequency.
Input Supply. This supplies power to all the internal control circuitry, both BS regulators and
VIN
the high-side switch. A decoupling capacitor to ground must be placed close to this pin to
minimize the switching spikes.
Bootstrap. This is the positive power supply for the internal floating high-side MOSFET driver.
BST
Connect a bypass capacitor between this pin and SW pin.
MP24875 Rev. 1.0
9/21/2010
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MP24875 – 1.5A, 200kHz, 55V HIGH-POWER LEDS DRIVER
TYPICAL PERFORMANCE CHARACTERISTICS
VIN=44V, ILED=330mA, 10 WLED series, unless otherwise noted.
PWM Dimming
Steady State Operation
EN Start Up
fPWM=200Hz,DPWM=50%
VEN
5V/div
VSW
20V/div
VEN
5V/div
VSW
50V/div
VCOMP
500mV/div
VIN
20V/div
ILED
200mA/div
IINDUCTOR
200mA/div
ILED
200mA/div
VSW
20V/div
ILED
200mA/div
IINDUCTOR
200mA/div
2ms/div
4us/div
EN Shutdown
Efficiency vs. Input Voltage
10 WLED@330mA
VEN
5V/div
VSW
50V/div
ILED
200mA/div
IINDUCTOR
200mA/div
EFFICIENCY(%)
100
98
96
94
92
90
30
35
40
45
50
55
60
INPUT VOLTAGE(V)
MP24875 Rev. 1.0
9/21/2010
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MP24875 – 1.5A, 200kHz, 55V HIGH-POWER LEDS DRIVER
BLOCK DIAGRAM
VIN
REFERENCE
UVLO
EN
INTERNAL
REGULATORS
BST
ISW
0.5ms SS
-+
SS
LOGIC
SW
FB
SS
0V2
--
COMP
+
OSCILLATOR
COMP
GND
FREQ
Figure 1—Functional Block Diagram
MP24875 Rev. 1.0
9/21/2010
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MP24875 – 1.5A, 200kHz, 55V HIGH-POWER LEDS DRIVER
mode but
slightly.
OPERATION
PWM Control Mode
MP24875 operates in a fixed frequency, peak
current control mode to regulate the LED current.
A PWM cycle is initiated by the internal clock.
The power MOSFET is turned on and remains on
until its current reaches the value set by the
COMP voltage. When the power switch is off, it
remains off for at least 100ns before the next
cycle starts. In one PWM period, the current in
the power MOSFET does not reach the COMP
set current value, the power MOSFET remains
on, saving a turn-off operation.
Error Amplifier
The error amplifier compares the FB pin voltage
with the internal reference (REF) and outputs a
current proportional to the difference between the
two. This output current is then used to charge
the external compensation network to form the
COMP voltage, which is used to control the
power MOSFET current.
During operation, the minimum COMP voltage is
clamped to 0.9V and its maximum is clamped to
2.0V. COMP is internally pulled down to GND in
shutdown mode. COMP should not be pulled up
beyond 2.6V.
Internal Regulator
Most of the internal circuitries are powered from
the 2.6V internal regulator. This regulator takes
the VIN input and operates in the full VIN range.
When VIN is greater than 3.0V, the output of the
regulator is in full regulation. When VIN is lower
than 3.0V, the output decreases.
Enable Control
The MP24875 has a dedicated enable control pin
(EN). With high enough input voltage, the chip
can be enabled and disabled by EN which has
positive logic. Its falling threshold is a precision
1.2V, and its rising threshold is 1.5V (300mV
higher).
When floating, EN is pulled up to about 3.0V by
an internal 1µA current source so it is enabled.
To pull it down, 1µA current capability is needed.
When EN is pulled down below 1.2V, the chip is
put into the lowest shutdown current mode.
When EN is higher than zero but lower than its
rising threshold, the chip is still in shutdown
MP24875 Rev. 1.0
9/21/2010
the
shutdown
current
increases
Under-Voltage Lockout (UVLO)
Under-voltage lockout (UVLO) is implemented to
protect the chip from operating at insufficient
supply voltage. The UVLO rising threshold is
about 3.0V while its falling threshold is a
consistent 2.6V.
Internal Soft-Start
The soft-start is implemented to prevent the LED
current from overshooting during startup and
short circuit recovery. When the chip starts, the
internal circuitry generates a soft-start voltage
(SS) ramping up from 0V to 2.6V. When it is
lower than the internal reference (REF), SS
overrides REF so the error amplifier uses SS as
the reference. When SS is higher than REF, REF
regains control.
Thermal Shutdown
Thermal shutdown is implemented to prevent the
chip from operating at exceedingly high
temperatures. When the silicon die temperature
is higher than its upper threshold, it shuts down
the whole chip. When the temperature is lower
than its lower threshold, the chip is enabled
again.
Floating Driver and Bootstrap Charging
The floating power MOSFET driver is powered by
an external bootstrap capacitor. This floating
driver has its own UVLO protection. This UVLO’s
rising threshold is 2.2V with a threshold of
150mV. The driver’s UVLO is soft-start related. In
case the bootstrap voltage hits its UVLO, the
soft-start circuit is reset. To prevent noise, there
is 20µs delay before the reset action. When
bootstrap UVLO is gone, the reset is off and then
soft-start process resumes.
The bootstrap capacitor is charged and regulated
to about 4V by the dedicated internal bootstrap
regulator. When the voltage between the BST
and SW nodes is lower than its regulation, a
PMOS pass transistor connected from VIN to
BST is turned on. The charging current path is
from VIN, BST and then to SW. External circuit
should provide enough voltage headroom to
facilitate the charging.
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MP24875 – 1.5A, 200kHz, 55V HIGH-POWER LEDS DRIVER
As long as VIN is sufficiently higher than SW, the
bootstrap capacitor can be charged. When the
power MOSFET is ON, VIN is about equal to SW
so the bootstrap capacitor cannot be charged.
When the external diode is on, the difference
between VIN and SW is largest, thus making it
the best period to charge. When there is no
current in the inductor, SW equals the output
voltage VOUT so the difference between VIN and
VOUT can be used to charge the bootstrap
capacitor.
At higher duty cycle operation condition, the time
period available to the bootstrap charging is less
so the bootstrap capacitor may not be sufficiently
charged.
In case the internal circuit does not have
sufficient voltage and the bootstrap capacitor is
not charged, extra external circuitry can be used
to ensure the bootstrap voltage is in the normal
operational region. Refer to External Bootstrap
Diode in Application section.
Current Comparator and Current Limit
The power MOSFET current is accurately sensed
via a current sense MOSFET. It is then fed to the
high speed current comparator for the current
mode control purpose. The current comparator
takes this sensed current as one of its inputs.
When the power MOSFET is turned on, the
comparator is first blanked till the end of the turnon transition to avoid noise issues. The
comparator then compares the power switch
current with the COMP voltage. When the
sensed current is higher than the COMP voltage,
the comparator output is low, turning off the
power MOSFET. The cycle-by-cycle maximum
current of the internal power MOSFET is
internally limited.
While the internal supply rail is up, an internal
timer holds the power MOSFET OFF for about
50µs to blank the startup glitches. When the
internal soft-start block is enabled, it first holds its
SS output low to ensure the remaining circuitries
are ready and then slowly ramps up.
Three events can shut down the chip: EN low,
VIN low and thermal shutdown. In the shutdown
procedure, power MOSFET is turned off first to
avoid any fault triggering. The COMP voltage and
the internal supply rail are then pulled down.
Programmable Oscillator
The MP24875 oscillating frequency is set by an
external resistor, RFREQ from the FREQ pin to
ground. The value of RFREQ can be calculated
from:
RFREQ (kΩ) =
100000
-5
fS (kHz)
To get fS=200kHz, RFREQ=495kΩ.
A ceramic capacitor CFREQ should be Parallel to
RFREQ to decouple the noise, 1nF is enough for
most applications.
Startup and Shutdown
If both VIN and EN are higher than their
appropriate thresholds, the chip starts. The
reference block starts first, generating stable
reference voltage and currents, and then the
internal regulator is enabled. The regulator
provides stable supply for the remaining
circuitries.
MP24875 Rev. 1.0
9/21/2010
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MP24875 – 1.5A, 200kHz, 55V HIGH-POWER LEDS DRIVER
APPLICATION INFORMATION
Output Rectifier Diode
COMPONENT SELECTION
The output rectifier diode supplies the current to
the inductor when the high-side switch is off. To
reduce losses due to the diode forward voltage
and recovery times, use a Schottky diode.
Setting the LED Current
The LED current is set using a sensing resistor,
which is in series with the LEDs and connected
to GND. The voltage on the sensing resistor is
connected to FB pin.
ILED =
VFB
RFB
For example, for a 700mA LED current, RFB is
287mΩ.
Inductor L1
The inductor is required to supply constant
current to the output load while being driven by
the switched input voltage. A larger value
inductor will help to reduce the output filter
capacitance for the same LED current ripple.
However, the larger value inductor will have a
larger physical size, higher series resistance,
and/or lower saturation current.
A good rule for determining the inductance to use
is to allow the peak-to-peak ripple current in the
inductor to be approximately 30% to 40% of the
LED current. Also, make sure that the peak
inductor current is below the maximum switch
current limit. The inductance value can be
calculated by:
L1=
VOUT
fs × ∆IL
× (1-
VOUT
VIN
),
VOUT = n × VF
Where VOUT is the output voltage to drive the
LEDs, VIN is the input voltage, fS is the switching
frequency, VF is one LED diode forward voltage
drop, n is the numbers of LEDs in series, and ∆IL
is the peak-to-peak inductor ripple current.
Choose an inductor that will not saturate under
the maximum inductor peak current. The peak
inductor current can be calculated by:
ILP = ILED +
Choose a diode whose maximum reverse voltage
rating is greater than the maximum input voltage,
and whose current rating is greater than the
maximum load current. Table 2 lists example
Schottky diodes and manufacturers.
Table 1—Diode Selection Guide
Voltage/
Current
Rating
80V, 1A
90V, 2A
Diodes
B180-7-F
B290-F
Manufacturer
Diodes Inc.
Diodes Inc.
Input Capacitor CIN
The input current to the step-down converter is
discontinuous, therefore a capacitor is required to
supply the AC current to the step-down converter
while maintaining the DC input voltage. Use low
ESR capacitors for the best performance.
Ceramic capacitors are preferred, but tantalum or
low-ESR electrolytic capacitors may also suffice.
For simplification, choose the input capacitor with
RMS current rating greater than half of the
maximum load current. The input capacitor (C1)
can be electrolytic, tantalum or ceramic.
When using electrolytic or tantalum capacitors, a
small, high quality ceramic capacitor, i.e. 0.1µF,
should be placed as close to the IC as possible.
When using ceramic capacitors, make sure that
they have enough capacitance to provide
sufficient charge to prevent excessive voltage
ripple at input. The input voltage ripple caused by
capacitance can be estimated by:
∆VIN =
⎛
⎞
V
V
ILED
× OUT × ⎜ 1 − OUT ⎟
fs × CIN
VIN ⎝
VIN ⎠
VOUT
V
× (1 − OUT )
2 × fs × L1
VIN
Where ILED is the LED current.
MP24875 Rev. 1.0
9/21/2010
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MP24875 – 1.5A, 200kHz, 55V HIGH-POWER LEDS DRIVER
Output Capacitor COUT
The output capacitor (COUT) is required to reduce
the LED current ripple. Ceramic, tantalum, or low
ESR electrolytic capacitors are recommended.
Low ESR capacitors are preferred to keep the
output voltage ripple low so that the AC ripple
current through the LEDs is small. The output
voltage ripple can be estimated by:
∆VOUT =
VOUT
8 × fS2 × L1× COUT
⎛
⎞
V
× ⎜ 1 − OUT ⎟
VIN ⎠
⎝
For most application, a 2.2µF~4.7µF ceramic
capacitor is recommended.
Compensation Components
MP24875 employs current mode control for easy
compensation and fast transient response. The
system stability and transient response are
controlled through the COMP pin. COMP pin is
the output of the internal error amplifier. A series
capacitor-resistor combination (RCOM and CCOM1)
sets a pole-zero combination to control the
characteristics of the control system. The DC
gain of the current feedback loop is given by:
A VDC = RFB × GCS × A VEA
Where AVEA is the error amplifier voltage gain,
is
the
current
sense
400V/V;
GCS
transconductance, 8A/V; RFB is the current
sensing resistor value.
The system has two poles of importance. One is
due to the compensation capacitor (CCOM1) and
the
output
resistor
of
error
amplifier
(REA=AVEA/GEA). GEA is the error amplifier
transconductance, 500µA/V. The other is due to
the output capacitor and the LEDs’ AC resistor
(RLED=∆VOUT/∆ILED). These poles are located at:
fP1 =
fP2 =
1
2π × CCOM1 × REA
1
2π × COUT × RLED
The system has one zero of importance, due to
the compensation capacitor (CCOM1) and the
compensation resistor (RCOM). This zero is
located at:
MP24875 Rev. 1.0
9/21/2010
fZ1 =
1
2π × CCOM1 × RCOM
The system may have another zero of
importance, if the output capacitor has a large
capacitance and/or a high ESR value. The zero,
due to the ESR and capacitance of the output
capacitor, is located at:
fESR =
1
2π × COUT × RESR
In this case, a third pole set by the compensation
capacitor (CCOM2) and the compensation resistor
(RCOM) is used to compensate the effect of the
ESR zero on the loop gain. This pole is located
at:
fP3 =
1
2π × CCOM2 × RCOM
The goal of compensation design is to shape the
converter transfer function to get a desired loop
gain and phase margin. The system crossover
frequency where the feedback loop has the unity
gain is important. Lower crossover frequencies
result in slower line and load transient responses,
while higher crossover frequencies could cause
system unstable. A good rule of thumb is to set
the crossover frequency to approximately onetenth of the switching frequency. To optimize the
compensation components for conditions, the
following procedure can be used.
1. Choose the compensation resistor (RCOM) to
set the desired crossover frequency. Determine
the RCOM value by the following equation:
RCOM =
2π × COUT × RLED × fC
RFB × GEA × GCS
Where fC is the desired crossover frequency.
2. Choose the compensation capacitor (CCOM1) to
achieve the desired phase margin. For
applications with typical inductor values, setting
the compensation zero, fZ1, below one forth of the
crossover frequency provides sufficient phase
margin. Determine the CCOM1 value by the
following equation:
CCOM1 >
4
2π × RCOM × fC
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MP24875 – 1.5A, 200kHz, 55V HIGH-POWER LEDS DRIVER
3. Determine if the second compensation
capacitor (CCOM2) is required, which is connected
from COMP pin to GND. It is required if the ESR
zero of the output capacitor is located at less
than half of the switching frequency:
1
2π × COUT × RESR
<
fS
2
If this is the case, then add the second
compensation capacitor (CCOM2) to set the pole
fP3 at the location of the ESR zero. Determine the
CCOM2 value by the equation:
CCOM2 =
COUT × RESR
RCOM
external BST diode is recommended from the 5V
to BST pin:
z
There is a 5V rail available in the system;
z
VIN is not greater than 5V;
z
VOUT is between 3.3V and 5V;
The bootstrap diode can be a low cost one such
as IN4148 or BAT54.
MP24875
External Bootstrap Diode
An external bootstrap diode may enhance the
efficiency of the regulator. In below cases, an
MP24875 Rev. 1.0
9/21/2010
Figure 2—External Bootstrap Diode
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MP24875 – 1.5A, 200kHz, 55V HIGH-POWER LEDS DRIVER
TYPICAL APPLICATION CIRCUITS
MP24875
Figure 3—700mA WLED Driver Application
MP24875 Rev. 1.0
9/21/2010
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MP24875 – 1.5A, 200kHz, 55V HIGH-POWER LEDS DRIVER
PACKAGE INFORMATION
SOIC8 (EXPOSED PAD)
0.189(4.80)
0.197(5.00)
0.124(3.15)
0.136(3.45)
8
5
0.150(3.80)
0.157(4.00)
PIN 1 ID
1
0.228(5.80)
0.244(6.20)
0.089(2.26)
0.101(2.56)
4
TOP VIEW
BOTTOM VIEW
SEE DETAIL "A"
0.051(1.30)
0.067(1.70)
SEATING PLANE
0.000(0.00)
0.006(0.15)
0.013(0.33)
0.020(0.51)
0.0075(0.19)
0.0098(0.25)
SIDE VIEW
0.050(1.27)
BSC
FRONT VIEW
0.010(0.25)
x 45o
0.020(0.50)
GAUGE PLANE
0.010(0.25) BSC
0.050(1.27)
0.024(0.61)
0o-8o
0.016(0.41)
0.050(1.27)
0.063(1.60)
DETAIL "A"
0.103(2.62)
0.213(5.40)
NOTE:
0.138(3.51)
RECOMMENDED LAND PATTERN
1) CONTROL DIMENSION IS IN INCHES. DIMENSION IN
BRACKET IS IN MILLIMETERS.
2) PACKAGE LENGTH DOES NOT INCLUDE MOLD FLASH,
PROTRUSIONS OR GATE BURRS.
3) PACKAGE WIDTH DOES NOT INCLUDE INTERLEAD FLASH
OR PROTRUSIONS.
4) LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING)
SHALL BE 0.004" INCHES MAX.
5) DRAWING CONFORMS TO JEDEC MS-012, VARIATION BA.
6) DRAWING IS NOT TO SCALE.
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.
MP24875 Rev. 1.0
9/21/2010
www.MonolithicPower.com
MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
© 2010 MPS. All Rights Reserved.
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