MP2365
3A, 28V, 1.4MHz
Step-Down Converter
The Future of Analog IC Technology
DESCRIPTION
FEATURES
The MP2365 is a 1.4MHz step-down regulator
with a built-in power MOSFET. It achieves 3A
continuous output current over a wide input
supply range with excellent load and line
regulation.
Current mode operation provides fast transient
response and eases loop stabilization.
Fault condition protection includes cycle-bycycle current limiting and thermal shutdown.
Adjustable soft-start reduces the stress on the
input source at turn-on. In shutdown mode, the
regulator draws 20μA of supply current.
The MP2365 is available in an 8-pin SOIC
package with an exposed pad, and requires a
minimum number of readily available external
components to complete a 3A step-down DC to
DC converter solution.
APPLICATIONS
EVALUATION BOARD REFERENCE
Board Number
Dimensions
EV2365DN-00A
2.0” x 1.9” x 0.4”
3A Continuous Output Current, 4A Peak
Output Current
Programmable Soft-Start
100mΩ Internal Power MOSFET Switch
Stable with Low ESR Output Ceramic
Capacitors
Up to 91% Efficiency
20μA Shutdown Mode
Fixed 1.4MHz Frequency
Thermal Shutdown
Cycle-by-Cycle Over Current Protection
Wide 4.75V to 28V Operating Input Range
Output is Adjustable From 0.92V to 21V
Under Voltage Lockout
Distributed Power Systems
Battery Chargers
Pre-Regulator for Linear Regulators
All MPS parts are lead-free and adhere to the RoHS directive. For MPS green
status, please visit MPS website under Quality Assurance. “MPS” and “The
Future of Analog IC Technology” are Registered Trademarks of Monolithic
Power Systems, Inc.
TYPICAL APPLICATION
Efficiency Curve
INPUT
100
C1
CERAMIC
8
IN
BS
SW
EN
MP2365
SS
GND
FB
COMP
4
C4
6
C6
OPEN
3
L1
5
R1
D1
C3
OUTPUT
3.3V
3A
R2
C2
CERAMIC
VOUT=5V
EFFICIENCY (%)
7
OPEN =
AUTOMATIC
STARTUP
C5
1
2
90
70
R3
VIN=12V
60
MP2365 Rev. 1.01
2/26/2020
VOUT=3.3V
80
0
0.5 1.0 1.5 2.0 2.5
LOAD CURRENT (A)
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3.0
1
MP2365 – 3A, 28V, 1.4MHz STEP-DOWN CONVERTER
ORDERING INFORMATION
Part Number*
Package
Top Marking
Free Air Temperature (TA)
MP2365DN
SOIC8N
MP2365DN
–40C to +85C
* For Tape & Reel, add suffix –Z (eg. MP2365DN–Z).
For RoHS compliant packaging, add suffix –LF (eg. MP2365DN–LF–Z)
PACKAGE REFERENCE
TOP VIEW
BS
1
8
SS
IN
2
7
EN
SW
3
6
COMP
GND
4
5
FB
EXPOSED PAD
CONNECT TO PIN 4
ABSOLUTE MAXIMUM RATINGS (1)
Supply Voltage VIN ....................... –0.3V to +30V
Switch Voltage VSW .............. –0.5V to VIN + 0.3V
Boost Voltage VBS .......... VSW – 0.3V to VSW + 6V
All Other Pins ................................. –0.3V to +6V
Continuous Power Dissipation (TA = +25°C) (2)
………………………………………..……....2.5W
Junction Temperature ............................... 150C
Lead Temperature .................................... 260C
Storage Temperature .............. –65C to +150C
Recommended Operating Conditions (3)
Input Voltage VIN ............................ 4.75V to 28V
Ambient Operating Temp........... –40C to +85C
MP2365 Rev. 1.01
2/26/2020
Thermal Resistance (4)
θJA
θJC
SOIC8N ................................... 50 ....... 10 ... C/W
Notes:
1) Exceeding these ratings may damage the device.
2) The maximum allowable power dissipation is a function of the
maximum junction temperature TJ (MAX), the junction-toambient thermal resistance θJA, and the ambient temperature
TA. The maximum allowable continuous power dissipation at
any ambient temperature is calculated by PD (MAX) = (TJ
(MAX)-TA)/θJA. Exceeding the maximum allowable power
dissipation will cause excessive die temperature, and the
regulator will go into thermal shutdown. Internal thermal
shutdown circuitry protects the device from permanent
damage.
3) The device is not guaranteed to function outside of its
operating conditions.
4) Measured on JESD51-7, 4-layer PCB.
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2
MP2365 – 3A, 28V, 1.4MHz STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS
VIN = 12V, TA = +25C, unless otherwise noted.
Parameters
Shutdown Supply Current
Supply Current
Feedback Voltage
Error Amplifier Voltage Gain
Error Amplifier Transconductance
High-Side Switch-On Resistance
Low-Side Switch-On Resistance
High-Side Switch Leakage Current
Current Limit (5)
Current Sense to COMP Transconductance
Oscillation Frequency
Short Circuit Oscillation Frequency
Maximum Duty Cycle
Minimum On Time
EN Shutdown Threshold Voltage
Enable Pull Up Current
EN Lockout Threshold Voltage
EN Lockout Hysteresis
Symbol Condition
VEN = 0V
VEN = 3V, VFB =1.4V
4.75V VIN 28V,
VFB
VCOMP < 2V
AVEA
GEA
ICOMP = 10μA
RDS(ON)1
RDS(ON)2
VEN = 0V, VSW = 0V
Min
Soft-Start Period
Thermal Shutdown
CSS = 0.1µF, L=4.7μH
CO=47μF, IO=3A(CR)
Units
µA
mA
V
400
530
100
10
0.1
6.5
6.0
1.4
180
65
130
1.2
1.6
2.6
210
V/V
µA/V
mΩ
Ω
µA
A
A/V
MHz
kHz
%
ns
V
µA
V
mV
330
VFB = 0V
VFB = 0.8V
VEN = 0V
Max
30
1.5
0.892 0.920 0.948
GCS
fS
DMAX
TON
Typ
20
1.3
0.9
0.9
2.3
730
10
1.5
2.3
2.9
5
ms
160
C
Note:
5) Slope compensation changes current limit above 40% duty cycle.
MP2365 Rev. 1.01
2/26/2020
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3
MP2365 – 3A, 28V, 1.4MHz STEP-DOWN CONVERTER
PIN FUNCTIONS
Pin #
1
2
3
4
5
6
7
8
Name
Description
High-Side Gate Drive Boost Input. BS supplies the drive for the high-side N-Channel MOSFET
BS
switch. Connect a 10nF or greater capacitor from SW to BS to power the high side switch.
Power Input. IN supplies the power to the IC, as well as the step-down converter switches.
IN
Drive IN with a 4.75V to 28V power source. Bypass IN to GND with a suitably large capacitor
to eliminate noise on the input to the IC. See Input Capacitor
Power Switching Output. SW is the switching node that supplies power to the output. Connect
SW
the output LC filter from SW to the output load. Note that a capacitor is required from SW to BS
to power the high-side switch.
GND Ground. Connect the exposed pad on backside to Pin 4.
Feedback Input. FB senses the output voltage to regulate said voltage. Drive FB with a
FB
resistive voltage divider from the output voltage. The feedback threshold is 0.92V. See Setting
the Output Voltage
Compensation Node. COMP is used to compensate the regulation control loop. Connect a
COMP series RC network from COMP to GND to compensate the regulation control loop. In some
cases, an additional capacitor from COMP to GND is required. See Compensation
Enable Input. EN is a digital input that turns the regulator on or off. Drive EN higher than 2.9V
EN
to turn on the regulator, lower than 0.9V to turn it off. For automatic startup, leave EN
unconnected.
Soft-Start Control Input. SS controls the soft start period. Connect a capacitor from SS to GND
SS
to set the soft-start period. A 0.1μF capacitor sets the soft-start period to 5ms.
MP2365 Rev. 1.01
2/26/2020
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4
MP2365 – 3A, 28V, 1.4MHz STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS
VIN = 12V, C1 = 10µF, C2 = 47µF, L = 4.7µH and TA = +25C, unless otherwise noted.
Steady State Operation
VOUT=3.3V, IOUT = 3A
8.0
VOUT
10mV/div.
7.5
7.0
VIN
10V/div.
VSW
10V/div.
6.5
6.0
5.5
5.0
0
20
40
60
IINDUCTOR
2A/div.
80
VOUT
2V/div.
VOUT
2V/div.
VEN
10V/div.
VOUT
2V/div.
VEN
10V/div.
VEN
10V/div.
VSW
10V/div.
VSW
10V/div.
IINDUCTOR
1A/div.
IINDUCTOR
2A/div.
VOUT=3.3V, IOUT = 3A
VOUT=3.3V, IOUT = 0A
VOUT=3.3V, IOUT = 3A (CR)
VSW
10V/div.
Shutdown Through Enable
Startup Through Enable
Startup Through Enable
400ns/div
IINDUCTOR
2A/div.
4ms/div.
Shutdown Through Enable
Load Transient Response
SHORT
VOUT=3.3V, IOUT = 1.5A ~3A
VOUT=3.3V, IOUT = 0A
VOUT
1V/div.
VOUT
2V/div.
VOUT
50mV/div.
VIN
10V/div.
VSW
10V/div.
VEN
10V/div.
VSW
10V/div.
IINDUCTOR
1A/div.
IINDUCTOR
5A/div.
IINDUCTOR
1A/div.
1s/div.
MP2365 Rev. 1.01
2/26/2020
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5
MP2365 – 3A, 28V, 1.4MHz STEP-DOWN CONVERTER
OPERATION
The converter uses an internal N-Channel
MOSFET switch to step-down the input voltage
to the regulated output voltage. Since the
MOSFET requires a gate voltage greater than
the input voltage, a boost capacitor connected
between SW and BS drives the gate. The
capacitor is internally charged while SW is low.
The MP2365 is a current-mode step-down
regulator. It regulates input voltages from 4.75V to
28V down to an output voltage as low as 0.92V,
and is able to supply up to 3A of load current.
The MP2365 uses current-mode control to
regulate the output voltage. The output voltage is
measured at FB through a resistive voltage
divider and amplified through the internal error
amplifier.
The
output
current
of
the
transconductance error amplifier is presented at
COMP where a network compensates the
regulation control system. The voltage at COMP
is compared to the switch current measured
internally to control the output voltage.
An internal 10Ω switch from SW to GND is used
to insure that SW is pulled to GND when SW is
low to fully charge the BS capacitor.
IN 2
CURRENT
SENSE
AMPLIFIER
INTERNAL
REGULATORS
OSCILLATOR
180KHz/
1.4MHz
1.2V
--
EN 7
-2.60V/
2.39V
SLOPE
COMP
5V
--
CLK
+
+
+
SHUTDOWN
COMPARATOR
--
1 BS
S
Q
R
Q
3 SW
CURRENT
COMPARATOR
LOCKOUT
COMPARATOR
+
--
+
4 GND
1.8V
FREQUENCY
FOLDBACK
COMPARATOR
--
0.6V
0.92V
5
FB
+
ERROR
AMPLIFIER
6
COMP
8
SS
Figure 1—Functional Block Diagram
MP2365 Rev. 1.01
2/26/2020
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6
MP2365 – 3A, 28V, 1.4MHz STEP-DOWN CONVERTER
APPLICATION INFORMATION
COMPONENT SELECTION
(Refer to “Typical Application on Page 1”
Setting the Output Voltage
The output voltage is set using a resistive
voltage divider from the output voltage to FB pin.
The voltage divider divides the output voltage
down to the feedback voltage by the ratio:
VFB VOUT
R2
R1 R2
Where VFB is the feedback voltage and VOUT is
the output voltage.
Thus the output voltage is:
VOUT 0.92
R1 R2
R2
A typical value for R2 can be as high as 100kΩ,
but a typical value is 10kΩ. Using that value, R1
is determined by:
R1 10.87 ( VOUT 0.92)(k )
Inductor
The inductor is required to supply constant
current to the output load while being driven by
the switched input voltage. A larger value
inductor will result in less ripple current that will
result in lower output ripple voltage. However,
the larger value inductor will have a larger
physical size, higher series resistance, and/or
lower saturation current. A good rule for
determining the inductance to use is to allow
the peak-to-peak ripple current in the inductor
to be approximately 30% of the maximum
switch current limit. Also, make sure that the
peak inductor current is below the maximum
switch current limit. The inductance value can
be calculated by:
L1
VOUT
V
1 OUT
fS ∆IL
VIN
Where VIN is the input voltage, fS is the 1.4MHz
switching frequency and ∆IL is the peak-to-peak
inductor ripple current.
Choose an inductor that will not saturate under
the maximum inductor peak current. The peak
inductor current can be calculated by:
ILP ILOAD
MP2365 Rev. 1.01
2/26/2020
Where ILOAD is the load current.
Table 1 lists a number of suitable inductors
from various manufacturers. The choice of
which style inductor to use mainly depends on
the price vs. size requirements and any EMI
requirement.
Table 1—Inductor Selection Guide
Vendor/
Model
Core
Type
Core
Material
Package
Dimensions
(mm)
W
L
H
Sumida
CR75
Open
Ferrite
7.0
7.8
5.5
CDH74
Open
Ferrite
7.3
8.0
5.2
CDRH5D28 Shielded
Ferrite
5.5
5.7
5.5
CDRH5D28 Shielded
Ferrite
5.5
5.7
5.5
CDRH6D28 Shielded
Ferrite
6.7
6.7
3.0
CDRH104R Shielded
Ferrite
10.1 10.0
3.0
Toko
D53LC
Type A
Shielded
Ferrite
5.0
5.0
3.0
D75C
Shielded
Ferrite
7.6
7.6
5.1
D104C
Shielded
Ferrite
10.0 10.0
4.3
D10FL
Open
Ferrite
9.7
1.5
4.0
DO3308
Open
Ferrite
9.4
13.0
3.0
DO3316
Open
Ferrite
9.4
13.0
5.1
Coilcraft
Output Rectifier Diode
The output rectifier diode supplies the current to
the inductor when the high-side switch is off. To
reduce losses due to the diode forward voltage
and recovery times, use a Schottky diode.
Choose a diode whose maximum reverse
voltage rating is greater than the maximum
input voltage, and whose current rating is
greater than the maximum load current. Table 2
lists
example
Schottky
diodes
and
manufacturers.
Input Capacitor
The input current to the step-down converter is
discontinuous, therefore a capacitor is required
to supply the AC current to the step-down
VOUT
V
1 OUT
2 fS L
VIN
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7
MP2365 – 3A, 28V, 1.4MHz STEP-DOWN CONVERTER
Table 2—Diode Selection Guide
Voltage/Current Manufacture
Rating
Diode
SK33
SK34
B330
B340
MBRS330
MBRS340
30V, 3A
40V, 3A
30V, 3A
40V, 3A
30V, 3A
40V, 3A
Diodes Inc.
Diodes Inc.
Diodes Inc.
Diodes Inc.
On Semiconductor
On Semiconductor
converter while maintaining the DC input
voltage. Use low ESR capacitors for the best
performance. Ceramic capacitors are preferred,
but tantalum or low-ESR electrolytic capacitors
may also suffice.
Since the input capacitor (C1) absorbs the input
switching current, it requires an adequate ripple
current rating. The RMS current in the input
capacitor can be estimated by:
I C1 ILOAD
VOUT VOUT
1
VIN
VIN
The worst-case condition occurs at VIN = 2VOUT,
where:
IC1
ILOAD
2
For simplification, choose the input capacitor
whose RMS current rating greater than half of
the maximum load current.
The input capacitor can be electrolytic, tantalum
or ceramic. When using electrolytic or tantalum
capacitors, a small, high quality ceramic
capacitor, i.e. 0.1μF, should be placed as close
to the IC as possible. When using ceramic
capacitors, make sure that they have enough
capacitance to provide sufficient charge to
prevent excessive voltage ripple at input. The
input voltage ripple caused by capacitance can
be estimated by:
VIN
ILOAD
V
V
OUT 1 OUT
fS C1 VIN
VIN
Output Capacitor
The output capacitor (C2) is required to
maintain the DC output voltage. Ceramic,
tantalum, or low ESR electrolytic capacitors are
recommended. Low ESR capacitors are
preferred to keep the output voltage ripple low.
The output voltage ripple can be estimated by:
MP2365 Rev. 1.01
2/26/2020
VOUT
VOUT
V
1 OUT
fS L
VIN
1
R ESR
8 f S C2
Where L is the inductor value and RESR is the
equivalent series resistance (ESR) value of the
output capacitor.
In the case of ceramic capacitors, the
impedance at the switching frequency is
dominated by the capacitance. The output
voltage ripple is mainly caused by the
capacitance. For simplification, the output
voltage ripple can be estimated by:
∆VOUT
V
1 OUT
VIN
L C2
VOUT
8 fS
2
In the case of tantalum or electrolytic capacitors,
the ESR dominates the impedance at the
switching frequency. For simplification, the
output ripple can be approximated to:
∆VOUT
VOUT
V
1 OUT
fS L
VIN
R ESR
The characteristics of the output capacitor also
affect the stability of the regulation system. The
MP2365 can be optimized for a wide range of
capacitance and ESR values.
Compensation Components
MP2365 employs current mode control for easy
compensation and fast transient response. The
system stability and transient response are
controlled through the COMP pin. COMP pin is
the output of the internal transconductance
error amplifier. A series capacitor-resistor
combination sets a pole-zero combination to
control the characteristics of the control system.
The DC gain of the voltage feedback loop is
given by:
A VDC R LOAD G CS A VEA
VFB
VOUT
Where AVEA is the error amplifier voltage gain, GCS
is the current sense transconductance and RLOAD
is the load resistor value.
The system has two poles of importance. One is
due to the compensation capacitor (C3) and the
output resistor of error amplifier, and the other is
due to the output capacitor and the load resistor.
These poles are located at:
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8
MP2365 – 3A, 28V, 1.4MHz STEP-DOWN CONVERTER
fP1
GEA
2 C3 A VEA
Table 3—Compensation Values for Typical
Output Voltage/Capacitor Combinations
fP 2
1
2 C2 R LOAD
VOUT
(V)
Where GEA is the error amplifier transconductance,
530μA/V.
The system has one zero of importance, due to
the compensation capacitor (C3) and the
compensation resistor (R3). This zero is located
at:
f Z1
1
2 C3 R3
The system may have another zero of importance,
if the output capacitor has a large capacitance
and/or a high ESR value. The zero, due to the
ESR and capacitance of the output capacitor, is
located at:
fESR
1
2 C2 R ESR
In this case, a third pole set by the compensation
capacitor (C6) and the compensation resistor (R3)
is used to compensate the effect of the ESR zero
on the loop gain. This pole is located at:
fP3
1
2 C6 R3
The goal of compensation design is to shape
the converter transfer function to get a desired
loop gain. The system crossover frequency
where the feedback loop has the unity gain is
important.
Lower crossover frequencies result in slower
line and load transient responses, while higher
crossover frequencies could cause system
unstable. A good rule of thumb is to set the
crossover frequency to approximately one-tenth
of the switching frequency or lower. The
switching frequency for the MP2365 is 1.4MHz,
so the desired crossover frequency is equal to
or less than 140kHz.
Table 3 lists the typical values of compensation
components for some standard output voltages
with various output capacitors and inductors.
The values of the compensation components
have been optimized for fast transient
responses and good stability at given conditions.
MP2365 Rev. 1.01
2/26/2020
L (µH)
C2 (µF,
Ceramic)
R3
(kΩ)
C3
(nF)
C6
1.8
2.2
47
7.5
3.3
None
2.5
2.2 - 4.7
47
10
4.7
None
3.3
2.2 - 4.7
47
15
5.6
None
5
4.7 – 6.8
2 x 22
20
4.7
None
12
6.8 - 10
2 x 22
44.2
2.2
None
To optimize the compensation components for
conditions not listed in Table 3, the following
procedure can be used.
1. Choose the compensation resistor (R3) to set
the desired crossover frequency. Determine the
R3 value by the following equation:
R3
2 C2 f C VOUT
G EA G CS
VFB
Where fC is the desired crossover frequency.
2. Choose the compensation capacitor (C3) to
achieve the desired phase margin. For
applications with typical inductor values, setting
the compensation zero, fZ1, below one forth of the
crossover frequency provides sufficient phase
margin. Determine the C3 value by the following
equation:
C3
4
2 R3 f C
3. Determine if the second compensation
capacitor (C6) is required. It is required if the ESR
zero of the output capacitor is located at less than
half of the 1.4MHz switching frequency, or the
following relationship is valid:
f
1
S
2 C2 R ESR
2
If this is the case, then add the second
compensation capacitor (C6) to set the pole fP3
at the location of the ESR zero. Determine the
C6 value by the equation:
C6
C2 R ESR
R3
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9
MP2365 – 3A, 28V, 1.4MHz STEP-DOWN CONVERTER
PCB Layout Guide
PCB layout is very important to achieve stable
operation. Please follow these guidelines and
take Figure2 and 3 for references.
1) Keep the path of switching current short and
minimize the loop area formed by Input cap,
high-side MOSFET and schottky diode.
6
4)
Route SW away from sensitive analog areas
such as FB.
5)
Connect IN, SW, and especially GND
respectively to a large copper area to cool
the chip to improve thermal performance and
long-term reliability. For single layer, do not
solder exposed pad of the IC.
FB 5
COMP
SW
Ensure all feedback connections are short
and direct. Place the feedback resistors and
compensation components as close to the
chip as possible.
4 GND
3
EN 7
IN
C5
1 BST
C4
SS/REF 8
Keep the connection of schottky diode
between SW pin and input power ground as
short and wide as possible.
2
2)
3)
Figure2―PCB Layout for Single Layer
SW
3
FB 5
SGND
SGND
Vout
Feeback
4 GND
6
IN
2
COMP
BST
1
EN 7
R3
C6
R1
C5
C3
R2
C4
SS/REF 8
R4
L1
C1
D1
C2
PGND
Top Layer
Bottom Layer
Figure3―PCB Layout for Double Layer
MP2365 Rev. 1.01
2/26/2020
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10
MP2365 – 3A, 28V, 1.4MHz STEP-DOWN CONVERTER
External Bootstrap Diode
An external bootstrap diode may enhance the
efficiency of the regulator, the applicable
conditions of external BST diode are:
VOUT=5V or 3.3V;
V
Duty cycle is high: D OUT 65%
VIN
In these cases, an external BST diode is
recommended from the output of the voltage
regulator to BST pin, as shown in Fig.4.
External BST Diode
IN4148
BST
MP2365
SW
CBST
L
5V or 3.3V
COUT
Figure 4—External Bootstrap Diode
The recommended external BST diode is IN4148,
and the BST cap is 0.1~1µF.
MP2365 Rev. 1.01
2/26/2020
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11
MP2365 – 3A, 28V, 1.4MHz STEP-DOWN CONVERTER
TYPICAL APPLICATION CIRCUITS
C5
10nF
INPUT
4.75V to 28V
OPEN = AUTOMATIC
STARTUP
2
7
1
BS
3
SW
IN
EN
OUTPUT
2.5V
3A
MP2365
8
SS
GND
FB
COMP
4
5
6
C6
C3
4.7nF
OPEN
D1
B330A
Figure 5 —2.5V Output Typical Application Schematic
C5
10nF
INPUT
4.75V to 28V
OPEN = AUTOMATIC
STARTUP
2
7
1
BS
3
SW
IN
EN
OUTPUT
3.3V
3A
MP2365
8
SS
GND
FB
COMP
4
5
6
C6
OPEN
C3
5.6nF
D1
B330A
Figure 6—3.3V Output Typical Application Schematic
MP2365 Rev. 1.01
2/26/2020
www.MonolithicPower.com
MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited.
© 2020 MPS. All Rights Reserved.
12
MP2365 – 3A, 28V, 1.4MHz STEP-DOWN CONVERTER
PACKAGE INFORMATION
SOIC8N (EXPOSED PAD)
0.189(4.80)
0.197(5.00)
8
0.124(3.15)
0.136(3.45)
5
0.150(3.80)
0.157(4.00)
PIN 1 ID
1
0.228(5.80)
0.244(6.20)
0.089(2.26)
0.101(2.56)
4
TOP VIEW
BOTTOM VIEW
SEE DETAIL "A"
0.013(0.33)
0.020(0.51)
0.051(1.30)
0.067(1.70)
SEATING PLANE
0.000(0.00)
0.006(0.15)
0.0075(0.19)
0.0098(0.25)
SIDE VIEW
0.050(1.27)
BSC
FRONT VIEW
0.010(0.25)
x 45o
0.020(0.50)
GAUGE PLANE
0.010(0.25) BSC
0.024(0.61)
0.050(1.27)
0o-8o
0.016(0.41)
0.050(1.27)
0.063(1.60)
DETAIL "A"
0.103(2.62)
0.138(3.51)
RECOMMENDED LAND PATTERN
0.213(5.40)
NOTE:
1) CONTROL DIMENSION IS IN INCHES. DIMENSION IN
BRACKET IS IN MILLIMETERS.
2) PACKAGE LENGTH DOES NOT INCLUDE MOLD FLASH
,
PROTRUSIONS OR GATE BURRS.
3) PACKAGE WIDTH DOES NOT INCLUDE INTERLEAD FLASH
OR PROTRUSIONS.
4) LEAD COPLANARITY(BOTTOM OF LEADS AFTER FORMING)
SHALL BE 0.004" INCHES MAX.
5) DRAWING CONFORMS TO JEDEC MS-012, VARIATION BA.
6) DRAWING IS NOT TO SCALE.
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.
MP2365 Rev. 1.01
2/26/2020
www.MonolithicPower.com
MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited.
© 2020 MPS. All Rights Reserved.
13
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