MPQ4459
Industrial Grade,1.5A, 4MHz, 36V
Step-Down Converter
The Future of Analog IC Technology
DESCRIPTION
FEATURES
The MPQ4459 is a high frequency step-down
switching regulator with an integrated internal
high-side high voltage power MOSFET. It
provides 1.5A output with current mode control
for fast loop response and easy compensation.
The wide 3.8V to 36V input range
accommodates a variety of step-down
applications, including those in automotive
systems. A 120µA operational quiescent current
is suitable for use in battery-powered
applications.
The frequency foldback helps prevent inductor
current runaway during startup and thermal
shutdown provides reliable, fault tolerant
operation.
By switching at 4MHz, the MPQ4459 prevents
EMI (Electromagnetic Interference) noise
problems, such as those found in AM radio and
ADSL applications.
The MPQ4459 is available in thin 10-pin 3mm x
3mm TQFN package.
Guaranteed Industrial Temp Range
120μA Quiescent Current
Wide 3.8V to 36V Operating Input Range
150mΩ Internal Power MOSFET
Up to 4MHz Programmable Switching
Frequency
Ceramic Capacitor Stable
Internal Soft-Start
Precision Current Limit without a Current
Sensing Resistor
Up to 95% Efficiency
Output Adjustable from 0.8V to 30V
Available in 10-Pin 3x3 TQFN Package
APPLICATIONS
High Voltage Power Conversion
Automotive Systems
Industrial Power Systems
Distributed Power Systems
Battery Powered Systems
All MPS parts are lead-free and adhere to the RoHS directive. For MPS green
status, please visit MPS website under Quality Assurance. “MPS” and “The
Future of Analog IC Technology” are Registered Trademarks of Monolithic
Power Systems, Inc.
TYPICAL APPLICATION
CONTROL
100
4
3
7
5
COMP
EN
VIN
BST
MPQ4459
FREQ
FB
SW
GND
8, 9
VIN
4.5V to 36V
10
1, 2
6
C4
100nF
10MQ100N
VOUT
5V @ 1.5A
VI=5V
90
EFFICIENCY (%)
C3
150pF
Efficiency vs
Load Current
80
VI=24V
VI=12V
70
60
50
40
VO=3.3V
30
20
0
500
1000
LOAD CURRENT (mA)
MPQ4459 Rev. 1.01
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1500
1
MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
ORDERING INFORMATION
Part Number*
Package
Top Marking
MPQ4459DQT
TQFN10(3mm x 3mm)
N4
* For Tape & Reel, add suffix –Z (eg. MPQ4459DQT–Z)
For RoHS compliant packaging, add suffix –LF(eg. MPQ4459DQT–LF–Z)
PACKAGE REFERENCE
TOP VIEW
SW
1
10
BST
SW
2
9
VIN
EN
3
8
VIN
COMP
4
7
FREQ
FB
5
6
GND
EXPOSED PAD
ON BACKSIDE
ABSOLUTE MAXIMUM RATINGS (1)
Thermal Resistance
Supply Voltage (VIN).....................–0.3V to +40V
Switch Voltage (VSW)............ –0.3V to VIN + 0.3V
BST to SW .....................................–0.3V to +6V
All Other Pins .................................–0.3V to +6V
Continuous Power Dissipation
(TA = +25°C)(2)
……………………………………………….2.5W
Junction Temperature ...............................150C
Lead Temperature ....................................260C
Storage Temperature.............. –65°C to +150C
3x3 TQFN10 ...........................50 ...... 12 ... C/W
Recommended Operating Conditions
(3)
Supply Voltage VIN ...........................3.8V to 36V
Output Voltage VOUT .........................0.8V to 30V
Operating Junct. Temp (TJ)..... –40C to +125C
(4)
θJA
θJC
Notes:
1) Exceeding these ratings may damage the device
2) The maximum allowable power dissipation is a function of the
maximum junction temperature TJ(MAX), the junction-toambient thermal resistance θJA, and the ambient temperature
TA. The maximum allowable continuous power dissipation at
any
ambient
temperature
is
calculated
by
PD(MAX)=(TJ(MAX)-TA)/ θJA. Exceeding the maximum
allowable power dissipation will cause excessive die
temperature, and the regulator will go into thermal shutdown.
Internal thermal shutdown circuitry protects the device from
permanent damage.
3) The device is not guaranteed to function outside of its
operating conditions.
4) Measured on JESD51-7, 4-layer PCB.
MPQ4459 Rev. 1.01
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MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS
VIN = 12V, VEN = 2.5V, VCOMP = 1.4V, TJ =–40C to +85C, unless otherwise noted. Typical values are
at TJ =25C.
Parameter
Symbol Condition
TJ =25C
Feedback Voltage
VFB
4.5V < VIN < 36V
Feedback Bias Current
IFB
VFB = 0.8V
VBST – VSW = 5V
VEN = 0V, VSW = 0V, VIN = 36V
Duty Cycle = 50%
Upper Switch On Resistance
Upper Switch Leakage
Current Limit
COMP to Current Sense
Transconductance
Error Amp Voltage Gain (5)
Error Amp Transconductance
Error Amp Min Source current
Error Amp Min Sink current
VIN UVLO Threshold
VIN UVLO Hysteresis
Soft-Start Time (5)
RDS(ON)
Min
Typ
Max
Units
0.776
0.8
0.824
V
0.770
0.01
1.7
GCS
ICOMP = ±3µA
VFB = 0.7V
VFB = 0.9V
20
2.55
Oscillator Frequency
fS
Shutdown Supply Current
Quiescent Supply Current
IQ
0V < VFB < 0.8V
RFREQ = 45kΩ
RFREQ = 18kΩ
VEN = 0V
No load, VFB = 0.9V
1.55
3.1
0.830
V
1.0
μA
150
1
2.5
mΩ
μA
A
4.7
A/V
200
60
5
–5
3.0
0.35
1.5
2
4
12
120
V/V
µA/V
µA
µA
V
V
ms
MHz
MHz
µA
µA
100
3.45
2.45
4.9
18
165
Thermal Shutdown
150
C
Thermal Shutdown Hysteresis
15
Minimum Off Time
Minimum On Time (5)
EN Up Threshold
EN Threshold Hysteresis
100
100
1.5
300
C
ns
ns
V
mV
1.2
1.8
Note:
5) Guaranteed by design.
MPQ4459 Rev. 1.01
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MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
PIN FUNCTIONS
Pin #
1, 2
3
4
5
6
7
8, 9
10
Name
Description
Switch Node. This is the output from the high-side switch. A low Vf Schottky rectifier to ground
SW
is required. The rectifier must be close to the SW pins to reduce switching spikes.
Enable Input. Pulling this pin below the specified threshold shuts the chip down. Pulling it up
EN
above the specified threshold or leaving it floating enables the chip.
Compensation. This node is the output of the GM error amplifier. Control loop frequency
COMP
compensation is applied to this pin.
Feedback. This is the input to the error amplifier. An external resistive divider connected
FB
between the output and GND is compared to the internal +0.8V reference to set the regulation
voltage.
Ground. It should be connected as close as possible to the output capacitor avoiding the high
GND
current switch paths.
Switching Frequency Program Input. Connect a resistor from this pin to ground to set the
FREQ
switching frequency.
Input Supply. This supplies power to all the internal control circuitry, both BS regulators and
VIN
the high-side switch. A decoupling capacitor to ground must e placed close to this pin to
minimize switching spikes.
Bootstrap. This is the positive power supply for the internal floating high-side MOSFET driver.
BST
Connect a bypass capacitor between this pin and SW pin.
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MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CURVES
VIN = 12V, VOUT = 5V, fS = 500kHz, TA = +25C, unless otherwise noted.
Efficiency vs
Load Current
Efficiency vs
Load Current
Efficiency vs
Load Current
100
100
100
90
90
90
80
80
80
70
70
70
60
60
60
50
50
50
40
40
40
30
30
30
20
0
500
1000
1500
20
0
500
1000
1500
20
VOUT
AC Coupled
20mV/div.
VOUT
AC Coupled
20mV/div.
VOUT
AC Coupled
20mV/div.
VSW
10V/div.
VSW
10V/div.
VSW
10V/div.
IL
1A/div.
IL
1A/div.
IL
1A/div.
4.0
0
500
1000
1500
Oscillating Frequency
vs RFREQ
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0.0
10
100
1000
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MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CURVES (continued)
VIN = 12V, VOUT = 5V, fS = 500kHz, TA = +25C, unless otherwise noted.
Startup Through EN
Shutdown Through EN
IOUT = 0.1A
IOUT = 0.1A
Startup Through EN
IOUT = 1A
VEN
5V/div.
VEN
5V/div.
VEN
5V/div.
VOUT
2V/div.
VOUT
2V/div.
VOUT
2V/div.
VSW
10V/div.
VSW
10V/div.
VSW
10V/div.
IL
1A/div.
IL
1A/div.
IL
1A/div.
Shutdown Through EN
Startup Through EN
Shutdown Through EN
IOUT = 1A
IOUT = 1.5A
IOUT = 1.5A
VEN
5V/div.
VEN
5V/div.
VEN
5V/div.
VOUT
2V/div.
VOUT
2V/div.
VOUT
2V/div.
VSW
10V/div.
IL
1A/div.
VSW
10V/div.
VSW
10V/div.
IL
2A/div.
IL
2A/div.
Short Circuit Entry
Shrot Circuit Recovery
Transient Response
IOUT = 0.1A
IOUT = 0.1A
IOUT = 0.5A to 1.5A
VOUT
2V/div.
VOUT
2V/div.
IL
1A/div.
IL
1A/div.
VOUT AC
100mV/div.
IL
1A/div.
IOUT
1A/div.
MPQ4459 Rev. 1.01
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MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
BLOCK DIAGRAM
VIN
VIN
EN
REFERENCE UVLO/
THERMAL
SHUTDOWN
5V +
-2.6V
INTERNAL
REGULATORS
+
-BST
SW
VOUT
1.5ms SS
SS
--
ISW
+
ISW
Level
Shift
FB
SW
Gm Error Amp
SS
0V8
--
COMP
+
OSCILLATOR
CLK
VOUT
COMP
GND
FREQ
Figure 1—Functional Block Diagram
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MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
OPERATION
The MPQ4459 is a variable frequency,
non-synchronous, step-down switching regulator
with an integrated high-side high voltage power
MOSFET. It provides a single highly efficient
solution with current mode control for fast loop
response and easy compensation. It features a
wide input voltage range, internal soft-start
control and precision current limiting. Its very low
operational quiescent current makes it suitable
for battery powered applications.
PWM Control
At moderate to high output current, the MPQ4459
operates in a fixed frequency, peak current
control mode to regulate the output voltage. A
PWM cycle is initiated by the internal clock. The
power MOSFET is turned on and remains on
until its current reaches the value set by the
COMP voltage. When the power switch is off, it
remains off for at least 100ns before the next
cycle starts. If, in one PWM period, the current in
the power MOSFET does not reach the COMP
set current value, the power MOSFET remains
on, saving a turn-off operation. Error Amplifier
The error amplifier compares the FB pin voltage
with the internal reference (REF) and outputs a
current proportional to the difference between the
two. This output current is then used to charge
the external compensation network to form the
COMP voltage, which is used to control the
power MOSFET current.
During operation, the minimum COMP voltage is
clamped to 0.9V and its maximum is clamped to
2.0V. COMP is internally pulled down to GND in
shutdown mode. COMP should not be pulled up
beyond 2.6V.
Internal Regulator
Most of the internal circuitries are powered from
the 2.6V internal regulator. This regulator takes
the VIN input and operates in the full VIN range.
When VIN is greater than 3.0V, the output of the
regulator is in full regulation. When VIN is lower
than 3.0V, the output decreases.
Enable Control
The MPQ4459 has a dedicated enable control
pin (EN). With high enough input voltage, the
chip can be enabled and disabled by EN which
has positive logic. Its falling threshold is a
precision 1.2V, and its rising threshold is 1.5V
(300mV higher).
When floating, EN is pulled up to about 3.0V by
an internal 1µA current source so it is enabled.
To pull it down, 1µA current capability is needed.
When EN is pulled down below 1.2V, the chip is
put into the lowest shutdown current mode.
When EN is higher than zero but lower than its
rising threshold, the chip is still in shutdown
mode but the shutdown current increases slightly.
Under-Voltage Lockout (UVLO)
Under-voltage lockout (UVLO) is implemented to
protect the chip from operating at insufficient
supply voltage. The UVLO rising threshold is
about 3.0V while its falling threshold is a
consistent 2.6V.
Internal Soft-Start
The soft-start is implemented to prevent the
converter output voltage from overshooting
during startup. When the chip starts, the internal
circuitry generates a soft-start voltage (SS)
ramping up from 0V to 2.6V. When it is lower
than the internal reference (REF), SS overrides
REF so the error amplifier uses SS as the
reference. When SS is higher than REF, REF
regains control.
Thermal Shutdown
Thermal shutdown is implemented to prevent the
chip from operating at exceedingly high
temperatures. When the silicon die temperature
is higher than its upper threshold, it shuts down
the whole chip. When the temperature is lower
than its lower threshold, the chip is enabled again.
Floating Driver and Bootstrap Charging
The floating power MOSFET driver is powered by
an external bootstrap capacitor. This floating
driver has its own UVLO protection. This UVLO’s
rising threshold is 2.2V with a threshold of
150mV.
The bootstrap capacitor is charged and regulated
to about 5V by the dedicated internal bootstrap
regulator. When the voltage between the BST
and SW nodes is lower than its regulation, a
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MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
PMOS pass transistor connected from VIN to
BST is turned on. The charging current path is
from VIN, BST and then to SW. External circuit
should provide enough voltage headroom to
facilitate the charging.
As long as VIN is sufficiently higher than SW, the
bootstrap capacitor can be charged. When the
power MOSFET is ON, VIN is about equal to SW
so the bootstrap capacitor cannot be charged.
When the external diode is on, the difference
between VIN and SW is largest, thus making it
the best period to charge. When there is no
current in the inductor, SW equals the output
voltage VOUT so the difference between VIN and
VOUT can be used to charge the bootstrap
capacitor.
At higher duty cycle operation condition, the time
period available to the bootstrap charging is less
so the bootstrap capacitor may not be sufficiently
charged.
In case the internal circuit does not have
sufficient voltage and the bootstrap capacitor is
not charged, extra external circuitry can be used
to ensure the bootstrap voltage is in the normal
operational region. Refer to External Bootstrap
Diode in Application section.
The DC quiescent current of the floating driver is
about 20µA. Make sure the bleeding current
power MOSFET. The cycle-by-cycle maximum
current of the internal power MOSFET is
internally limited.
Startup and Shutdown
If both VIN and EN are higher than their
appropriate thresholds, the chip starts. The
reference block starts first, generating stable
reference voltage and currents, and then the
internal regulator is enabled. The regulator
provides stable supply for the remaining
circuitries.
While the internal supply rail is up, an internal
timer holds the power MOSFET OFF for about
50µs to blank the startup glitches. When the
internal soft-start block is enabled, it first holds its
SS output low to ensure the remaining circuitries
are ready and then slowly ramps up.
Three events can shut down the chip: EN low,
VIN low and thermal shutdown. In the shutdown
procedure, power MOSFET is turned off first to
avoid any fault triggering. The COMP voltage and
the internal supply rail are then pulled down.
Programmable Oscillator
The MPQ4459 oscillating frequency is set by an
external resistor, RFREQ from the FREQ pin to
ground. The relationship between RFREQ and fS
refer to table1 in Application section.
at the SW node is higher than this value, such
that:
IO
VO
20A
(R1 R2)
Current Comparator and Current Limit
The power MOSFET current is accurately sensed
via a current sense MOSFET. It is then fed to the
high speed current comparator for the current
mode control purpose. The current comparator
takes this sensed current as one of its inputs.
When the power MOSFET is turned on, the
comparator is first blanked till the end of the turnon transition to avoid noise issues. The
comparator then compares the power switch
current with the COMP voltage. When the
sensed current is higher than the COMP voltage,
the comparator output is low, turning off the
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MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
APPLICATION INFORMATION
Setting the Frequency
The MPQ4459 has an externally adjustable
frequency. The switching frequency (fS) can be
set using a resistor at FREQ pin (RFREQ). The
recommended RFREQ value for various fS see
table1.
A few µA of current from the high-side BS
circuitry can be seen at the output when the
MPQ4459 is at no load. In order to absorb this
small amount of current, keep R2 under 40kΩ.
A typical value for R2 can be 40.2kΩ. With this
value, R1 can be determined by:
Table 1—fS vs. RFREQ
R1 50.25 ( VOUT 0.8)(k)
RFREQ (kΩ)
fS (MHz)
18
4
20
3.8
22.1
3.5
24
3.3
26.7
3
30
2.8
33.2
2.5
39
2.2
45.3
2
51
1.8
57.6
1.6
68
1.4
80.6
1.2
100
1
133
0.8
200
0.5
340
0.3
536
0.2
Setting the Output Voltage
The output voltage is set using a resistive
voltage divider from the output voltage to FB pin.
The voltage divider divides the output voltage
down to the feedback voltage by the ratio:
VFB VOUT
R2
R1 R2
Where VFB is the feedback voltage and VOUT is
the output voltage.
Thus the output voltage is:
VOUT VFB
For example, for a 3.3V output voltage, R2 is
40.2kΩ, and R1 is 127kΩ.
Inductor
The inductor is required to supply constant
current to the output load while being driven by
the switched input voltage. A larger value
inductor will result in less ripple current that will
result in lower output ripple voltage. However,
the larger value inductor will have a larger
physical size, higher series resistance, and/or
lower saturation current. A good rule for
determining the inductance to use is to allow
the peak-to-peak ripple current in the inductor
to be approximately 30% of the maximum
switch current limit. Also, make sure that the
peak inductor current is below the maximum
switch current limit. The inductance value can
be calculated by:
L1
VOUT
V
1 OUT
fS ΔIL
VIN
Where VIN is the input voltage, fS is the switching
frequency, and ΔIL is the peak-to-peak inductor
ripple current. Choose an inductor that will not
saturate under the maximum inductor peak
current. The peak inductor current can be
calculated by:
ILP ILOAD
VOUT
V
1 OUT
2 fS L1
VIN
Where ILOAD is the load current. Table 2 lists a
number of suitable inductors from various
manufacturers. The choice of which style
inductor to use mainly depends on the price vs.
size requirements and any EMI requirement.
(R1 R2)
R2
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MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
Table 2—Selected Inductors
Manufacturer
Part Number
Inductance
(µH)
Max DCR
(Ω)
Current Rating
(A)
Dimensions
L x W x H (mm3)
Wurth Electronics
Wurth Electronics
Wurth Electronics
Wurth Electronics
Wurth Electronics
Wurth Electronics
7447789002
7447789003
7447789004
744066100
744771115
744771122
2.2µH
3.3µH
4.7µH
10µH
15µH
22µH
0.019
0.024
0.033
0.035
0.025
0.031
4A
3.42A
2.9A
3.6A
3.75
3.37
7.3x7.3x3.2
7.3x7.3x3.2
7.3x7.3x3.2
10x10x3.8
12x12x6
12x12x6
TDK
RLF7030T-2R2
2.2µH
0.012
5.4A
7.3x6.8x3.2
TDK
TDK
TDK
TDK
RLF7030T-3R3
RLF7030T-4R7
SLF10145T-100
SLF12565T-150M4R2
3.3µH
4.7µH
10µH
15µH
0.02
0.031
0.0364
0.0237
4.1A
3.4A
3A
4.2
7.3x6.8x3.2
7.3x6.8x3.2
10.1x10.1x4.5
12.5x12.5x6.5
TDK
SLF12565T-220M3R5
22µH
0.0316
3.5
12.5x12.5x6.5
TOKO
TOKO
TOKO
TOKO
TOKO
TOKO
FDV0630-2R2M
FDV0630-3R3M
FDV0630-4R7M
#919AS-100M
#919AS-160M
#919AS-220M
2.2µH
3.3µH
4.7µH
10µH
16µH
22µH
0.021
0.031
0.049
0.0265
0.0492
0.0776
5.3
4.3
3.3
4.3
3.3
3.0
7.7x7x3
7.7x7x3
7.7x7x3
10.3x10.3x4.5
10.3x10.3x4.5
10.3x10.3x4.5
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MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
Output Rectifier Diode
The output rectifier diode supplies the current to
the inductor when the high-side switch is off. To
reduce losses due to the diode forward voltage
and recovery times, use a Schottky diode.
Choose a diode who’s maximum reverse
voltage rating is greater than the maximum
input voltage, and who’s current rating is
greater than the maximum load current. Table 3
lists
example
Schottky
diodes
and
manufacturers.
Table 3—Output Diodes
Manufacturer
Part Number
For simplification, choose the input capacitor
whose RMS current rating greater than half of
the maximum load current. The input capacitor
can be electrolytic, tantalum or ceramic. When
using electrolytic or tantalum capacitors, a small,
high quality ceramic capacitor, i.e. 0.1μF,
should be placed as close to the IC as possible.
When using ceramic capacitors, make sure that
they have enough capacitance to provide
sufficient charge to prevent excessive voltage
ripple at input. The input voltage ripple caused
by capacitance can be estimated by:
Voltage Current
Rating Rating Package
(V)
(A)
VIN
ILOAD
V
V
OUT 1 OUT
fS C1 VIN
VIN
Diodes Inc.
B240A-13-F
40V
2A
SMA
Where CIN is the input capacitance value.
Diodes Inc.
B340A-13-F
40V
3A
SMA
Central semi
CMSH2-40M
40V
2A
SMA
Output Capacitor
Central semi
CMSH3-40MA
40V
3A
SMA
Input Capacitor
The input current to the step-down converter is
discontinuous, therefore a capacitor is required
to supply the AC current to the step-down
converter while maintaining the DC input
voltage. Use low ESR capacitors for the best
performance. Ceramic capacitors are preferred,
but tantalum or low-ESR electrolytic capacitors
may also suffice. Since the input capacitor
absorbs the input switching current it requires
an adequate ripple current rating. The RMS
current in the input capacitor can be estimated
by:
I C1 ILOAD
VOUT VOUT
1
VIN
VIN
The worse case condition occurs at VIN = 2VOUT,
where:
IC1
I
LOAD
2
The output capacitor is required to maintain the
DC output voltage. Ceramic, tantalum, or low
ESR electrolytic capacitors are recommended.
Low ESR capacitors are preferred to keep the
output voltage ripple low. The output voltage
ripple can be estimated by:
VOUT
VOUT
V
1 OUT
f S L1
VIN
1
R ESR
8 f S C2
Where L is the inductor value, CO is the output
capacitance value, and RESR is the equivalent
series resistance (ESR) value of the output
capacitor.
In the case of ceramic capacitors, the
impedance at the switching frequency is
dominated by the capacitance. The output
voltage ripple is mainly caused by the
capacitance. For simplification, the output
voltage ripple can be estimated by:
ΔVOUT
V
1 OUT
VIN
L1 C2
VOUT
8 fS
2
MPQ4459 Rev. 1.01
www.MonolithicPower.com
1/24/2014
MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited.
© 2014 MPS. All Rights Reserved.
12
MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
In the case of tantalum or electrolytic capacitors,
the ESR dominates the impedance at the
switching frequency. For simplification, the
output ripple can be approximated to:
ΔVOUT
VOUT
V
1 OUT RESR
fS L1
VIN
The characteristics of the output capacitor also
affect the stability of the regulation system. The
MP1593 can be optimized for a wide range of
capacitance and ESR values.
Compensation Components
MPQ4459 employs current mode control for
easy compensation and fast transient response.
The system stability and transient response are
controlled through the COMP pin. COMP pin is
the output of the internal error amplifier. A
series capacitor-resistor combination sets a
pole-zero
combination
to
control
the
characteristics of the control system. The DC
gain of the voltage feedback loop is given by:
A VDC R LOAD G CS A VEA
VFB
VOUT
Where AVEA is the error amplifier voltage gain,
GCS is the current sense transconductance, and
RLOAD is the load resistor value. The system has
two poles of importance. One is due to the
compensation capacitor (C3), the output
resistor of error amplifier. The other is due to
the output capacitor and the load resistor.
These poles are located at:
fP1
GEA
2 C3 A VEA
fP2
1
2 C2 R LOAD
and
The system has one zero of importance, due to
the compensation capacitor (C3) and the
compensation resistor (R3). This zero is located
at:
f Z1
1
2 C3 R3
The system may have another zero of
importance, if the output capacitor has a large
capacitance and/or a high ESR value. The zero,
due to the ESR and capacitance of the output
capacitor, is located at:
fESR
1
2 C2 R ESR
In this case (as shown in Figure 2), a third pole
set by the compensation capacitor (C6) and the
compensation resistor (R3) is used to
compensate the effect of the ESR zero on the
loop gain. This pole is located at:
f P3
1
2 C6 R3
The goal of compensation design is to shape
the converter transfer function to get a desired
loop gain. The system crossover frequency
where the feedback loop has the unity gain is
important. Lower crossover frequencies result
in slower line and load transient responses,
while higher crossover frequencies could cause
system unstable. A good rule of thumb is to set
the crossover frequency to approximately onetenth of the switching frequency or lower. The
Table 4 lists the typical values of compensation
components for some standard output voltages
with various output capacitors and inductors.
The values of the compensation components
have been optimized for fast transient
responses and good stability at given conditions.
MPQ4459 Rev. 1.01
www.MonolithicPower.com
1/24/2014
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© 2014 MPS. All Rights Reserved.
13
MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
Table 4—Compensation Values for Typical
Output Voltage/Capacitor Combinations
VOUT
L
CO
R3
C3
C6
1.8V
4.7µH
47µF
ceramic
105k
100pF
None
2.5V
4.7µH6.8µH
22µF
ceramic
54.9k
220pF
None
6.8µH10µH
15µH22µH
22µH33µH
22µF
ceramic
22µF
ceramic
22µF
ceramic
68.1k
220pF
None
100k
150pF
None
147k
150pF
None
3.3V
5V
12V
Note: The selection of L is based on fs = 500KHz. Please
refer to “Inductor section” on page7 to select proper
inductor if fs is higher than that.
If this is the case, then add the second
compensation capacitor (C6) to set the pole fP3
at the location of the ESR zero. Determine the
C6 value by the equation:
C6
High Frequency Operation
The switching frequency of MPQ4459 can be
programmed up to 4MHz by an external resistor.
Please pay attention to the following if the
switching frequency is above 2MHz.
The minimum on time of MPQ4459 is about
80ns (typ). Pulse skipping operation can be
seen more easily at higher switching frequency
due to the minimum on time. Recommended
operating voltage at 4MHz is 12V or below, and
24V or below at 2MHz.
To optimize the compensation components for
conditions not listed in Table 3, the following
procedure can be used.
2 C2 f C VOUT
R3
G EA G CS
VFB
Where fC is the desired crossover frequency
(which typically has a value no higher than
1/10th of switching frequency).
2. Choose the compensation capacitor (C3) to
achieve the desired phase margin. For
applications with typical inductor values, setting
the compensation zero, fZ1, below one forth of
the crossover frequency provides sufficient
phase margin. Determine the C3 value by the
following equation:
C3
Input Max vs
Switching Frequency
30
MAX INPUT VOLTAGE (V)
1. Choose the compensation resistor (R3) to set
the desired crossover frequency. Determine the
R3 value by the following equation:
C2 R ESR
R3
25
20
VO=3.3V
15
10
5
1.5
VO=2.5V
2.0
2.5
3.0
fS (MHz)
3.5
4.0
Figure 2—Recommended Input vs. fS
4
2 R3 f C
Where R3 is the compensation resistor value.
3. Determine if the second compensation
capacitor (C6) is required. It is required if the
ESR zero of the output capacitor is located at
less than half of the switching frequency, or the
following relationship is valid:
f
1
S
2 C2 R ESR
2
MPQ4459 Rev. 1.01
www.MonolithicPower.com
1/24/2014
MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited.
© 2014 MPS. All Rights Reserved.
14
MPQ4459 – INDUSTRIAL GRADE, 1.5A, 4MHz, 36V STEP-DOWN CONVERTER
Since the internal bootstrap circuitry has higher
impedance, which may not be adequate to
charge the bootstrap capacitor during each
charging period, an external bootstrap charging
diode is strongly recommended if the switching
frequency is above 2MHz (see External
Bootstrap
Diode
section
for
detailed
implementation information).
With higher switching frequencies, the inductive
reactance (XL) of a capacitor dominates, such
that the ESL of the input/output capacitor
determines the input/output ripple voltage at
higher switching frequencies. As a result, high
frequency ceramic capacitors are strongly
recommended as input decoupling capacitors
and output filtering capacitors.
Layout becomes more important when the
device switches at higher frequency. It is
essential to place the input decoupling
capacitor, catch diode and the MPQ4459 as
close together as possible, with traces that are
very short and fairly wide. This can help to
greatly reduce the voltage spikes on SW and
also lower the EMI noise level.
Try to run the feedback trace as far from the
inductor and noisy power traces as possible. It
is a good idea to run the feedback trace on the
side of the PCB opposite of the inductor with a
ground plane separating the two. The
compensation components should be placed
close to the MPQ4459. Do not place the
compensation components close to or under
the high dv/dt SW node, or inside the high di/dt
power loop. If you have to do so, the proper
ground plane must be in place to isolate these
nodes. Switching losses are expected to
increase at high switching frequencies. To help
improve the thermal conduction, a grid of
thermal vias can be created right under the
exposed pad. It is recommended that they be
small (15mil barrel diameter) so that the hole is
essentially filled up during the plating process,
thus aiding conduction to the other side. Too
large a hole can cause solder wicking problems
during the reflow soldering process. The pitch
(distance between the centers) of several such
thermal vias in an area is typically 40mil.
External Bootstrap Diode
It is recommended that an external bootstrap
diode be added when the input voltage is no
greater than 5V or the 5V rail is available in the
system. This helps improve the efficiency of the
regulator. The bootstrap diode can be a low
cost one such as IN4148 or BAT54.
5V
BS
MPQ4459
0.1μ F
SW
Figure 3—External Bootstrap Diode
This diode is also recommended for high duty
cycle operation (when VOUT/VIN >65%) or low
VIN (
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