MPQ4560DQ-LF-Z

MPQ4560-AEC1 Industrial-Grade, 2A, 2MHz, 55V Step-Down Converter Available in AEC-Q100 DESCRIPTION The MPQ4560 is a high-frequency, step-down, switching regulator with an integrated, highside, high-voltage, power MOSFET. It provides a 2A output with current mode control for fast loop response and easy compensation. FEATURES     The wide 3.8V-to-55V input range accommodates a variety of step-down applications, including those in automotive input environment. A 12µA shutdown mode supply current allows use in battery-powered applications.       High-power conversion efficiency over a wide load range is achieved by scaling down the switching frequency in light load conditions to reduce the switching and gate driving losses.  Frequency foldback prevents inductor current runaway during startup and thermal shutdown provides reliable, fault tolerant operation. By switching at 2MHz, the MPQ4560 can prevent electromagnetic interference problems, such as those found in AM radio and ADSL applications. The MPQ4560 is available in small 3mm x 3mm QFN10 and SOIC8E packages. Guaranteed Industrial Automotive Temperature Range Limits Wide 3.8V-to-55V Operating Input Range 250mΩ Internal Power MOSFET Up to 2MHz Programmable Switching Frequency 140μA Quiescent Current Ceramic Capacitor Stable Internal Soft-Start Up to 95% Efficiency Output Adjustable from 0.8V to 52V Available in QFN10 (3mmx3mm) and SOIC8E Packages AEC-Q100 Qualified APPLICATIONS      High-Voltage Power Conversion Automotive Systems Industrial Power Systems Distributed Power Systems Battery Powered Systems All MPS parts are lead-free and adhere to the RoHS directive. For MPS green status, please visit MPS website under Products, Quality Assurance page. “MPS” and “The Future of Analog IC Technology” are Registered Trademarks of Monolithic Power Systems, Inc. TYPICAL APPLICATION MPQ4560 Rev. 1.1 www.MonolithicPower.com 5/24/2016 MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 1 MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER ORDERING INFORMATION Part Number Package Top Marking MPQ4560DN* SOIC8E MP4560DN MPQ4560DQ** QFN10 (3×3mm) T8 MPQ4560DN-AEC1 SOIC8E MP4560DN MPQ4560DQ-AEC1 QFN10 (3×3mm) T8 Junction Temperature (TJ) –40°C to +125°C * For Tape & Reel, add suffix –Z (e.g. MPQ4560DN-Z) For RoHS Compliant Packaging, add suffix –LF, (e.g. MPQ4560DN-LF–Z) ** For Tape & Reel, add suffix –Z (e.g. MPQ4560DQ-Z) For RoHS Compliant Packaging, add suffix –LF, (e.g. MPQ4560DQ-LF–Z) PACKAGE REFERENCE QFN10 (3x3mm) SOIC8E ABSOLUTE MAXIMUM RATINGS (1) Thermal Resistance Supply Voltage (VIN).................... –0.3V to +60V Switch Voltage (VSW)......... –0.5V to (VIN + 0.5V) BST to SW .................................... –0.3V to +5V All Other Pins ................................ –0.3V to +5V Continuous Power Dissipation .......(TJ = 25°C)(2) QFN10 (3×3mm) ........................................2.5W SOIC8E .....................................................2.5W Junction Temperature .............................. 150°C Lead Temperature ................................... 260°C Storage Temperature .............. –65°C to +150°C QFN10 (3x3mm) ..................... 50 ...... 12 ... °C/W SOIC8E .................................. 50 ...... 10 ... °C/W Recommended Operating Conditions (3) Supply Voltage VIN .......................... 3.8V to 55V Output Voltage VOUT........................ 0.8V to 52V Maximum Junction Temp. (TJ) .............. +125°C (4) θJA θJC Notes: 1) Exceeding these ratings may damage the device 2) The maximum allowable power dissipation is a function of the maximum junction temperature TJ(MAX), the junction-toambient thermal resistance θJA, and the ambient temperature TA. The maximum allowable continuous power dissipation at any ambient temperature is calculated by PD(MAX)=(TJ(MAX)TA)/ θJA. Exceeding the maximum allowable power dissipation will cause excessive die temperature, and the regulator will go into thermal shutdown. Internal thermal shutdown circuitry protects the device from permanent damage. 3) The device is not guaranteed to function outside of its operating conditions. 4) Measured on JESD51-7 4-layer board. MPQ4560 Rev. 1.1 www.MonolithicPower.com 5/24/2016 MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 2 MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER ELECTRICAL CHARACTERISTICS VIN = 12V, VEN = 2.5V, VCOMP = 1.4V, TJ= –40°C to +125°C, unless otherwise noted. Typical Values are at TJ=25°C. Parameter Symbol Feedback Voltage VFB Feedback Leakage Current Upper Switch On Resistance 4.5V < VIN < 55V Min Typ Max TJ=25°C 0.780 0.797 0.820 −40°C ≤ TJ ≤85°C 0.772 0.829 −40°C ≤ TJ ≤125°C 0.766 0.829 IFB (5) RDS(ON) Upper Switch Leakage ISW Current Limit ILIM COMP to Current Sense (5) Transconductance GCS Error Amp Voltage Gain Condition VBST – VSW = TJ=25°C 5V 175 Duty Cycle ≤ 60% 1.0 250 330 160 VEN = 0V, VSW = 0V TJ=25°C 0.1 400 3.2 2.2 (6) V μA mΩ μA 1 2.6 Units 4.5 4.7 A 5.7 A/V 400 V/V Error Amp Transconductance ICOMP = ±3µA 120 µA/V Error Amp Min Source current VFB = 0.7V 10 µA Error Amp Min Sink current VFB = 0.9V −10 µA TJ=25°C VIN UVLO Threshold 2.7 2.4 VIN UVLO Hysteresis Soft-Start Time (5) 0V < VFB < 0.8V Oscillator Frequency 3.0 fSW RFREQ = 95kΩ TJ=25°C 3.3 V 3.6 0.35 V 0.19 0.5 ms 0.8 1 0.7 1.2 MHz 1.3 Shutdown Supply Current IS VEN < 0.3V 12 20 µA Quiescent Supply Current IQ No load, VFB = 0.9V (no switching) 140 200 µA Hysteresis = 20°C 150 °C Thermal Shutdown (5) Minimum Off Time (5) tOFF 100 ns Minimum On Time (5) tON 100 ns EN Rising Threshold TJ=25°C EN Threshold Hysteresis 1.4 1.55 1.3 1.7 V 1.8 320 mV Note: 5) Derived from bench characterization. Not tested in production. 6) Guaranteed by design. Not tested in production. MPQ4560 Rev. 1.1 www.MonolithicPower.com 5/24/2016 MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 3 MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER PIN FUNCTIONS QFN Pin # SOIC8 Pin # 1, 2 1 3 2 4 3 5 4 6 5 7 6 8, 9 7 10 8 Name Description Switch Node. Output from the high-side switch. A low VF Schottky rectifier to ground is required. The rectifier must be close to the SW pins to reduce switching spikes. Enable Input. Pull this pin below the specified threshold to shutdown the chip. Pull it EN up above the specified threshold or leaving it floating to enable the chip. Compensation. Output of the GM error amplifier. Control loop frequency COMP compensation is applied to this pin. Feedback. Input to the error amplifier. Sets the regulator voltage by comparing the FB tap of an external resistive divider connected between the output and GND to the internal +0.8V reference. GND, Ground. Connect as close as possible to the output capacitor and avoid the highExposed current switch paths. Connect exposed pad to GND plane for optimal thermal pad performance. Switching Frequency Program Input. Connect a resistor from this pin to ground to set FREQ the switching frequency. Input Supply. This supplies power to all the internal control circuitry, both BS VIN regulators, and the high-side switch. Place a decoupling capacitor to ground close to this pin to minimize switching spikes. SW BST Bootstrap. Positive power supply for the internal floating high-side MOSFET driver. Connect a bypass capacitor between this pin and SW pin. MPQ4560 Rev. 1.1 www.MonolithicPower.com 5/24/2016 MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 4 MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER TYPICAL CHARACTERISTICS MPQ4560 Rev. 1.1 www.MonolithicPower.com 5/24/2016 MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 5 MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER TYPICAL CHARACTERISTICS (continued) MPQ4560 Rev. 1.1 www.MonolithicPower.com 5/24/2016 MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 6 MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER TYPICAL PERFORMANCE CHARACTERISTICS VIN = 12V, VOUT =3.3V, C1 = 4.7µF, C2 = 22µF, L1 = 10µH and TA = 25°C, unless otherwise noted. MPQ4560 Rev. 1.1 www.MonolithicPower.com 5/24/2016 MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 7 MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER TYPICAL PERFORMANCE CHARACTERISTICS (continued) VIN = 12V, VOUT =3.3V, C1 = 4.7µF, C2 = 22µF, L1 = 10µH and TA = 25°C, unless otherwise noted. MPQ4560 Rev. 1.1 www.MonolithicPower.com 5/24/2016 MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 8 MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER BLOCK DIAGRAM Figure 1: Functional Block Diagram MPQ4560 Rev. 1.1 www.MonolithicPower.com 5/24/2016 MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 9 MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER OPERATION The MPQ4560 is an asynchronous, step-down, switching regulator with an integrated high-side, high-voltage, power MOSFET and a programmable frequency. It provides a single highly-efficient solution with current-mode control for fast loop response and easy compensation. It features a wide input voltage range, internal softstart control, and precise current limiting. Its very low operational quiescent current makes it suitable for battery-powered applications. PWM Control The MPQ4560 operates in a fixed-frequency, peak-current-control mode to regulate the output voltage at moderate-to-high output current. The internal clock initiates a PWM cycle. The power MOSFET turns ON and remains ON until its current reaches the value set by the COMP voltage. When the power switch is OFF, it remains OFF for at least 100ns before the next cycle starts. If the current in the power MOSFET does not reach the COMP-set current value within one PWM period, the power MOSFET remains ON, saving a turn-off operation. Enable Control The MPQ4560 has a dedicated enable control pin (EN) that can enable or disable the chip when the input voltage exceeds an upper threshold. Its falling threshold (turn-off) is 1.2V, and its rising threshold (turn-on) is 1.5V (300mV higher). When floating, an internal 1µA current source pulls EN up to ~3.0V to enable the chip. Pulldown requires a 1µA current. When EN is pulled below 1.2V, the chip enters its lowest shutdown current mode. When EN exceeds 0V but remains lower than its rising threshold, the chip remains in shutdown mode but the shutdown current increases slightly. Under-Voltage Lockout Under-voltage lockout (UVLO) protects the chip from operating at insufficient supply voltage. The UVLO rising threshold is about 3.0V while its falling threshold is a consistent 2.6V. Pulse-Skipping Mode Under light-load condition the switching frequency stretches the zero-voltage period to reduce the switching loss and driving loss. Internal Soft-Start Soft-start prevents the converter output voltage from overshooting during startup and short-circuit recovery. When the chip starts, the internal circuit generates a soft-start voltage (SS) ramping up from 0V to 2.6V. When it is less than the VREF, SS overrides VREF so the error amplifier uses SS as the reference. When SS exceeds VREF, VREF regains control. Error Amplifier The error amplifier compares the FB pin voltage (VFB) to the internal reference (VREF) and outputs a current proportional to the difference. This output current charges the external compensation network to form VCOMP, which controls the power MOSFET current. Thermal Shutdown Thermal shutdown prevents the chip from operating at exceedingly high temperatures. When the silicon die temperature exceeds its upper threshold, the whole chip shuts down. When the temperature is less than its lower threshold, the chip is enabled again. During operation, the minimum VCOMP is clamped to 0.9V and its maximum is clamped to 2.0V. COMP is internally pulled down to GND in shutdown mode. Do not pull VCOMP above 2.6V. Floating Driver and Bootstrap Charging An external bootstrap capacitor powers the floating power MOSFET driver. This floating driver has its own UVLO protection. This UVLO’s rising threshold is 2.2V with a hysteresis of 150mV. The driver’s UVLO is soft-start related: When the bootstrap voltage hits its UVLO threshold, the soft-start circuit resets. To prevent noise, there is 20µs delay before the reset action. When bootstrap UVLO is gone, the reset is off and then the soft-start process resumes. Internal Regulator An internal 2.6V regulator powers most of the internal circuits. This regulator takes the VIN input and operates in the full VIN range. When VIN exceeds 3.0V, the output of the regulator is in full regulation. When VIN is less than 3.0V, the output decreases. The dedicated internal bootstrap regulator regulates and charges the bootstrap capacitor to MPQ4560 Rev. 1.1 www.MonolithicPower.com 5/24/2016 MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 10 MPQ4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER ~5V. When the voltage between the BST and SW nodes is less than its regulation, a PMOS pass transistor from VIN to BST turns ON. The charging current path is from VIN, BST and then to SW. An external circuit must provide enough voltage headroom to facilitate charging. During a short circuit, the VFB voltage is low and pulls down VSS to ~100mV above VFB. Removing the short circuit causes the output voltage to recover with VSS. When VFB is high enough, the frequency and current limit return to normal values. As long as VIN is sufficiently higher than VSW, the bootstrap capacitor can charge. When the power MOSFET is ON, VIN≈VSW so the bootstrap capacitor cannot charge. When the external diode is ON, the difference between VIN and VSW is at its largest, thus making it the best period to charge. When there is no current in the inductor, VSW=VOUT so the difference between VIN and VOUT can charge the bootstrap capacitor. Startup and Shutdown If both VIN and VEN exceed their respective thresholds, the chip starts. The reference block initiates to generate a stable reference voltage and currents, and then the internal regulator is enabled. The regulator provides a stable supply for the remaining circuitries. At higher duty cycles, the time period available for bootstrap charging is shorter so the bootstrap capacitor may not sufficiently charge. If the internal circuit does not have sufficient voltage and the bootstrap capacitor is not charged, extra external circuitry can ensure the bootstrap voltage is within the normal operational region. The DC quiescent current of the floating driver is about 20µA. Make sure the bleeding current at the SW node exceeds this value, such that: VO IO   20A (R1  R2) Current Comparator and Current Limit A current-sense MOSFET accurately senses the power MOSFET’s current. The result goes to the high-speed current comparator for current-mode control.: When the power MOSFET turns ON, the comparator is first blanked till the end of the turnon transition to avoid noise issues. The comparator then compares the power switch current to VCOMP. When the sensed current exceeds VCOMP, the comparator output is LOW, turning OFF the power MOSFET. The cycle-by-cycle maximum current of the internal power MOSFET is internally limited. While the internal supply rail is up, an internal timer holds the power MOSFET OFF for about 50µs to blank the startup noise. When the internal soft-start block is enabled, it first holds its SS output low to ensure the remaining circuitries are ready and then slowly ramps up. Three events can shut down the chip: VEN LOW, VIN LOW and thermal shutdown. During shutdown, the power MOSFET turns OFF first to avoid any fault triggering. Then VCOMP and the internal supply rail drop. Programmable Oscillator An external resistor (RFREQ) from the FREQ pin to ground sets the MPQ4560 oscillating frequency. The value of RFREQ can be calculated from: RFREQ (kΩ) = 100000 -5 fS (kHz) For example, for fSW=500kHz, RFREQ=195kΩ. Short Circuit Protection When the output is shorted to the ground, the switching frequency folds back and the current limit falls to lower the short-circuit current. When VFB is zero, the current limit drops to about 50% of its full current limit. When VFB exceeds 0.4V, current limit reaches 100%. MPQ4560 Rev. 1.1 5/24/2016 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 11 MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER APPLICATION INFORMATION COMPONENT SELECTION Setting the Output Voltage A resistive voltage divider from the output voltage to FB pin sets the output voltage. The voltage divider divides the output voltage down to the feedback voltage by the ratio: VFB =VOUT  R2 R1+R2 Thus the output voltage is: VOUT =VFB  R1+R2 R2 For example, the value for R2 can be 10kΩ. With this value, R1 is: R1=12.5  (VOUT -0.8)(KΩ) So for a 3.3V output voltage, R2 is 10kΩ, and R1 is 31.6kΩ. Inductor The inductor provides constant current to the output load while being driven by the switched input voltage. A larger-value inductor will result in lower ripple current that will lower the output ripple voltage. However, a larger inductor value will be physically larger, have higher series resistance, or lower saturation current. To determine the inductance, allow the inductor’s peak-to-peak ripple current to approximately equal 30% of the maximum switch current limit. Make sure that the peak inductor current is less than the maximum switch current limit. The inductance value can be calculated by: L1= VOUT fs  ΔIL  (1- VOUT VIN ) Where VOUT is the output voltage, VIN is the input voltage, fS is the switching frequency, and ∆IL is the peak-to-peak inductor ripple current. Choose an inductor that will not saturate under the maximum inductor peak current. The peak inductor current can be calculated by: ILP  ILOAD   VOUT V  1  OUT 2  fS  L1  VIN    Where ILOAD is the load current. Table 1 lists several suitable inductors from various manufacturers. The different inductor choices include price vs. size requirements and any EMI requirements. MPQ4560 Rev. 1.1 www.MonolithicPower.com 5/24/2016 MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 12 MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER Table 1: Inductor Selection Guide Inductance Max DCR Current Rating Dimensions (µH) (Ω) (A) L × W × H (mm3) 7447789004 4.7 0.033 2.9 7.3×7.3×3.2 744066100 10 0.035 3.6 10×10×3.8 744771115 15 0.025 3.75 12×12×6 744771122 22 0.031 3.37 12×12×6 RLF7030T-4R7 4.7 0.031 3.4 7.3×6.8×3.2 SLF10145T-100 10 0.0364 3 10.1×10.1×4.5 SLF12565T-150M4R2 15 0.0237 4.2 12.5×12.5×6.5 SLF12565T-220M3R5 22 0.0316 3.5 12.5×12.5×6.5 FDV0630-4R7M 4.7 0.049 3.3 7.7×7×3 919AS-100M 10 0.0265 4.3 10.3×10.3×4.5 919AS-160M 16 0.0492 3.3 10.3×10.3×4.5 919AS-220M 22 0.0776 3 10.3×10.3×4.5 Part Number Wurth Electronics TDK Toko Output Rectifier Diode The output rectifier diode supplies the current to the inductor when the high-side switch is OFF. Use a Schottky diode to reduce losses from the diode forward voltage and recovery times. Choose a diode whose maximum reverse voltage rating exceeds the maximum input voltage, and whose current rating exceeds the maximum load current. Table 2 lists example Schottky diodes and manufacturers. Table 2: Diode Selection Guide Diodes Voltage/ Current Rating Manufacturer B290-13-F 90V, 2A Diodes Inc. B380-13-F 80V, 3A Diodes Inc. CMSH2-100M 100V, 2A Central Semi CMSH3-100MA 100V, 3A Central Semi MPQ4560 Rev. 1.1 www.MonolithicPower.com 5/24/2016 MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 13 MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER Input Capacitor The input current to the step-down converter is discontinuous and requires a capacitor to supply the AC current to the step-down converter while maintaining the DC input voltage. Use capacitors with low equivalent series resistances (ESR) for the best performance. Ceramic capacitors are best, but tantalum or low-ESR electrolytic capacitors may also suffice. For simplification, choose the input capacitor with an RMS current rating greater than half of the maximum load current. The input capacitor (C1) can be electrolytic, tantalum, or ceramic. When using electrolytic or tantalum capacitors, place a small, high-quality, ceramic capacitor (0.1μF) as close to the IC as possible. When using ceramic capacitors, make sure that they have enough capacitance to provide sufficient charge to prevent excessive voltage ripple at the input. The input voltage ripple caused by capacitance is approximately: VIN   ILOAD V V  OUT  1  OUT fS  C1 VIN  VIN    Output Capacitor The output capacitor (C2) maintains the DC output voltage. Use ceramic, tantalum, or lowESR electrolytic capacitors. Low-ESR capacitors are preferred to keep the output voltage ripple low. The output voltage ripple can be estimated as: VOUT  VOUT  V  1  OUT fS  L  VIN    1     R ESR   8  f S  C2    Where L is the inductor value and RESR is the ESR value of the output capacitor. For ceramic capacitors, the capacitance dominates the impedance at the switching frequency and contributes the most to the output voltage ripple. For simplification, the output voltage ripple can be estimated by: ΔVOUT   V  1  OUT VIN  L  C2  VOUT 8  fS 2    For tantalum or electrolytic capacitors, the ESR dominates the impedance at the switching frequency. For simplification, the output ripple is approximately: ΔVOUT  VOUT  V  1  OUT fS  L  VIN    R ESR  The characteristics of the output capacitor also affect the stability of the regulation system. The MPQ4560 can be optimized for a wide range of capacitances and ESR values. Compensation Components MPQ4560 employs current-mode control for easy compensation and fast transient response. The COMP pin controls the system stability and transient response. The COMP pin is the output of the internal error amplifier. A series capacitorresistor combination sets a pole-zero combination to control the control system’s characteristics. The DC gain of the voltage feedback loop is: A VDC  R LOAD  GCS  A VEA  VFB VOUT Where  AVEA is the error-amplifier voltage gain, 400V/V;  GCS is the current-sense transconductance, 5.6A/V; and  RLOAD is the load resistor value. The system has two important poles: One from the compensation capacitor (C3) and the output resistor of error amplifier, and the other due to the output capacitor and the load resistor. These poles are located at: fP1  GEA 2 π C3  A VEA fP 2  1 2 π C2  RLOAD Where, GEA is the transconductance, 120μA/V. error-amplifier The system has one important zero due to the compensation capacitor and the compensation resistor (R3). This zero is located at: fZ1  1 2 π C3  R3 MPQ4560 Rev. 1.1 www.MonolithicPower.com 5/24/2016 MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited. © 2016 MPS. All Rights Reserved. 14 MPQ4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER The system may have another significant zero if the output capacitor has a large capacitance or a high ESR value. This zero is located at: fESR C3  1  2π  C2  RESR In this case, a third pole set by the compensation capacitor (C5) and the compensation resistor can compensate for the effect of the ESR zero. This pole is located at: fP 3  1 2 π C5  R3 The goal of compensation design is to shape the converter transfer function for a desired loop gain. The system crossover frequency where the feedback loop has unity gain is important: Lower crossover frequencies result in slower line and load transient responses, while higher crossover frequencies lead to system instability. Generally, set the crossover frequency to ~0.1×fSW. Table 3: Compensation Values for Typical Output Voltage/Capacitor Combinations VOUT (V) L (µH) C2 (µF) R3 (kΩ) C3 (pF) C6 (pF) 1.8 4.7 33 32.4 680 None 2.5 4.7 - 6.8 22 26.1 680 None 3.3 6.8 -10 22 68.1 220 None 5 15 - 22 33 47.5 330 None 12 10 22 16 470 2 To optimize the compensation components for conditions not listed in Table 3, follow these steps: 1. Choose R3 to set the desired crossover frequency: 2 π C2  fC VOUT R3   GEA  GCS VFB Where fC is the desired crossover frequency. 2. Choose C3 to achieve the desired phase margin. For applications with typical inductor MPQ4560 Rev. 1.1 5/24/2016 values, set the compensation zero (fZ1)