LM25085SDE

LM25085 42V Constant On-Time PFET Buck Switching Controller December 17, 2009 LM25085 42V Constant On-Time PFET Buck Switching Controller General Description The LM25085 is a high efficiency PFET switching regulator controller that can be used to quickly and easily develop a small, efficient buck regulator for a wide range of applications. This high voltage controller contains a PFET gate driver and a high voltage bias regulator which operates over a wide 4.5V to 42V input range. The constant on-time regulation principle requires no loop compensation, simplifies circuit implementation, and results in ultra-fast load transient response. The operating frequency remains nearly constant with line and load variations due to the inverse relationship between the input voltage and the on-time. The PFET architecture allows 100% duty cycle operation for a low dropout voltage. Either the RDS(ON) of the PFET or an external sense resistor can be used to sense current for over-current detection. Features ■ LM25085Q is an Automotive Grade product that is AEC■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ Q100 grade 1 qualified (-40°C to 125°C operating junction temperature) Wide 4.5V to 42V input voltage range Adjustable current limit using RDS(ON) or a current sense resistor Programmable switching frequency to 1MHz No loop compensation required Ultra-Fast transient response Nearly constant operating frequency with line and load variations Adjustable output voltage from 1.25V Precision ±2% feedback reference Capable of 100% duty cycle operation Internal soft-start timer Integrated high voltage bias regulator Thermal shutdown Package ■ MSOP-8EP ■ MSOP-8 ■ LLP-8 (3 mm x 3 mm) Typical Application, Basic Step Down Controller 30079301 © 2009 National Semiconductor Corporation 300793 www.national.com LM25085 Connection Diagrams 30079302 Top View 8-Lead MSOP-EP 30079304 Top View 8-Lead LLP 30079303 Top View 8-Lead MSOP Ordering Information Order Number LM25085QMYE LM25085QMY LM25085QMYX LM25085MYE LM25085MY LM25085MYX LM25085MME LM25085MM LM25085MMX LM25085SDE LM25085SD LM25085SDX Package Type MSOP-8EP MSOP-8EP MSOP-8EP MSOP-8EP MSOP-8EP MSOP-8EP MSOP-8 MSOP-8 MSOP-8 LLP-8 LLP-8 LLP-8 NSC Package Drawing MUY08A MUY08A MUY08A MUY08A MUY08A MUY08A MUA08A MUA08A MUA08A SDA08A SDA08A SDA08A -40°C to +125°C -40°C to +125°C Junction Temperature Range Supplied As 250 units on tape and reel 1000 units on tape and reel 3500 units on tape and reel 250 units on tape and reel 1000 units on tape and reel 3500 units on tape and reel 250 units on tape and reel 1000 units on tape and reel 3500 units on tape and reel 250 units on tape and reel 1000 units on tape and reel 4500 units on tape and reel Feature AEC-Q100 Grade 1 qualified. Automotive Grade Production Flow* *Automotive Grade (Q) product incorporates enhanced manufacturing and support processes for the automotive market, including defect detection methodologies. Reliability qualification is compliant with the requirements and temperature grades defined in the AEC-Q100 standard. Automotive grade products are identified with the letter Q. For more information go to http://www.national.com/automotive. www.national.com 2 LM25085 Pin Descriptions Pin No. 1 2 3 4 5 6 7 Name ADJ RT FB GND ISEN Description Current Limit Adjust On-time control and shutdown Voltage Feedback from the regulated output Circuit Ground Current sense input for current limit detection. Output of the gate driver bias regulator Input supply voltage Exposed Pad Application Information The current limit threshold is set by an external resistor from VIN to ADJ in conjunction with the external sense resistor or the PFET’s RDS(ON). An external resistor from VIN to RT sets the buck switch on-time and switching frequency. Grounding this pin shuts down the controller. Input to the regulation and over-voltage comparators. The regulation level is 1.25V. Ground reference for all internal circuitry Connect to the PFET drain when using RDS(ON) current sense. Connect to the PFET source and the sense resistor when using a current sense resistor. Connect to the gate of the external PFET. Output of the negative voltage regulator (relative to VIN) that biases the PFET gate driver. A low ESR capacitor is required from VIN to VCC, located as close as possible to the pins. The operating input range is from 4.5V to 42V. A low ESR bypass capacitor must be located as close as possible to the VIN and GND pins. Exposed pad on the underside of the package (MSOP-8EP and LLP only). This pad is to be soldered to the PC board ground plane to aid in heat dissipation. PGATE Gate Driver Output VCC 8 VIN EP 3 www.national.com LM25085 Absolute Maximum Ratings (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN to GND ISEN to GND ADJ to GND RT, FB to GND -0.3V to 45V -0.3V to VIN + 0.3V -0.3V to VIN + 0.3V -0.3V to 7V VIN to VCC, VIN to PGATE ESD Rating (Note 2) Human Body Model Storage Temperature Range -0.3V to 10V 2kV -65°C to +150°C (Note 1) 4.5V to 42V −40°C to + 125°C Operating Ratings VIN Voltage Junction Temperature Electrical Characteristics Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are guaranteed through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated the following conditions apply: VIN = 24V, RT = 100 kΩ. Symbol VIN Pin IIN IQ VCC(reg) Operating current Shutdown current VIN - VCC Non-switching, FB = 1.4V (Note 6) RT = 0V (Note 6) Vin = 9V, FB = 1.4V, ICC = 0 mA Vin = 9V, FB = 1.4V, ICC = 20 mA Vin = 42V, FB = 1.4V, ICC = 0 mA UVLOVcc VCC under-voltage lock-out threshold UVLOVcc hysteresis VCC(CL) PGATE Pin VPGATE(HI) VPGATE(LO) VPGATE(HI)4.5 VPGATE(LO)4.5 IPGATE RPGATE PGATE High voltage PGATE Low voltage PGATE Pin = Open PGATE Pin = Open VIN -0.1 VIN -0.1 VIN VCC VIN VCC 1.75 1.5 2.3 2.3 32 -9 40 0 0.73 50 VIN = 4.5V, RT = 100 kΩ VIN = 24V, RT = 100 kΩ VIN = 42V, RT = 100 kΩ Minimum on-time in current limit (Note 7) VIN = 24V, 25 mV overdrive at ISEN 3.5 560 329 55 5 720 415 140 7.15 870 500 235 48 9 VCC+0.1 VCC+0.1 V V V V A A Ω Ω µA mV V mV µs ns ns ns VCC Current Limit VCC increasing VCC decreasing FB = 1.4V 20 6.9 1.25 175 7.7 7.7 7.7 3.8 260 40 1.75 300 8.5 mA µA V V V V mV mA Parameter Conditions Min Typ Max Units VCC Regulator (Note 3) PGATE High Voltage at Vin = 4.5V PGATE Pin = Open PGATE Low Voltage at Vin = 4.5V PGATE Pin = Open Driver Output Source Current Driver Output Sink Current Driver Output Resistance VIN = 12V, PGATE = VIN - 3.5V VIN = 12V, PGATE = VIN - 3.5V Source current = 500 mA Sink current = 500 mA Current Limit Detection IADJ VCL OFFSET RT Pin RTSD RTHYS On-Time tON – 1 tON – 2 tON - 3 tON - 4 On-time Shutdown threshold Shutdown threshold hysteresis RT Pin voltage rising ADJUST pin current source Current limit comparator offset VADJ = 22.5V VADJ = 22.5V, VADJ - VISEN www.national.com 4 LM25085 Symbol Off-Time tOFF(CL1) tOFF(CL2) tOFF(CL3) tOFF(CL4) Parameter Off-time (current limit) (Note 7) Conditions VIN = 12V, VFB = 0V VIN = 12V, VFB = 1V VIN = 24V, VFB = 0V VIN = 24V, VFB = 1V Min 5.35 1.42 8.9 2.22 1.225 Typ 7.9 1.9 13 3.2 1.25 350 10 Max 10.84 3.03 17.7 4.68 1.275 Units µs µs µs µs V mV nA Regulation and Over-Voltage Comparators (FB Pin) VREF VOV IFB tSS TSD THYS θJA FB regulation threshold FB over-voltage threshold FB bias current Soft-start time Junction shutdown temperature Junction shutdown hysteresis Junction to ambient, 0 LFPM air flow (Note 5) Junction to case, 0 LFPM air flow (Note 5) MSOP-8 package MSOP-8EP package LLP-8 package θJC MSOP-8 package MSOP-8EP package LLP-8 package Junction temperature rising 1.4 Measured with respect to VREF Soft-Start Function 2.5 170 20 126 46 54 29 5.5 9.1 °C/W 4.3 ms °C °C °C/W Thermal Shutdown Thermal Resistance Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics. Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Note 3: VCC provides self bias for the internal gate drive. Note 4: For detailed information on soldering plastic MSOP and LLP packages refer to the Packaging Data Book available from National Semiconductor Corporation. Note 5: Tested on a 4 layer JEDEC board. Four vias provided under the exposed pad. See JEDEC standards JESD51-5 and JESD51-7. Note 6: Operating current and shutdown current do not include the current in the RT resistor. Note 7: The tolerance of the minimum on-time (tON-4) and the current limit off-times (tOFF(CL1) through (tOFF(CL4)) track each other over process and temperature variations. A device which has an on-time at the high end of the range will have an off-time that is at the high end of its range. 5 www.national.com LM25085 Typical Performance Characteristics Unless otherwise specified the following conditions apply: TJ = 25°C, VIN = 24V. Efficiency (Circuit of Figure 4) Input Operating Current vs. VIN 30079305 30079306 Shutdown Current vs. VIN VCC vs. VIN 30079307 30079308 VCC vs. ICC On-Time vs. RT and VIN 30079309 30079311 www.national.com 6 LM25085 Off-Time vs. VIN and VFB Voltage at the RT Pin 30079312 30079313 ADJ Pin Current vs. VIN Input Operating Current vs. Temperature 30079317 30079318 Shutdown Current vs. Temperature VCC vs. Temperature 30079352 30079353 7 www.national.com LM25085 On-Time vs. Temperature Minimum On-Time vs. Temperature 30079354 30079355 Off-Time vs. Temperature Current Limit Comparator Offset vs. Temperature 30079356 30079357 ADJ Pin Current vs. Temperature PGATE Driver Output Resistance vs. Temperature 30079314 30079310 www.national.com 8 LM25085 Feedback Reference Voltage vs. Temperature Soft-Start Time vs. Temperature 30079315 30079316 RT Pin Shutdown Threshold vs. Temperature 30079358 9 www.national.com LM25085 Block Diagram 30079319 Sense resistor method shown for current limit detection. Minimum output ripple configuration shown. www.national.com 10 LM25085 Functional Description OVERVIEW The LM25085 is a PFET buck (step-down) DC-DC controller using the constant on-time (COT) control principle. The input operating voltage range of the LM25085 is 4.5V to 42V. The use of a PFET in a buck regulator greatly simplifies the gate drive requirements and allows for 100% duty cycle operation to extend the regulation range when operating at low input voltage. However, PFET transistors typically have higher onresistance and gate charge when compared to similarly rated NFET transistors. Consideration of available PFETs, input voltage range, gate drive capability of the LM25085, and thermal resistances indicate an upper limit of 10A for the load current for LM25085 applications. Constant on-time control is implemented using an on-time one-shot that is triggered by the feedback signal. During the off-time, when the PFET (Q1) is off, the load current is supplied by the inductor and the output capacitor. As the output voltage falls, the voltage at the feedback comparator input (FB) falls below the regulation threshold. When this occurs Q1 is turned on for the one-shot period which is determined by the input voltage (VIN) and the RT resistor. During the on-time the increasing inductor current increases the voltage at FB above the feedback comparator threshold. For a buck regulator the basic relationship between the on-time, off-time, input voltage and output voltage is: sistors. The output voltage ripple is a result of the inductor’s ripple current passing through the output capacitor’s ESR, or through a resistor in series with the output capacitor. Multiple methods are available to ensure sufficient ripple is supplied to the FB pin, and three different configurations are discussed in the Applications Information section. When in regulation, the LM25085 operates in continuous conduction mode at medium to heavy load currents and discontinuous conduction mode at light load currents. In continuous conduction mode the inductor’s current is always greater than zero, and the operating frequency remains relatively constant with load and line variations. The minimum load current for continuous conduction mode is one-half the inductor’s ripple current amplitude. In discontinuous conduction mode, where the inductor’s current reaches zero during the off-time, the operating frequency is lower than in continuous conduction mode and varies with load current. Conversion efficiency is maintained at light loads since the switching losses are reduced with the reduction in load and frequency. If the voltage at the FB pin exceeds 1.6V due to a transient overshoot or excessive ripple at VOUT the internal over-voltage comparator immediately switches off Q1. The next ontime period starts when the voltage at FB falls below the feedback reference voltage. ON-TIME TIMER The on-time of the PFET gate drive output (PGATE pin) is determined by the resistor (RT) and the input voltage (VIN), and is calculated from: (1) where Fs is the switching frequency. Equation 1 is valid only in continuous conduction mode (inductor current does not reach zero). Since the LM25085 controls the on-time inversely proportional to VIN, the switching frequency remains relatively constant as VIN is varied. If the input voltage falls to a level that is equal to or less than the regulated output voltage Q1 is held on continuously (100% duty cycle) and VOUT is approximately equal to VIN. The COT control scheme, with the feedback signal applied to a comparator rather than an error amplifier, requires no loop compensation, resulting in very fast load transient response. The LM25085 is available in both an 8 pin MSOP package and an 8 pin LLP package with an exposed pad to aid in heat dissipation. An 8 pin MSOP package without an exposed pad is also available. REGULATION CONTROL CIRCUIT The LM25085 buck DC-DC controller employs a control scheme based on a comparator and a one-shot on-timer, with the output voltage feedback compared to an internal reference voltage (1.25V). When the FB pin voltage falls below the feedback reference, Q1 is switched on for a time period determined by the input voltage and a programming resistor (RT). Following the on-time Q1 remains off until the FB voltage falls below the reference. Q1 is then switched on for another on-time period. The output voltage is set by the feedback resistors (RFB1, RFB2 in the Block Diagram). The regulated output voltage is calculated as follows: VOUT = 1.25V x (RFB2+ RFB1)/ RFB1 (2) The feedback voltage supplied to the FB pin is applied to a comparator rather than a linear amplifier. For proper operation sufficient ripple amplitude is necessary at the FB pin to switch the comparator at regular intervals with minimum delay and noise susceptibility. This ripple is normally obtained from the output voltage ripple attenuated through the feedback re11 (3) where RT is in kΩ. The minimum on-time, which occurs at maximum VIN, should not be set less than 150 ns (see Current Limiting section). The buck regulator effective on-time, measured at the SW node (junction of Q1, L1, and D1) is typically longer than that calculated in Equation 3 due to the asymmetric delay of the PFET. The on-time difference caused by the PFET switching delay can be estimated as the difference of the turn-off and turn-on delays listed in the PFET data sheet. Measuring the difference between the on-time at the PGATE pin versus the SW node in the actual application circuit is also recommended. In continuous conduction mode, the inverse relationship of tON with VIN results in a nearly constant switching frequency as VIN is varied. The operating frequency can be calculated from: (4) where RT is in kΩ, and tD is equal to 50 ns plus the PFET’s delay difference. To set a specific continuous conduction mode switching frequency (FS), the RT resistor is determined from the following: (5) where RT is in kΩ. A simplified version of Equation 5 at VIN = 12V, and tD = 100 ns, is: www.national.com LM25085 For VIN = 42V and tD = 100 ns, the simplified equation is: SHUTDOWN The LM25085 can be shutdown by grounding the RT pin (see Figure 1). In this mode the PFET is held off, and the VCC regulator is disabled. The internal operating current is reduced to the value shown in the graph “Shutdown current vs. VIN”. The shutdown threshold at the RT pin is ≊0.73V, with ≊50 mV of hysteresis. Releasing the pin enables normal operation. The RT pin must not be forced high during normal operation. When using equation 6 or 7, the tolerances for the ADJ pin current sink and the offset of the current limit comparator should be included to ensure the resulting minimum current limit is not less than the required maximum switch current. Simultaneously increasing the values of RADJ and RSEN decreases the effects of the current limit comparator offset, but at the expense of higher power dissipation. When using a sense resistor, the RSEN resistor value should be chosen within the practical limitations of power dissipation and physical size. For example, for a 10A current limit, setting RSEN = 0.005Ω results in a power dissipation as high as 0.5W. Current sense connections to the RSEN resistor, or to Q1, must be Kelvin connections to ensure accuracy. The CADJ capacitor filters noise from the ADJ pin, and helps prevent unintended switching of the current limit comparator due to input voltage transients. The recommended value for CADJ is 1000 pF. CURRENT LIMIT OFF-TIME When the current through Q1 exceeds the current limit threshold, the LM25085 forces an off-time longer than the normal off-time defined by Equation 1. See the graph “Off-Time vs. VIN and VFB”, or calculate the current limit off-time from the following equation: 30079324 (8) where VIN is the input voltage, and VFB is the voltage at the FB pin at the time current limit was detected. This feature is necessary to allow the inductor current to decrease sufficiently to offset the current increase which occurred during the ontime. During the on-time, the inductor current increases an amount equal to: FIGURE 1. Shutdown Implementation CURRENT LIMITING The LM25085 current limiting operates by sensing the voltage across either the RDS(ON) of Q1, or a sense resistor, during the on-time and comparing it to the voltage across the resistor RADJ (see Figure 2). The current limit function is much more accurate and stable over temperature when a sense resistor is used. The RDS(ON) of a MOSFET has a wide process variation and a large temperature coefficient. If the voltage across RDS(ON) of Q1, or the sense resistor, is greater than the voltage across RADJ, the current limit comparator switches to turn off Q1. Current sensing is disabled for a blanking time of ≊100 ns at the beginning of the on-time to prevent false triggering of the current limit comparator due to leading edge current spikes. Because of the blanking time and the turn-on and turn-off delays created by the PFET, the on-time at the PGATE pin should not be set less than 150 ns. An on-time shorter than that may prevent the current limit detection circuit from properly detecting an over-current condition. The duration of the subsequent forced off-time is a function of the input voltage and the voltage at the FB pin, as shown in the graph “Off-time vs. VIN and VFB”. The longerthan-normal forced off-time allows the inductor current to decrease to a low level before the next on-time. This cycleby-cycle monitoring, followed by a forced off-time, provides effective protection from output load faults over a wide range of operating conditions. The voltage across the RADJ resistor is set by an internal 40 µA current sink at the ADJ pin. When using Q1’s RDS(ON) for sensing, the current at which the current limit comparator switches is calculated from: ICL = 40 µA x RADJ/RDS(ON) (6) (9) During the off-time the inductor current decreases due to the reverse voltage applied across the inductor by the output voltage, the freewheeling diode’s forward voltage (VFD), and the voltage drop due to the inductor’s series resistance (VESR). The current decrease is equal to: (10) The on-time in Equation 9 is shorter than the normal on-time since the PFET is shut off when the current limit threshold is crossed. If the off-time is not long enough, such that the current decrease (Equation 10) is less than the current increase (Equation 9), the current levels are higher at the start of the next on-time. This results in a further decrease in on-time, since the current limit threshold is crossed sooner. A balance is reached when the current changes in Equations 9 and 10 are equal. The worst case situation is that of a direct short circuit at the output terminals, where VOUT = 0 volts, as that results in the largest current increase during the on-time, and the smallest decrease during the off-time. The sum of the diode’s forward voltage and the inductor’s ESR voltage must be sufficient to ensure current runaway does not occur. Using Equations 9 and 10, this requirement can be stated as: When using a sense resistor (RSEN) the thrshold of the current limit comparator is calculated from: ICL = 40 µA x RADJ/RSEN www.national.com (7) 12 (11) LM25085 For tON in Equation 11 use the minimum on-time at the SW node. To determine this time period add the “Minimum ontime in current limit” specified in the Electrical Characteristics (tON-4) to the difference of the turn-off and turn-on delays of the PFET. For tOFF use the value in the graph “Off-Time vs. VIN and VFB”, or use Equation 8, where VFB is equal to zero volts. When using the minimum or maximum limits of those specifications to determine worst case situations, the tolerance of the minimum on-time (tON-4) and the current limit offtimes (tOFF(CL1) through tOFF(CL4)) track each other over the process and temperature variations. A device which has an on-time at the high end of the range will have an off-time that is at the high end of its range. 30079325 FIGURE 2. Current Limit Sensing VCC REGULATOR The VCC regulator provides a regulated voltage between the VIN and the VCC pins to provide the bias and gate current for the PFET gate driver. The 0.47 µF capacitor at the VCC pin must be a low ESR capacitor, preferably ceramic as it provides the high surge current for the PFET’s gate at each turnon. The capacitor must be located as close as possible to the VIN and VCC pins to minimize inductance in the PC board traces. Referring to the graph “VCC vs. VIN”, the voltage across the VCC regulator (VIN – VCC) is equal to VIN until VIN reaches approximately 8.5V. At higher values of VIN, the voltage at the VCC pin is regulated at approximately 7.7V below VIN. The VCC regulator has a maximum current capability of at least 20 mA. The regulator is disabled when the LM25085 is shutdown using the RT pin, or when the thermal shutdown is activated. PGATE DRIVER OUTPUT The PGATE pin output swings between VIN (Q1 off) and the VCC pin voltage (Q1 on). The rise and fall times depend on the PFET gate capacitance and the source and sink currents provided by the internal gate driver. See the Electrical Charateristics for the current capability of the driver. P-CHANNEL MOSFET SELECTION The PFET must be rated for the maximum input voltage, with some margin above that to allow for transients and ringing which can occur on the supply line and the switching node. The gate-to-source voltage (VGS) normally provided to the PFET is 7.7 volts for VIN greater than 8.5V. However, if the circuit is to be operated at lower values of VIN, the selected PFET must be able to fully turn-on with a VGS voltage equal to VIN. The minimum input operating voltage for the LM25085 is 4.5V. Similar to NFETs, the case or exposed thermal pad for a PFET is electrically connected to the drain terminal. When 13 designing a PFET buck regulator the drain terminal is connected to the switching node. This situation requires a tradeoff between thermal and EMI performance since increasing the PC board area of the switching node to aid the PFET power dissipation also increases radiated noise, possibly disrupting the circuit operation. Typically the switching node area is kept to a reasonable minimum and the PFET peak current is derated to stay within the recommended temperature rating of the PFET. The RDS(ON) of the PFET determines a portion of the power dissipation in the PFET. However, PFETs with very low RDS(ON) usually have large values of gate charge. A PFET with a higher gate charge has a corresponding slower switching speed, leading to higher switching losses and affecting the PFET power dissipation. If the PFET RDS(ON) is used for current limit detection, note that it typically has a positive temperature coefficient. At 100° C the RDS(ON) may be as much as 50% higher than the value at 25°C which could result in incorrect current limiting if not accounted for when determining the value of the RADJ resistor. The PFET Total Gate Charge determines most of the power dissipation in the LM25085 due to the repetitive charge and discharge of the PFET’s gate capacitance by the gate driver (powered from the VCC regulator). The LM25085’s internal power dissipation can be calculated from the following: PDISS = VIN x ((QG x FS) + IIN) (12) where QG is the PFET's Total Gate Charge obtained from its datasheet, FS is the switching frequency, and IIN is the LM25085's operating current obtained from the graph "Input Operating Current vs. VIN". Using the Thermal Resistance specifications in the Electrical Characteristics table, the approximate junction temperature can be determined. If the calculated junction temperature is near the maximum operating temperature of 125°C, either the switching frequency must be reduced, or a PFET with a smaller Total Gate Charge must be used. www.national.com LM25085 SOFT-START The internal soft-start feature of the LM25085 allows the regulator to gradually reach a steady state operating point at power up, thereby reducing startup stresses and current surges. Upon turn-on, when Vcc reaches its under-voltage lockout threshold, the internal soft-start circuit ramps the feedback reference voltage from 0V to 1.25V, causing VOUT to ramp up in a proportional manner. The soft-start ramp time is typically 2.5 ms. In addition to controlling the initial power up cycle, the softstart circuit also activates when the LM25085 is enabled by releasing the RT pin, and when the circuit is shutdown and restarted by the internal Thermal Shutdown circuit. If the voltage at FB is below the regulation threshold value due to an over-current condition or a short circuit at Vout, the internal reference voltage provided by the soft-start circuit to the regulation comparator is reduced along with FB. When the over-current or short circuit condition is removed, VOUT returns to the regulated value at a rate determined by the softstart ramp. This feature helps prevent the output voltage from over-shooting following an overload event. THERMAL SHUTDOWN The LM25085 should be operated such that the junction temperature does not exceed 125°C. If the junction temperature increases above that, an internal Thermal Shutdown circuit activates at 170°C (typical) to disable the VCC regulator and the gate driver, and discharge the soft-start capacitor. This feature helps prevent catastrophic failures from accidental device overheating. When the junction temperature falls below 150°C (typical hysteresis = 20°C), the gate driver is enabled, the soft-start circuit is released, and normal operation resumes. A standard value 90.9 kΩ resistor is selected. Using equation 3 the minimum on-time at the PGATE pin, which occurs at maximum input voltage (42V), is calculated to be 381 ns. This minimum one-shot period is sufficiently longer than the minimum recommended value of 150 ns. The minimum on-time at the SW node is longer due to the delay added by the PFET (57 ns). Therefore the minimum SW node on-time is 438 ns at 42V. At the SW node the maximum on-time is calculated to be 2.55 µs at 7V. L1: The main parameter controlled by the inductor value is the current ripple amplitude (IOR). See Figure 3. The minimum load current for continuous conduction mode is used to determine the maximum allowable ripple such that the inductor current’s lower peak does not fall below 0 mA. Continuous conduction mode operation at minimum load current is not a requirement of the LM25085, but serves as a guideline for selecting L1. For this example, the maximum ripple current is: IOR(max) = 2 x IOUT(min) = 1.2 Amp (13) If an application’s minimum load current is zero, a good initial estimate for the maximum ripple current (IOR(max)) is 20% of the maximum load current. The ripple calculated in equation 13 is then used in the following equation to calculate L1: (14) A standard value 15 µH inductor is selected. Using this inductance value, the maximum ripple current amplitude, which occurs at maximum input voltage, calculates to 1.08 Ap-p. The peak current (IPK) at maximum load current is 5.54A. However, the current rating of the selected inductor must be based on the maximum current limit value calculated below. Applications Information EXTERNAL COMPONENTS The procedure for calculating the external components is illustrated with the following design example. Referring to the Block Diagram, the circuit is to be configured for the following specifications: VOUT = 5V VIN = 7V to 42V, 12V Nominal Maximum load current (IOUT(max)) = 5A Minimum load current (IOUT(min)) = 600 mA (for continuous conduction mode) Switching Frequency (FSW) = 300 kHz Maximum allowable output ripple (VOS) = 5 mVp-p Selected PFET: Vishay Si7465 RFB1 and RFB2: These resistors set the output voltage. The ratio of these resistors is calculated from: RFB2/RFB1 = (VOUT/1.25V) - 1 For this example, RFB2 / RFB1 = 3. Typically, RFB1 and RFB2 should be chosen from standard value resistors in the range of 1 kΩ to 20 kΩ which satisfy the above ratio. For this example, RFB2 = 10 kΩ, and RFB1 = 3.4 kΩ. RT, PFET: Before selecting the RT resistor, the PFET must be selected as its turn-on and turn-off delays affect the calculated value of RT. For the Vishay Si7465 PFET, the difference of its typical turn-off and turn-on delays is 57 ns. Using equation 5 at nominal input voltage, RT calculates to be: 30079332 FIGURE 3. Inductor Current Waveform RSEN, RADJ: To achieve good current limit accuracy and avoid over designing the power stage components, the sense resistor method is used for current limiting in this example. A standard value 10 mΩ resistor is selected for RSEN, resulting in a 50 mV drop at maximum load current, and a maximum 0.25W power dissipation in the resistor. Since the LM25085 uses peak current detection, the minimum value for the current limit threshold must be equal to the maximum load current (5A) plus half the maximum ripple amplitude calculated above: ICL(min) = 5A + 1.08A/2 = 5.54A At this current level the voltage across RSEN is 55.4 mV. Adding the current limit comparator offset of 9 mV (max) in- www.national.com 14 LM25085 creases the required current limit threshold to 6.44A. Using equation 7 with the minimum value for the ADJ pin current (32 µA), the required RADJ resistor calculates to: where VSW is the absolute value of the voltage at the SW node during the off-time, typically 0.5V to 1V depending on the diode D1. Using a typical value of 0.65V, VA calculates to 4.81V. VA is the nominal DC voltage at the R3/C1 junction, and is used in the next equation: A standard value 2.1 kΩ resistor is selected. The nominal current limit threshold calculates to: where tON is the maximum on-time (at minimum input voltage), and ΔV is the desired ripple amplitude at the R3/C1 junction, typically 25 mVp-p. For this example Using the tolerances for the ADJ pin current and the current limit comparator offset, the maximum current limit threshold calculates to: R3 and C1 are then selected from standard value components to produce the product calculated above. Typical values for C1 are 3000 pF to 10,000 pF, and R3 is typically from 10 kΩ to 300 kΩ. C2 is then chosen large compared to C1, typically 0.1 µF. For this example, 3300 pF is chosen for C1, requiring R3 to be 67.7 kΩ. A standard value 66.5 kΩ resistor is selected. CIN, CBYP: These capacitors limit the voltage ripple at VIN by supplying most of the switch current during the on-time. At maximum load current, when Q1 is switched on, the current through Q1 suddenly increases to the lower peak of the inductor’s ripple current, then ramps up to the upper peak, and then drops to zero at turn-off. The average current during the on-time is the load current. For a worst case calculation, these capacitors must supply this average load current during the maximum on-time, while limiting the voltage drop at VIN. For this example, 0.5V is selected as the maximum allowable droop at VIN. Their minimum value is calculated from: The minimum current limit thresholds calculate to: The load current in each case is equal to the current limit threshold minus half the current ripple amplitude. The recommended value of 1000 pF for CADJ is used in this example. COUT: Since the maximum allowed output ripple voltage is very low in this example (5 mVp-p), the minimum ripple configuration (R3, C1, and C2 in the Block Diagram) must be used. The resulting ripple at VOUT is then due to the inductor’s ripple current passing through COUT. This capacitor’s value can be selected based on the maximum allowable ripple voltage at VOUT, or based on transient response requirements. The following calculation, based on ripple voltage, provides a first order result for the value of COUT: where IOR(max) is the maximum ripple current calculated above, and VRIPPLE is the allowable ripple at VOUT. A 100 µF capacitor is selected. Typically the ripple amplitude will be higher than the calculations indicate due to the capacitor’s ESR. R3, C1, C2: The minimum ripple configuration uses these three components to generate the ripple voltage required at the FB pin since there is insufficient ripple at VOUT. A minimum of 25 mVp-p must be applied to the FB pin to obtain stable constant frequency operation. R3 and C1 are selected to generate a sawtooth waveform at their junction, and that waveform is AC coupled to the FB pin via C2. The values of the three components are determined using the following procedure: Calculate VA = VOUT - (VSW x (1 – (VOUT/VIN(min)))) 15 A 33 µF electrolytic capacitor is selected for CIN, and a 1 µF ceramic capacitor is selected for CBYP. Due to the ESR of CIN, the ripple at VIN will likely be higher than the calculation indicates, and therefore it may be desirable to increase CIN to 47 µF or 68 µF. CBYP must be located as close as possible to the VIN and GND pins of the LM25085. The voltage rating for both capacitors must be at least 42V. The RMS ripple current rating for the input capacitors must also be considered. A good approximation for the required ripple current rating is IRMS > IOUT/2. D1: A Schottky diode is recommended. Ultra-fast recovery diodes are not recommended as the high speed transitions at the SW pin may affect the regulator’s operation due to the diode’s reverse recovery transients. The diode must be rated for the maximum input voltage, and the worst case current limit level. The average power dissipation in the diode is calculated from: PD1 = VF x IOUT x (1-D) where VF is the diode’s forward voltage drop, and D is the ontime duty cycle. Using Equation 1, the minimum duty cycle occurs at maximum input voltage, and is calculated to be ≊11.9% in this example. The diode power dissipation calculates to be: PD1 = 0.65V x 5A x (1- 0.119) = 2.86W www.national.com LM25085 CVCC: The capacitor at the VCC pin (from VIN to VCC) provides not only noise filtering and stability for the VCC regulator, but also provides the surge current for the PFET gate drive. The typical recommended value for CVCC is 0.47 µF. A good quality, low ESR, ceramic capacitor is recommended. CVCC must be located as close as possible to the VIN and VCC pins. If the selected PFET has a Total Gate Charge specification of 100 nC or larger, or if the circuit is required to operate at input voltages below 7 volts, a larger capacitor may be required. The maximum recommended value for CVCC is 1 µF. IC Power Dissipation: The maximum power dissipated in the LM25085 package is calculated using Equation 12 at the maximum input voltage. The Total Gate Charge for the Si7465 PFET is specified to be 40 nC (max) in its data sheet. Therefore the total power dissipation within the LM25085 is calculated to be: PDISS = 42V x ((40 nC x 300 kHz) + 1.3 mA) = 559 mW Using an MSOP-8EP package with a θJA of 46°C/W produces a temperature rise of 26°C from junction to ambient. Final Design Example Circuit The final circuit is shown in Figure 4, and its performance is presented in Figure 5 through Figure 8. 30079342 FIGURE 4. Example Circuit www.national.com 16 LM25085 30079305 FIGURE 5. Efficiency vs. Load Current and VIN (Circuit of Figure 4) 30079344 FIGURE 6. Frequency vs. VIN (Circuit of Figure 4) 17 www.national.com LM25085 30079345 FIGURE 7. Current Limit vs. VIN (Circuit of Figure 4) 30079346 FIGURE 8. LM25085 Power Dissipation (Circuit of Figure 4) www.national.com 18 LM25085 Alternate Output Ripple Configurations The minimum ripple configuration, using C1, C2 and R3, used in the example circuit figure 4, results in a low ripple amplitude at VOUT determined mainly by the characteristics of the output capacitor and the ripple current in L1. This configuration allows multiple ceramic capacitors to be used for VOUT if the output voltage is provided to several places on the PC board. However, if a slightly higher level of ripple at VOUT is acceptable in the application, and distributed capacitance is not used, the ripple required for the FB comparator pin can be generated with fewer external components using the circuits shown below. a) Reduced ripple configuration: In Figure 9, R3, C1 and C2 are removed (compared to Figure 4). A low value resistor (R4) is added in series with COUT, and a capacitor (Cff) is added across RFB2. Ripple is generated at VOUT by the inductor’s ripple current flowing through R4, and that ripple voltage is passed to the FB pin via Cff. The ripple at VOUT can be set as low as 25 mVp-p since it is not attenuated by RFB2 and RFB1. The minimum value for R4 is calculated from: where IOR(min) is the minimum ripple current, which occurs at minimum input voltage. The minimum value for Cff is determined from: where tON(max) is the maximum on-time, which occurs at minimum VIN. The next larger standard value capacitor should be used for Cff. 30079349 FIGURE 9. Reduced Ripple Configuration b) Lowest cost configuration: This configuration, shown in Figure 10, is the same as Figure 9 except Cff is removed. Since the ripple voltage at VOUT is attenuated by RFB2 and RFB1, the minimum ripple required at VOUT is equal to: VRIP(min) = 25 mV x (RFB2 + RFB1)/RFB1 where IOR(min) is the minimum ripple current, which occurs at minimum input voltage. The minimum value for R4 is calculated from: 19 www.national.com LM25085 30079351 FIGURE 10. Lowest Cost Ripple Generating Configuration PC Board Layout In most applications, the heat sink pad or tab of Q1 is connected to the switch node, i.e. the junction of Q1, L1 and D1. While it is common to extend the PC board pad from under these devices to aid in heat dissipation, the pad size should be limited to minimize EMI radiation from this switching node. If the PC board layout allows, a similarly sized copper pad can be placed on the underside of the PC board, and connected with as many vias as possible to aid in heat dissipation. The voltage regulation, over-voltage, and current limit comparators are very fast and can respond to short duration noise pulses. Layout considerations are therefore critical for optimum performance. The layout must be as neat and compact as possible with all the components as close as possible to their associated pins. Two major current loops conduct currents which switch very fast, requiring the loops to be as small as possible to minimize conducted and radiated EMI. The first loop is that formed by CIN, Q1, L1, COUT, and back to CIN. The second loop is that formed by D1, L1, COUT, and back to D1. The connection from the anode of D1 to the ground end of CIN must be short and direct. CIN must be as close as possible to the VIN and GND pins, and CVCC must be as close as possible to the VIN and VCC pins. If the anticipated internal power dissipation of the LM25085 will produce excessive junction temperatures during normal operation, a package option with an exposed pad must be used (MSOP-8EP or LLP-8). Effective use of the PC board ground plane can help dissipate heat. Additionally, the use of wide PC board traces, where possible, helps conduct heat away from the IC. Judicious positioning of the PC board within the end product, along with the use of any available air flow (forced or natural convection) also helps reduce the junction temperature. www.national.com 20 LM25085 Physical Dimensions inches (millimeters) unless otherwise noted 8-Lead MSOP-EP Package NS Package Number MUY08A 8-Lead MSOP Package NS Package Number MUA08A 21 www.national.com LM25085 8-Lead LLP Package NS Package Number SDA08A www.national.com 22 LM25085 Notes 23 www.national.com LM25085 42V Constant On-Time PFET Buck Switching Controller Notes For more National Semiconductor product information and proven design tools, visit the following Web sites at: www.national.com Products Amplifiers Audio Clock and Timing Data Converters Interface LVDS Power Management Switching Regulators LDOs LED Lighting Voltage References PowerWise® Solutions Temperature Sensors PLL/VCO www.national.com/amplifiers www.national.com/audio www.national.com/timing www.national.com/adc www.national.com/interface www.national.com/lvds www.national.com/power www.national.com/switchers www.national.com/ldo www.national.com/led www.national.com/vref www.national.com/powerwise WEBENCH® Tools App Notes Reference Designs Samples Eval Boards Packaging Green Compliance Distributors Quality and Reliability Feedback/Support Design Made Easy Design Support www.national.com/webench www.national.com/appnotes www.national.com/refdesigns www.national.com/samples www.national.com/evalboards www.national.com/packaging www.national.com/quality/green www.national.com/contacts www.national.com/quality www.national.com/feedback www.national.com/easy www.national.com/solutions www.national.com/milaero www.national.com/solarmagic www.national.com/training Applications & Markets Mil/Aero PowerWise® Design University Serial Digital Interface (SDI) www.national.com/sdi www.national.com/wireless www.national.com/tempsensors SolarMagic™ THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. 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