LM25116MH

LM25116 Wide Range Synchronous Buck Controller April 2007 LM25116 Wide Range Synchronous Buck Controller General Description The LM25116 is a synchronous buck controller intended for step-down regulator applications from a high voltage or widely varying input supply. The control method is based upon current mode control utilizing an emulated current ramp. Current mode control provides inherent line feed-forward, cycle by cycle current limiting and ease of loop compensation. The use of an emulated control ramp reduces noise sensitivity of the pulse-width modulation circuit, allowing reliable control of very small duty cycles necessary in high input voltage applications. The operating frequency is programmable from 50kHz to 1MHz. The LM25116 drives external high-side and low-side NMOS power switches with adaptive dead-time control. A user-selectable diode emulation mode enables discontinuous mode operation for improved efficiency at light load conditions. A low quiescent current shutdown disables the controller and consumes less than 10µA of total input current. Additional features include a high voltage bias regulator, automatic switch-over to external bias for improved efficiency, thermal shutdown, frequency synchronization, cycle by cycle current limit and adjustable line under-voltage lockout. The device is available in a power enhanced TSSOP-20 package featuring an exposed die attach pad to aid thermal dissipation. Features ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ Emulated peak current mode Wide operating range up to 42V Low IQ shutdown ( 4.5V, VCCX is internally connected to VCC and the internal VCC regulator is disabled. If VCCX is unused, it should be connected to ground. High-side driver supply for bootstrap gate drive. Connect to the cathode of the bootstrap diode and the positive terminal of the bootstrap capacitor. The bootstrap capacitor supplies current to charge the high-side MOSFET gate and should be placed as close to the controller as possible. Connect to the gate of the high-side synchronous MOSFET through a short, low inductance path Switch node. Connect to the negative terminal of the bootstrap capacitor and the source terminal of the highside MOSFET. Exposed pad. Solder to ground plane. 19 20 EP HO SW EP 3 www.national.com LM25116 Absolute Maximum Ratings (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN to GND VCC, VCCX, UVLO to GND (Note 3) SW, CS to GND HB to SW HO to SW VOUT to GND CSG to GND LO to GND SS to GND FB to GND DEMB to GND -0.3V to 45V -0.3 to 16V -3.0 to 45V -0.3 to 16V -0.3 to HB+0.3V -0.3 to 45V -1V to 1V -0.3 to VCC+0.3V -0.3 to 7V -0.3 to 7V -0.3 to VCC RT to GND EN to GND ESD Rating HBM (Note 2) Storage Temperature Range Junction Temperature -0.3 to 7V -0.3 to 45V 2 kV -55°C to +150°C +150°C (Note 1) 6V to 42V 4.75V to 15V 4.75V to 15V -0.3V to 2V -40°C to +125°C Operating Ratings VIN VCC, VCCX HB to SW DEMB to GND Junction Temperature Note: RAMP, COMP are output pins. As such they are not specified to have an external voltage applied. Electrical Characteristics Symbol VIN Supply IBIAS IBIASX ISTDBY VCC Regulator VCC(REG) VCC Regulation Parameter Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature range of -40°C to +125°C and are provided for reference only. Unless otherwise specified, the following conditions apply: VIN = 24V, VCC = 7.4V, VCCX = 0V, EN = 5V, RT = 16kΩ, no load on LO and HO. Conditions VCCX = 0V VCCX = 5V EN = 0V 7.1 VIN = 6V VCC = 0V VCCX Rising ICCX = 10mA VCCX = 0V VCCX = 3V VCC Rising HB-SW = 15V 4.3 5.0 15 4.3 Min Typ 4.6 1 1 7.4 10.6 5.9 26 4.5 0.25 3.8 -200 100 4.5 0.2 125 200 0.5 3.3 VEN = 3V VEN = 0.5V VEN = 42V UVLO Rising UVLO = 0V 1.170 -7.5 -1 -3 0 15 1.215 0.1 5.4 80 210 1.262 1 1 4.7 6.2 4.7 6.0 Max 6.5 1.5 10 7.7 Units mA mA µA V V V mA V V Ω nA kΩ V V µA V V µA µA µA V V µA Ω VIN Operating Current VIN Operating Current VIN Shutdown Current VCC LDO Mode Turn-off VCC Regulation VCC Sourcing Current Limit VCCX Switch Threshold VCCX Switch Hysteresis VCCX Switch RDS(ON) VCCX Leakage VCCX Pull- down Resistance VCC Under-voltage Threshold VCC Under-voltage Hysteresis HB DC Bias Current EN Input VIL max VIH min EN Input Low Threshold EN Input High Threshold EN Input Bias Current EN Input Bias Current EN Input Bias Current UVLO Thresholds UVLO Standby Threshold UVLO Threshold Hysteresis UVLO Pull-up Current Source UVLO Pull-down RDS(ON) www.national.com 4 LM25116 Symbol Soft Start Parameter SS Current Source SS Diode Emulation Ramp Disable Threshold SS to FB Offset SS Output Low Voltage Conditions SS = 0V SS Rising FB = 1.25V Sinking 100µA, UVLO = 0V Measured at FB pin, FB = COMP FB = 2V Min 8 Typ 11 3 160 45 Max 14 Units µA V mV mV Error Amplifier VREF FB Reference Voltage FB Input Bias Current COMP Sink/Source Current AOL fBW PWM Comparators tHO(OFF) tON(min) Oscillator fSW1 fSW2 Frequency 1 Frequency 2 RT output voltage RT sync positive threshold Current Limit VCS(TH) VCS(THX) Cycle-by-cycle Sense Voltage Threshold (CSG-CS) Cycle-by-cycle Sense Voltage Threshold (CSG-CS) CS Bias Current CS Bias Current CSG Bias Current Current Limit Fault Timer RAMP Generator IR1 IR2 RAMP Current 1 RAMP Current 2 VOUT Bias Current RAMP Output Low Voltage Diode Emulation SW Zero Cross Threshold DEMB Output Current DEMB Output Current DEMB Output Current LO Gate Driver VOLL VOHL LO Low-state Output Voltage LO High-state Output Voltage LO Rise Time LO Fall Time IOHL IOLL Peak LO Source Current Peak LO Sink Current ILO = 100mA ILO = -100mA, VOHL = VCC VLO C-load = 1000pF C-load = 1000pF VLO = 0V VLO = VCC 0.08 0.25 18 12 1.8 3.5 0.17 V V ns ns A A DEMB = 0V, SS = 1.25V DEMB =0V, SS = 2.8V DEMB = 0V, SS = Regulated by FB 1.6 28 45 -6 2.7 38 65 3.8 48 85 mV µA µA µA VIN = 40V, VOUT=10V VIN = 10V, VOUT = 10V VOUT = 36V VIN = 40V, VOUT = 10V 150 21 180 28 200 265 220 35 µA µA µA mV VCCX = 0V, RAMP = 0V VCCX = 5V, RAMP = 0V CS = 42V CS = 0V CSG = 0V RT = 16kΩ, (200kHz), (256 clock cycles) 94 105 -1 90 90 1.28 110 122 126 139 1 125 125 mV mV µA µA µA ms RT = 16kΩ RT = 5kΩ 180 480 1.191 3.0 200 535 1.215 3.5 220 590 1.239 4.0 kHz kHz V V Forced HO Off-time Minimum HO On-time VIN = 42V, CRAMP = 50pF 320 450 100 580 ns ns DC Gain Unity Gain Bandwidth 1.195 1.215 15 3 80 3 1.231 500 V nA mA dB MHz 5 www.national.com LM25116 Symbol HO Gate Driver VOLH VOHH Parameter HO Low-state Output Voltage HO High-state Output Voltage HO Rise Time HO High-side Fall Time Conditions IHO = 100mA IHO = -100mA, VOHH = VHB – VHO C-load = 1000pF C-load = 1000pF VHO = 0V VHO = VCC Min Typ 0.17 0.45 19 13 1 2.2 3 Max 0.27 Units V V ns ns A A V ns ns °C °C °C/W °C/W IOHH IOLH Peak HO Source Current Peak HO Sink Current HB to SW under-voltage Switching Characteristics LO Fall to HO Rise Delay HO Fall to LO Rise Delay Thermal TSD θJA θJC Thermal Shutdown Thermal Shutdown Hysteresis Junction to Ambient Junction to Case Rising 170 15 40 4 C-load = 0 C-load = 0 75 70 Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Operating Ratings are conditions under which operation of the device is guaranteed. Operating Ratings do not imply guaranteed performance limits. For guaranteed performance limits and associated test conditions, see the Electrical Characteristics tables. Note 2: The human body model is a 100pF capacitor discharged through a 1.5kΩ resistor into each pin. LO, HO and HB are rated at 1kV. 2kV rating for all pins except VIN which is rated for 1.5kV. Note 3: These pins must not exceed VIN. www.national.com 6 LM25116 Typical Performance Characteristics Typical Application Circuit Efficiency Driver Source Current vs VCC 30015603 30015604 Driver Dead-time vs Temperature HO High RDS(ON) vs VCC 30015605 30015606 Driver Sink Current vs VCC HO Low RDS(ON) vs VCC 30015607 30015608 7 www.national.com LM25116 LO High RDS(ON) vs VCC EN Input Threshold vs Temperature 30015609 30015610 LO Low RDS(ON) vs VCC HB to SW UVLO vs Temperature 30015611 30015612 Forced HO Off-time vs Temperature VCCX = 5V HB DC Bias Current vs Temperature 30015614 30015613 www.national.com 8 LM25116 Frequency vs RT Error Amp Gain vs Frequency 30015615 30015616 Frequency vs Temperature Error Amp Phase vs Frequency 30015617 30015618 Frequency vs Temperature Current Limit Threshold vs. Temperature 30015619 30015620 9 www.national.com LM25116 VIN Operating Current vs Temperature VCC vs Temperature 30015621 30015622 VCC UVLO vs Temperature VCC vs VIN 30015623 30015624 VCC vs ICC VCCX Switch RDS(ON) vs VCCX 30015625 30015626 www.national.com 10 LM25116 Block Diagram and Typical Application Circuit 30015627 11 FIGURE 1. www.national.com LM25116 Detailed Operating Description The LM25116 high voltage switching regulator features all of the functions necessary to implement an efficient high voltage buck regulator using a minimum of external components. This easy to use regulator integrates high-side and low-side MOSFET drivers capable of supplying peak currents of 2 Amps. The regulator control method is based on current mode control utilizing an emulated current ramp. Emulated peak current mode control provides inherent line feed-forward, cycle by cycle current limiting and ease of loop compensation. The use of an emulated control ramp reduces noise sensitivity of the pulse-width modulation circuit, allowing reliable processing of the very small duty cycles necessary in high input voltage applications. The operating frequency is user programmable from 50kHz to 1MHz. An oscillator/synchronization pin allows the operating frequency to be set by a single resistor or synchronized to an external clock. Fault protection features include current limiting, thermal shutdown and remote shutdown capability. An under-voltage lockout input allows regulator shutdown when the input voltage is below a user selected threshold, and an enable function will put the regulator into an extremely low current shutdown via the enable input. The TSSOP-20EP package features an exposed pad to aid in thermal dissipation. An output voltage derived bias supply can be applied to the VCCX pin to reduce the IC power dissipation. If the bias supply voltage is greater than 4.5V, the internal regulator will essentially shut off, reducing the IC power dissipation. The VCC regulator series pass transistor includes a diode between VCC and VIN that should not be forward biased in normal operation. For an output voltage between 5V and 15V, VOUT can be connected directly to VCCX. For VOUT < 5V, a bias winding on the output inductor can be added to VOUT. If the bias winding can supply VCCX greater than VIN, an external blocking diode is required from the input power supply to the VIN pin to prevent VCC from discharging into the input supply. In high voltage applications extra care should be taken to ensure the VIN pin does not exceed the absolute maximum voltage rating of 45V. During line or load transients, voltage ringing on the VIN line that exceeds the Absolute Maximum Ratings can damage the IC. Both careful PC board layout and the use of quality bypass capacitors located close to the VIN and GND pins are essential. Enable The LM25116 contains an enable function allowing a very low input current shutdown. If the enable pin is pulled below 0.5V, the regulator enters shutdown, drawing less than 10µA from the VIN pin. Raising the EN input above 3.3V returns the regulator to normal operation. The EN pin can be tied directly to VIN if this function is not needed. It must not be left floating. A 1MΩ pull-up resistor to VIN can be used to interface with an open collector control signal. High Voltage Start-Up Regulator The LM25116 contains a dual mode internal high voltage startup regulator that provides the VCC bias supply for the PWM controller and a boot-strap gate drive for the high-side buck MOSFET. The input pin (VIN) can be connected directly to an input voltage source as high as 42 volts. For input voltages below 10.6V, a low dropout switch connects VCC directly to VIN. In this supply range, VCC is approximately equal to VIN. For VIN voltages greater than 10.6V, the low dropout switch is disabled and the VCC regulator is enabled to maintain VCC at approximately 7.4V. The wide operating range of 6V to 42V is achieved through the use of this dual mode regulator. The output of the VCC regulator is current limited to 26mA (typical). Upon power-up, the regulator sources current into the capacitor connected to the VCC pin. When the voltage at the VCC pin exceeds 4.5V and the UVLO pin is greater than 1.215V, the output switch is enabled and a soft-start sequence begins. The output switch remains enabled until VCC falls below 4.5V, EN is pulled low, the UVLO pin falls below 1.215V or the die temperature exceeds the thermal limit threshold. 30015649 FIGURE 3. Enable Circuit 30015648 30015650 FIGURE 2. VCCX Bias Supply with Additional Inductor Winding FIGURE 4. EN Bias Current vs Voltage www.national.com 12 LM25116 UVLO An under-voltage lockout pin is provided to disable the regulator without entering shutdown. If the UVLO pin is pulled below 1.215V, the regulator enters a standby mode of operation with the soft-start capacitor discharged and outputs disabled, but with the VCC regulator running. If the UVLO input is pulled above 1.215V, the controller will resume normal operation. A voltage divider from input to ground can be used to set a VIN threshold to disable the supply in brown-out conditions or for low input faults. The UVLO pin has a 5µA internal pull up current that allows this pin to left open if the input under-voltage lockout function is not needed. The UVLO pin can also be used to implement a “hiccup” current limit. If a current limit fault exists for more than 256 consecutive clock cycles, the UVLO pin will be internally pulled down to 200mV and then released. A capacitor to ground connected to the UVLO pin will set the timing for hiccup mode current limit. When this feature is used in conjunction with the voltage divider, a diode across the top resistor may be used to discharge the capacitor in the event of an input under-voltage condition. tion pulse detection. A 5V amplitude signal and 100pF coupling capacitor are recommended. The free-running frequency should be set nominally 15% below the external clock. Synchronizing above twice the free-running frequency may result in abnormal behavior of the pulse width modulator. Error Amplifier and PWM Comparator The internal high-gain error amplifier generates an error signal proportional to the difference between the regulated output voltage and an internal precision reference (1.215V). The output of the error amplifier is connected to the COMP pin allowing the user to provide loop compensation components, generally a type II network. This network creates a pole at very low frequency, a mid-band zero, and a noise reducing high frequency pole. The PWM comparator compares the emulated current sense signal from the RAMP generator to the error amplifier output voltage at the COMP pin. Ramp Generator The ramp signal used in the pulse width modulator for current mode control is typically derived directly from the buck switch current. This switch current corresponds to the positive slope portion of the inductor current. Using this signal for the PWM ramp simplifies the control loop transfer function to a single pole response and provides inherent input voltage feed-forward compensation. The disadvantage of using the buck switch current signal for PWM control is the large leading edge spike due to circuit parasitics that must be filtered or blanked. Also, the current measurement may introduce significant propagation delays. The filtering, blanking time and propagation delay limit the minimal achievable pulse width. In applications where the input voltage may be relatively large in comparison to the output voltage, controlling small pulse widths and duty cycles is necessary for regulation. The LM25116 utilizes a unique ramp generator which does not actually measure the buck switch current but rather reconstructs the signal. Representing or emulating the inductor current provides a ramp signal to the PWM comparator that is free of leading edge spikes and measurement or filtering delays. The current reconstruction is comprised of two elements, a sample-and-hold DC level and an emulated current ramp. Oscillator and Sync Capability The LM25116 oscillator frequency is set by a single external resistor connected between the RT/SYNC pin and the AGND pin. The resistor should be located very close to the device and connected directly to the pins of the IC (RT/SYNC and AGND). To set a desired oscillator frequency (fSW), the necessary value for the resistor can be calculated from the following equation: Where T = 1 / fSW and RT is in ohms. 450ns represents the fixed minimum off time. The RT/SYNC pin can be used to synchronize the internal oscillator to an external clock. The external clock must be a higher frequency than the free-running frequency set by the RT resistor. The internal oscillator can be synchronized to an external clock by AC coupling a positive edge into the RT/ SYNC pin. The voltage at the RT/SYNC pin is nominally 1.215V and must exceed 4V to trip the internal synchroniza- 30015646 FIGURE 5. Composition of Current Sense Signal 13 www.national.com LM25116 The sample-and-hold DC level is derived from a measurement of the recirculating current through either the low-side MOSFET or current sense resistor. The voltage level across the MOSFET or sense resistor is sampled and held just prior to the onset of the next conduction interval of the buck switch. The current sensing and sample-and-hold provide the DC level of the reconstructed current signal. The positive slope inductor current ramp is emulated by an external capacitor connected from the RAMP pin to the AGND and an internal voltage controlled current source. The ramp current source that emulates the inductor current is a function of the VIN and VOUT voltages per the following equation: IR = 5µA/V x (VIN-VOUT) + 25µA Proper selection of the RAMP capacitor (CRAMP) depends upon the value of the output inductor (L) and the current sense resistor (RS). For proper current emulation, the DC sample and hold value and the ramp amplitude must have the same dependence on the load current. That is: The DC current sample is obtained using the CS and CSG pins connected to either a source sense resistor (RS) or the RDS(ON) of the low-side MOSFET. For RDS(ON) sensing, RS = RDS(ON) of the low-side MOSFET. In this case it is sometimes helpful to adjust the current sense amplifier gain (A) to a lower value in order to obtain the desired current limit. Adding external resistors RG in series with CS and CSG, the current sense amplifier gain A becomes: Current Limit The LM25116 contains a current limit monitoring scheme to protect the circuit from possible over-current conditions. When set correctly, the emulated current sense signal is proportional to the buck switch current with a scale factor determined by the current limit sense resistor. The emulated ramp signal is applied to the current limit comparator. If the emulated ramp signal exceeds 1.6V, the current cycle is terminated (cycle-by-cycle current limiting). Since the ramp amplitude is proportional to VIN - VOUT, if VOUT is shorted, there is an immediate reduction in duty cycle. To further protect the external switches during prolonged current limit conditions, an internal counter counts clock pulses when in current limit. When the counter detects 256 consecutive clock cycles, the regulator enters a low power dissipation hiccup mode of current limit. The regulator is shut down by momentarily pulling UVLO low, and the soft-start capacitor discharged. The regulator is restarted with a full soft-start cycle once UVLO charges back to 1.215V. This process is repeated until the fault is removed. The hiccup off-time can be controlled by a capacitor to ground on the UVLO pin. In applications with low output inductance and high input voltage, the switch current may overshoot due to the propagation delay of the current limit comparator. If an overshoot should occur, the sampleand-hold circuit will detect the excess recirculating current. If the sample-and-hold DC level exceeds the internal current limit threshold, the buck switch will be disabled and skip pulses until the current has decayed below the current limit threshold. This approach prevents current runaway conditions due to propagation delays or inductor saturation since the inductor current is forced to decay following any current overshoot. Where gm is the ramp generator transconductance (5µA/V) and A is the current sense amplifier gain (10V/V). The ramp capacitor should be located very close to the device and connected directly to the pins of the IC (RAMP and AGND). The difference between the average inductor current and the DC value of the sampled inductor current can cause instability for certain operating conditions. This instability is known as sub-harmonic oscillation, which occurs when the inductor ripple current does not return to its initial value by the start of next switching cycle. Sub-harmonic oscillation is normally characterized by observing alternating wide and narrow pulses at the switch node. Adding a fixed slope voltage ramp (slope compensation) to the current sense signal prevents this oscillation. The 25µA of offset current provided from the emulated current source adds the optimal slope compensation to the ramp signal for a 5V output. For higher output voltages, additional slope compensation may be required. In these applications, the ramp capacitor can be decreased from its nominal value to increase the ramp slope compensation. 30015647 FIGURE 6. RDS(ON) Current Sensing without Diode Emulation www.national.com 14 LM25116 30015643 FIGURE 7. Current Limit and Ramp Circuit Using a current sense resistor in the source of the low-side MOSFET provides superior current limit accuracy compared to RDS(ON) sensing. RDS(ON) sensing is far less accurate due to the large variation of MOSFET RDS(ON) with temperature and part-to-part variation. The CS and CSG pins should be Kelvin connected to the current sense resistor or MOSFET drain and source. The peak current which triggers the current limit comparator is: connected to the SS pin resulting in a gradual rise of FB and the output voltage. Where tON is the on-time of the high-side MOSFET. The 1.1V threshold is the difference between the 1.6V reference at the current limit comparator and the 0.5V offset at the current sense amplifier. This offset at the current sense amplifier allows the inductor ripple current to go negative by 0.5V / (A x RS) when running full synchronous operation. Current limit hysteresis prevents chatter around the threshold when VCCX is powered from VOUT. When 4.5V < VCC < 5.8V, the 1.6V reference is increased to 1.72V. The peak current which triggers the current limit comparator becomes: 30015644 FIGURE 8. Diode Emulation Control During this initial charging of CSS to the internal reference voltage, the LM25116 will force diode emulation. That is, the low-side MOSFET will turn off for the remainder of a cycle if the sensed inductor current becomes negative. The inductor current is sensed by monitoring the voltage between SW and DEMB. As the SS capacitor continues to charge beyond 1.215V to 3V, the DEMB bias current will increase from 0µA up to 40µA. With the use of an external DEMB resistor (RDEMB), the current sense threshold for diode emulation will increase resulting in the gradual transition to synchronous operation. Forcing diode emulation during soft-start allows the LM25116 to start up into a pre-biased output without unnecessarily discharging the output capacitor. Full synchronous operation is obtained if the DEMB pin is always biased to a higher potential than the SW pin when LO is high. RDEMB = 10kΩ will bias the DEMB pin to 0.45V minimum, which is adequate for most applications. The DEMB bias potential should always be kept below 2V. When RDEMB = 0Ω, the LM25116 will always run in diode emulation. Once SS charges to 3V the SS latch is set, increasing the DEMB bias current to 65µA. An amplifier is enabled that regulates SS to 160mV above the FB voltage. This feature can prevent overshoot of the output voltage in the event the output This has the effect of a 10% fold-back of the peak current during a short circuit when VCCX is powered from a 5V output. Soft-Start and Diode Emulation The soft-start feature allows the regulator to gradually reach the initial steady state operating point, thus reducing start-up stresses and surges. The LM25116 will regulate the FB pin to the SS pin voltage or the internal 1.215V reference, whichever is lower. At the beginning of the soft-start sequence when SS = 0V, the internal 10µA soft-start current source gradually increases the voltage of an external soft-start capacitor (CSS) 15 www.national.com LM25116 voltage momentarily dips out of regulation. When a fault is detected (VCC under-voltage, UVLO pin < 1.215, or EN = 0V) the soft-start capacitor is discharged. Once the fault condition is no longer present, a new soft-start sequence begins. OUTPUT INDUCTOR The inductor value is determined based on the operating frequency, load current, ripple current and the input and output voltages. HO Ouput The LM25116 contains a high current, high-side driver and associated high voltage level shift. This gate driver circuit works in conjunction with an external diode and bootstrap capacitor. A 1µF ceramic capacitor, connected with short traces between the HB pin and SW pin, is recommended. During the off-time of the high-side MOSFET, the SW pin voltage is approximately -0.5V and the bootstrap capacitor charges from VCC through the external bootstrap diode. When operating with a high PWM duty cycle, the buck switch will be forced off each cycle for 450ns to ensure that the bootstrap capacitor is recharged. The LO and HO outputs are controlled with an adaptive deadtime methodology which insures that both outputs are never enabled at the same time. When the controller commands HO to be enabled, the adaptive block first disables LO and waits for the LO voltage to drop below approximately 25% of VCC. HO is then enabled after a small delay. Similarly, LO is enabled once HO has discharged. This methodology insures adequate dead-time for any size MOSFET. 30015645 FIGURE 9. Inductor Current Knowing the switching frequency (fSW), maximum ripple current (IPP), maximum input voltage (VIN(MAX)) and the nominal output voltage (VOUT), the inductor value can be calculated: Thermal Protection Internal thermal shutdown circuitry is provided to protect the integrated circuit in the event the maximum junction temperature is exceeded. When activated, typically at 170°C, the controller is forced into a low power reset state, disabling the output driver and the bias regulator. This is designed to prevent catastrophic failures from accidental device overheating. The maximum ripple current occurs at the maximum input voltage. Typically, IPP is 20% to 40% of the full load current. When running diode emulation mode, the maximum ripple current should be less than twice the minimum load current. For full synchronous operation, higher ripple current is acceptable. Higher ripple current allows for a smaller inductor size, but places more of a burden on the output capacitor to smooth the ripple current for low output ripple voltage. For this example, 40% ripple current was chosen for a smaller sized inductor. Application Information EXTERNAL COMPONENTS The procedure for calculating the external components is illustrated with the following design example. The Bill of Materials for this design is listed in Table 1. The circuit shown in Figure 15 is configured for the following specifications: • Output voltage = 5V • Input voltage = 7V to 42V • Maximum load current = 7A • Switching frequency = 250kHz Simplified equations are used as a general guideline for the design method. Comprehensive equations are provided at the end of this section. TIMING RESISTOR RT sets the oscillator switching frequency. Generally, higher frequency applications are smaller but have higher losses. Operation at 250kHz was selected for this example as a reasonable compromise for both small size and high efficiency. The value of RT for 250kHz switching frequency can be calculated as follows: The nearest standard value of 6µH will be used. The inductor must be rated for the peak current to prevent saturation. During normal operation, the peak current occurs at maximum load current plus maximum ripple. During overload conditions with properly scaled component values, the peak current is limited to VCS(TH) / RS (See next section). At the maximum input voltage with a shorted output, the valley current must fall below VCS(TH) / RS before the high-side MOSFET is allowed to turn on. The peak current in steady state will increase to VIN(MAX) x tON(min) / L above this level. The chosen inductor must be evaluated for this condition, especially at elevated temperature where the saturation current rating may drop significantly. CURRENT SENSE RESISTOR The current limit is set by the current sense resistor value (RS). The nearest standard value of 12.4kΩ was chosen for RT. www.national.com 16 LM25116 For a 5V output, the maximum current sense signal occurs at the minimum input voltage, so RS is calculated from: For this example VCCX = 0V, so VCS(TH) = 0.11V. The current sense resistor is calculated as: The next lowest standard value of 10mΩ was chosen for RS. RAMP CAPACITOR With the inductor and sense resistor value selected, the value of the ramp capacitor (CRAMP) necessary for the emulation ramp circuit is: INPUT CAPACITORS The regulator supply voltage has a large source impedance at the switching frequency. Good quality input capacitors are necessary to limit the ripple voltage at the VIN pin while supplying most of the switch current during the on-time. When the buck switch turns on, the current into the switch steps to the valley of the inductor current waveform, ramps up to the peak value, and then drops to zero at turn-off. The input capacitors should be selected for RMS current rating and minimum ripple voltage. A good approximation for the required ripple current rating is IRMS > IOUT / 2. Quality ceramic capacitors with a low ESR were selected for the input filter. To allow for capacitor tolerances and voltage rating, four 2.2µF ceramic capacitors were used for the typical application circuit. With ceramic capacitors, the input ripple voltage will be triangular and peak at 50% duty cycle. Taking into account the capacitance change with DC bias, the input ripple voltage is approximated as: Where L is the value of the output inductor in Henrys, gm is the ramp generator transconductance (5µA/V), and A is the current sense amplifier gain (10V/V). For the 5V output design example, the ramp capacitor is calculated as: When the converter is connected to an input power source, a resonant circuit is formed by the line impedance and the input capacitors. If step input voltage transients are expected near the maximum rating of the LM25116, a careful evaluation of the ringing and possible overshoot at the device VIN pin should be completed. To minimize overshoot make CIN > 10 x LIN. The characteristic source impedance and resonant frequency are: The next lowest standard value of 270pF was selected for CRAMP. A COG type capacitor with 5% or better tolerance is recommended. OUTPUT CAPACITORS The output capacitors smooth the inductor ripple current and provide a source of charge for transient loading conditions. For this design example, five 100µF ceramic capacitors where selected. Ceramic capacitors provide very low equivalent series resistance (ESR), but can exhibit a significant reduction in capacitance with DC bias. From the manufacturer’s data, the ESR at 250kHz is 2mΩ / 5 = 0.4mΩ, with a 36% reduction in capacitance at 5V. This is verified by measuring the output ripple voltage and frequency response of the circuit. The fundamental component of the output ripple voltage is calculated as: The converter exhibits a negative input impedance which is lowest at the minimum input voltage: The damping factor for the input filter is given by: With typical values for the 5V design example: When δ = 1, the input filter is critically damped. This may be difficult to achieve with practical component values. With δ < 0.2, the input filter will exhibit significant ringing. If δ is zero or negative, there is not enough resistance in the circuit and the input filter will sustain an oscillation. When operating near the minimum input voltage, an aluminum electrolytic capacitor across CIN may be needed to damp the input for a typical bench test setup. Any parallel capacitor should be evaluated for its RMS current rating. The current will split between the ceramic and aluminum capacitors based on the relative impedance at the switching frequency. VCC CAPACITOR The primary purpose of the VCC capacitor (CVCC) is to supply the peak transient currents of the LO driver and bootstrap diode (D1) as well as provide stability for the VCC regulator. 17 www.national.com LM25116 These current peaks can be several amperes. The recommended value of CVCC should be no smaller than 0.47µF, and should be a good quality, low ESR, ceramic capacitor located at the pins of the IC to minimize potentially damaging voltage transients caused by trace inductance. A value of 1µF was selected for this design. BOOTSTRAP CAPACITOR The bootstrap capacitor (CHB) between the HB and SW pins supplies the gate current to charge the high-side MOSFET gate at each cycle’s turn-on as well as supplying the recovery charge for the bootstrap diode (D1). These current peaks can be several amperes. The recommended value of the bootstrap capacitor is at least 0.1µF, and should be a good quality, low ESR, ceramic capacitor located at the pins of the IC to minimize potentially damaging voltage transients caused by trace inductance. The absolute minimum value for the bootstrap capacitor is calculated as: 2. 2. With an appropriate value for RUV2, RUV1 can be selected using the following equation: 3. Where VIN(MIN) is the desired shutdown voltage. Capacitor CFT provides filtering for the divider and determines the off-time of the “hiccup” duty cycle during current limit. When CFT is used in conjunction with the voltage divider, a diode across the top resistor should be used to discharge CFT in the event of an input undervoltage condition. If under-voltage shutdown is not required, RUV1 and RUV2 can be eliminated and the off-time becomes: Where Qg is the high-side MOSFET gate charge and ΔVHB is the tolerable voltage droop on CHB, which is typically less than 5% of VCC. A value of 1µF was selected for this design. SOFT START CAPACITOR The capacitor at the SS pin (CSS) determines the soft-start time, which is the time for the reference voltage and the output voltage to reach the final regulated value. The value of CSS for a given time is determined from: The voltage at the UVLO pin should never exceed 16V when using an external set-point divider. It may be necessary to clamp the UVLO pin at high input voltages. For the design example, RUV2 = 102kΩ and RUV1 = 21kΩ for a shut-down voltage of 6.6V. If sustained short circuit protection is required, CFT ≥ 1µF will limit the short circuit power dissipation. D2 may be installed when using CFT with RUV1 and RUV2. MOSFETs Selection of the power MOSFETs is governed by the same tradeoffs as switching frequency. Breaking down the losses in the high-side and low-side MOSFETs is one way to determine relative efficiencies between different devices. When using discrete SO-8 MOSFETs the LM25116 is most efficient for output currents of 2A to 10A. Losses in the power MOSFETs can be broken down into conduction loss, gate charging loss, and switching loss. Conduction, or I2R loss PDC, is approximately: PDC(HO-MOSFET) = D x (IO2 x RDS(ON) x 1.3) PDC(LO-MOSFET) = (1 - D) x (IO2 x RDS(ON) x 1.3) Where D is the duty cycle. The factor 1.3 accounts for the increase in MOSFET on-resistance due to heating. Alternatively, the factor of 1.3 can be ignored and the on-resistance of the MOSFET can be estimated using the RDS(ON) vs Temperature curves in the MOSFET datasheet. Gate charging loss, PGC, results from the current driving the gate capacitance of the power MOSFETs and is approximated as: PGC = n x VCC x Qg x fSW Qg refer to the total gate charge of an individual MOSFET, and ‘n’ is the number of MOSFETs. If different types of MOSFETs are used, the ‘n’ term can be ignored and their gate charges summed to form a cumulative Qg. Gate charge loss differs from conduction and switching losses in that the actual dissipation occurs in the LM25116 and not in the MOSFET itself. Further loss in the LM25116 is incurred as the gate driving current is supplied by the internal linear regulator. Switching loss occurs during the brief transition period as the MOSFET turns on and off. During the transition period both current and voltage are present in the channel of the MOSFET. The switching loss can be approximated as: 18 For this application, a value of 0.01µF was chosen for a softstart time of 1.2ms. OUTPUT VOLTAGE DIVIDER RFB1 and RFB2 set the output voltage level, the ratio of these resistors is calculated from: RFB1 is typically 1.21kΩ for a divider current of 1mA. The divider current can be reduced to 100µA with RFB1=12.1kΩ. For the 5V output design example used here, RFB1 = 1.21kΩ and RFB2 = 3.74kΩ. UVLO DIVIDER A voltage divider and filter can be connected to the UVLO pin to set a minimum operating voltage VIN(MIN) for the regulator. If this feature is required, the following procedure can be used to determine appropriate resistor values for RUV2, RUV1 and CFT. 1. RUV2 must be large enough such that in the event of a current limit, the internal UVLO switch can pull UVLO < 200mV. This can be guaranteed if: RUV2 > 500 x VIN(MAX) Where VIN(MAX) is the maximum input voltage and RUV2 is in ohms. www.national.com LM25116 PSW = 0.5 x VIN x IO x (tR + tF) x fSW Where tR and tF are the rise and fall times of the MOSFET. Switching loss is calculated for the high-side MOSFET only. Switching loss in the low-side MOSFET is negligible because the body diode of the low-side MOSFET turns on before the MOSFET itself, minimizing the voltage from drain to source before turn-on. For this example, the maximum drain-tosource voltage applied to either MOSFET is 42V. VCC provides the drive voltage at the gate of the MOSFETs. The selected MOSFETs must be able to withstand 42V plus any ringing from drain to source, and be able to handle at least VCC plus ringing from gate to source. A good choice of MOSFET for the 42V input design example is the Si7850DP. It has an RDS(ON) of 20mΩ, total gate charge of 14nC, and rise and fall times of 10ns and 12ns respectively. In applications where a high step-down ratio is maintained for normal operation, efficiency may be optimized by choosing a high-side MOSFET with lower Qg, and low-side MOSFET with lower RDS(ON). For higher voltage MOSFETs which are not true logic level, it is important to use the UVLO feature. Choose a minimum operating voltage which is high enough for VCC and the bootstrap (HB) supply to fully enhance the MOSFET gates. This will prevent operation in the linear region during power-on or power-off which can result in MOSFET failure. Similar consideration must be made when powering VCCX from the output voltage. MOSFET SNUBBER A resistor-capacitor snubber network across the low-side MOSFET reduces ringing and spikes at the switching node. Excessive ringing and spikes can cause erratic operation and couple spikes and noise to the output. Selecting the values for the snubber is best accomplished through empirical methods. First, make sure the lead lengths for the snubber connections are very short. Start with a resistor value between 5Ω and 50Ω. Increasing the value of the snubber capacitor results in more damping, but higher snubber losses. Select a minimum value for the snubber capacitor that provides adequate damping of the spikes on the switch waveform at high load. ERROR AMPLIFIER COMPENSATION RCOMP, CCOMP and CHF configure the error amplifier gain characteristics to accomplish a stable voltage loop gain. One advantage of current mode control is the ability to close the loop with only two feedback components, RCOMP and CCOMP. The voltage loop gain is the product of the modulator gain and the error amplifier gain. For the 5V output design example, the modulator is treated as an ideal voltage-to-current converter. The DC modulator gain of the LM25116 can be modeled as: DC Gain(MOD) = RLOAD / (A x RS) The dominant low frequency pole of the modulator is determined by the load resistance (RLOAD) and output capacitance (COUT). The corner frequency of this pole is: fP(MOD) = 1 / (2π x RLOAD x COUT) For RLOAD = 5V / 7A = 0.714Ω and COUT = 320µF (effective) then fP(MOD) = 700Hz DC Gain(MOD) = 0.714Ω / (10 x 10mΩ) = 7.14 = 17dB For the 5V design example the modulator gain vs. frequency characteristic was measured as shown in Figure 10. 30015664 30015663 FIGURE 10. Modulator Gain and Phase Components RCOMP and CCOMP configure the error amplifier as a type II configuration. The DC gain of the amplifier is 80dB which has a pole at low frequency and a zero at fZEA = 1 / (2π x RCOMP x CCOMP). The error amplifier zero cancels the modulator pole leaving a single pole response at the crossover frequency of the voltage loop. A single pole response at the crossover frequency yields a very stable loop with 90° of phase margin. For the design example, a target loop bandwidth (crossover frequency) of one-tenth the switching frequency or 25kHz was selected. The compensation network zero (fZEA) should be selected at least an order of magnitude less than the target crossover frequency. This constrains the product of RCOMP and CCOMP for a desired compensation network zero 1 / (2π x RCOMP x CCOMP) to be 2.5kHz. Increasing RCOMP, while proportionally decreasing CCOMP, increases the error amp gain. Conversely, decreasing RCOMP while proportionally increasing CCOMP, decreases the error amp gain. For the design example CCOMP was selected as 3300pF and RCOMP was selected as 18kΩ. These values configure the compensation network zero at 2.7kHz. The error amp gain at frequencies greater than fZEA is: RCOMP / RFB2, which is approximately 4.8 (13.6dB). FIGURE 11. Error Amplifier Gain and Phase The overall voltage loop gain can be predicted as the sum (in dB) of the modulator gain and the error amp gain. 19 www.national.com LM25116 The regulator has an exposed thermal pad to aid power dissipation. Selecting MOSFETs with exposed pads will aid the power dissipation of these devices. The resulting power losses are primarily in the switching MOSFETs. Careful attention to RDS(ON) at high temperature should be observed. Also, at 250 kHz, a MOSFET with low gate capacitance will result in lower switching losses. Comprehensive Equations CURRENT SENSE RESISTOR AND RAMP CAPACITOR T = 1 / fSW, gm = 5µA/V, A = 10V/V. IOUT is the maximum output current at current limit. General Method for VOUT < 5V: 30015665 FIGURE 12. Overall Voltage Loop Gain and Phase If a network analyzer is available, the modulator gain can be measured and the error amplifier gain can be configured for the desired loop transfer function. If a network analyzer is not available, the error amplifier compensation components can be designed with the guidelines given. Step load transient tests can be performed to verify acceptable performance. The step load goal is minimum overshoot with a damped response. CHF can be added to the compensation network to decrease noise susceptibility of the error amplifier. The value of CHF must be sufficiently small since the addition of this capacitor adds a pole in the error amplifier transfer function. This pole must be well beyond the loop crossover frequency. A good approximation of the location of the pole added by CHF is: fP2 = fZEA x CCOMP / CHF. The value of CHF was selected as 100pF for the design example. PCB BOARD LAYOUT and THERMAL CONSIDERATIONS In a buck regulator there are two loops where currents are switched very fast. The first loop starts from the input capacitors, through the high-side MOSFET, to the inductor then out to the load. The second loop starts from the output capacitor ground, to the regulator PGND pins, to the current sense resistor, through the low-side MOSFET, to the inductor and then out to the load. Minimizing the area of these two loops reduces the stray inductance and minimizes noise and possible erratic operation. A ground plane in the PC board is recommended as a means to connect the input filter capacitors to the output filter capacitors and the PGND pin of the regulator. Connect all of the low power ground connections (CSS, RT, CRAMP) directly to the regulator AGND pin. Connect the AGND and PGND pins together through to topside copper area covering the entire underside of the device. Place several vias in this underside copper area to the ground plane. The input capacitor ground connection should be as close as possible to the low-side source or current sense ground connection. The highest power dissipating components are the two power MOSFETs. The easiest way to determine the power dissipated in the MOSFETs is to measure the total conversion losses (PIN - POUT), then subtract the power losses in the output inductor and any snubber resistors. If a snubber is used, the power loss can be estimated with an oscilloscope by observation of the resistor voltage drop at both turn-on and turn-off transitions. Assuming that the RC time constant is 7.5V, install a resistor from the RAMP pin to VCC. www.national.com 20 LM25116 30015673 FIGURE 13. RRAMP to VCC for VOUT > 7.5V For VOUT < 7.5V, a negative VCC is required. This can be made with a simple charge pump from the LO gate output. Install a resistor from the RAMP pin to the negative VCC. Km is the effective DC gain of the modulating comparator. The duty cycle D = VOUT / VIN. KSL is the proportional slope compensation term. VSL is the fixed slope compensation term. Slope compensation is set by mc, which is the ratio of the external ramp to the natural ramp. The switching frequency sampling gain is characterized by ωn and Q, which accounts for the high frequency inductor pole. ERROR AMPLIFIER TRANSFER FUNCTION The following equations are used to calculate the error amplifier transfer function: 30015675 FIGURE 14. RRAMP to -VCC for VOUT < 7.5V If a large variation is expected in VCC, say for VIN < 11V, a Zener regulator may be added to supply a constant voltage for RRAMP. MODULATOR TRANSFER FUNCTION The following equations can be used to calculate the controlto-output transfer function: Where AOL = 10,000 (80dB) and ωBW = 2π x fBW. GEA(S) is the ideal error amplifier gain, which is modified at DC and high frequency by the open loop gain of the amplifier and the feedback divider ratio. 21 www.national.com LM25116 www.national.com 30015642 22 5V 7A Typical Application Schematic FIGURE 15. LM25116 TABLE 1. Bill of Materials for 7V-42V Input, 5V 7A Output, 250kHz ID C1, C2, C14 C3 C4 C5, C15 C6 C7 C8, C9, C10, C11 C12 C13 C16, C17, C18, C19, C20 C21, C22 C23 D1 D2 JMP1 L1 P1-P4 TP1-TP5 Q1, Q2 R1 R2 R3 R4 R5 R6, R7 R8 R9 R10 R11 R12 R13 R14 U1 LM25116MH CRCW0603105J CRCW06030R0J CRCW0603103J CRCW06031242F CRCW0603183J LRC-LRF2010-01R010-F HC2LP-6R0 1514-2 5012 Si7850DP CRCW06031023F CRCW06032102F CRCW06033741F CRCW06031211F CMPD2003 CMPD2003 C4532X7R2A225M C3225X7R2A105M C2012X7R2A104M C4532X6S0J107M Part Number C2012X7R1E105K VJ0603Y103KXAAT VJ0603A271JXAAT VJ0603Y101KXAT W1BC VJ0603Y332KXXAT Type Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Capacitor, Ceramic Capacitor, Tantalum Capacitor, Ceramic Diode, Switching Diode, Switching Connector, Jumper Inductor Turret Terminal Test Point N-CH MOSFET Resistor Resistor Resistor Resistor Resistor Resistor Resistor Resistor Resistor Resistor Resistor Resistor Resistor Synchronous Buck Controller .090” dia. .040” dia. SO-8 Power PAK 0603 0603 0603 0603 0603 0603 0603 0603 0603 2010 0603 0603 1206 TSSOP-20EP 10.3A, 60V 102kΩ, 1% 21.0kΩ, 1% 3.74kΩ, 1% 1.21kΩ, 1% Not Used 0Ω 10kΩ, 5% 12.4kΩ, 1% 18kΩ, 5% 0.010Ω, 1% Not Used 1MΩ, 5% Not Used Size 0805 0603 0603 0603 0603 0603 1812 1210 0805 1812 D Case 0805 SOT-23 SOT-23 Parameters 1µF, 25V, X7R 0.01µF, 50V, X7R 270pF, 50V, COG, 5% 100pF, 50V, X7R 3300pF, 25V, X7R Not Used 2.2µF, 100V X7R 1µF, 100V X7R 0.1µF, 100V X7R 100µF, 6.3V, X6S, 105°C Not Used Not Used 200mA, 200V Not Used 2 pin sq. post 6µH, 16.5A Qty 3 1 1 2 1 0 4 1 1 5 0 0 1 0 1 1 4 5 2 1 1 1 1 0 2 1 1 1 1 0 1 0 1 NSC Vishay Vishay Vishay Vishay Vishay IRC Cooper Keystone Keystone Vishay Siliconix Vishay Vishay Vishay Vishay Central Semi Central Semi TDK TDK TDK TDK Vendor TDK Vishay Vishay Vishay Vishay 23 www.national.com LM25116 Physical Dimensions inches (millimeters) unless otherwise noted TSSOP-20EP Outline Drawing NS Package Number MXA20A www.national.com 24 LM25116 Notes 25 www.national.com LM25116 Wide Range Synchronous Buck Controller Notes THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. NATIONAL MAKES NO REPRESENTATIONS OR WARRANTIES WITH RESPECT TO THE ACCURACY OR COMPLETENESS OF THE CONTENTS OF THIS PUBLICATION AND RESERVES THE RIGHT TO MAKE CHANGES TO SPECIFICATIONS AND PRODUCT DESCRIPTIONS AT ANY TIME WITHOUT NOTICE. NO LICENSE, WHETHER EXPRESS, IMPLIED, ARISING BY ESTOPPEL OR OTHERWISE, TO ANY INTELLECTUAL PROPERTY RIGHTS IS GRANTED BY THIS DOCUMENT. TESTING AND OTHER QUALITY CONTROLS ARE USED TO THE EXTENT NATIONAL DEEMS NECESSARY TO SUPPORT NATIONAL’S PRODUCT WARRANTY. EXCEPT WHERE MANDATED BY GOVERNMENT REQUIREMENTS, TESTING OF ALL PARAMETERS OF EACH PRODUCT IS NOT NECESSARILY PERFORMED. NATIONAL ASSUMES NO LIABILITY FOR APPLICATIONS ASSISTANCE OR BUYER PRODUCT DESIGN. BUYERS ARE RESPONSIBLE FOR THEIR PRODUCTS AND APPLICATIONS USING NATIONAL COMPONENTS. PRIOR TO USING OR DISTRIBUTING ANY PRODUCTS THAT INCLUDE NATIONAL COMPONENTS, BUYERS SHOULD PROVIDE ADEQUATE DESIGN, TESTING AND OPERATING SAFEGUARDS. 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Copyright© 2007 National Semiconductor Corporation For the most current product information visit us at www.national.com National Semiconductor Americas Customer Support Center Email: new.feedback@nsc.com Tel: 1-800-272-9959 National Semiconductor Europe Customer Support Center Fax: +49 (0) 180-530-85-86 Email: europe.support@nsc.com Deutsch Tel: +49 (0) 69 9508 6208 English Tel: +49 (0) 870 24 0 2171 Français Tel: +33 (0) 1 41 91 8790 National Semiconductor Asia Pacific Customer Support Center Email: ap.support@nsc.com National Semiconductor Japan Customer Support Center Fax: 81-3-5639-7507 Email: jpn.feedback@nsc.com Tel: 81-3-5639-7560 www.national.com