LM2854MHX-1000

LM2854 4A 500 kHz / 1 MHz PowerWise® Synchronous SIMPLE SWITCHER® Buck Regulator April 2, 2008 LM2854 4A 500 kHz / 1 MHz PowerWise® Synchronous SIMPLE SWITCHER® Buck Regulator General Description The LM2854 SIMPLE buck regulator is a 500 kHz or 1 MHz step-down switching voltage regulator capable of driving up to a 4A load with exceptional power conversion efficiency, line and load regulation, and output accuracy. The LM2854 can accept an input voltage rail between 2.95V and 5.5V and deliver an adjustable and highly accurate output voltage as low as 0.8V. Externally established soft-start with a small capacitor facilitates controlled start-up, and the LM2854 is capable of starting gracefully into a pre-biased output voltage. Partial internal compensation reduces the number of external passive components and PC board space typically necessary in a voltage mode buck converter application, yet preserving flexibility to deal with ceramic and/or electrolytic based load capacitors. Lossless cycle-by-cycle peak current limit is used to protect the load from an overcurrent or short-circuit fault, and an enable comparator simplifies sequencing applications. The LM2854 is available in an exposed pad TSSOP-16 package that enhances the thermal performance of the regulator. PowerWise® SWITCHER® Features ■ Input voltage range of 2.95V to 5.5V ■ Maximum load current of 4A ■ Wide bandwidth voltage mode control loop, partial internal ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ compensation Fixed switching frequency of 500 kHz or 1 MHz 35 mΩ integrated MOSFET switches Adjustable output voltage down to 0.8V Optimized reference voltage initial accuracy and temperature drift External soft-start control with tracking capability Enable pin with hysteresis Low standby current of 230 µA Pre-biased load startup capability Integrated UVLO, OCP and thermal shutdown 100% duty cycle capability eTSSOP-16 exposed pad package Applications ■ Low Voltage POL Regulation from 5V or 3.3V Rail ■ Local Solution for FPGA/DSP/ASIC/µP Core or I/O Power ■ Broadband Networking and Communications Infrastructure ■ Portable Computing Typical Application Circuit 30052801 SIMPLE SWITCHER® is a Registered Trademark of National Semiconductor Corporation. © 2008 National Semiconductor Corporation 300528 www.national.com LM2854 Connection Diagram 30052802 Top View 16-Lead eTSSOP Ordering Information Order Number LM2854MH-500 LM2854MHX-500 LM2854MH-1000 LM2854MHX-1000 1 MHz Frequency 500 kHz TSSOP-16 exposed pad MXA16A Package Type Package Drawing Supplied As 92 Units, Rail 2500 Units, Tape and Reel 92 Units, Rail 2500 Units, Tape and Reel www.national.com 2 LM2854 Pin Descriptions Pin Number 1 2,3,4 5,6,7 8,9 10 Name NC PGND PVIN NC AVIN Description Reserved for factory use, this pin should be connected to GND to ensure proper operation. Power ground pins for the internal power switches. These pins should be connected together locally at the device and tied to the PC board ground plane. Input voltage to the power switches inside the device. These pins should be connected together at the device. A low ESR input capacitance should be located as close as possible to these pins. Reserved for factory use, these pins should be connected to GND to ensure proper operation. Analog input voltage supply that generates the internal bias. The UVLO circuit derives its input from this pin also. Thus, if the voltage on AVIN falls below the UVLO threshold, both internal FETs are turned off. It is recommended to connect PVIN to AVIN through a low pass RC filter to minimize the influence of input rail ripple and noise on the analog control circuitry. The series resistor should be 1Ω and the bypass capacitor should be a X7R ceramic type 0.1 µF to 1.0 µF. Active high enable input for the device. Typically, turn-on threshold is 1.23V with 0.15V hysteresis. An external resistor divider from PVIN can be used to effectively increase the UVLO turn-on threshold. If not used, the EN pin should be connected to PVIN. Switch node pins. This is the PWM output of the internal MOSFET power switches. These pins should be tied together locally and connected to the filter inductor. Soft-start control pin. An internal 2 µA current source charges an external capacitor connected between this pin and AGND to set the output voltage ramp rate during startup. This pin can also be used to configure the tracking feature. Quiet analog ground for the internal bias circuitry. Feedback pin is connected to the inverting input of the voltage loop error amplifier. A 0.8V bandgap reference is connected to the non-inverting input of the error amplifier. Exposed metal pad on the underside of the package with a weak electrical connection to PGND. It is recommended to connect this pad to the PC board ground plane in order to improve thermal dissipation. 11 EN 12,13 14 SW SS 15 16 EXP AGND FB Exposed Pad 3 www.national.com LM2854 Absolute Maximum Ratings (Notes 1, 6) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. PVIN, AVIN, SW, EN, FB, SS to GND ESD Susceptibility (Note 2) Power Dissipation Junction Temperature Storage Temperature Range -0.3V to 6.0V ±2 kV Internally Limited 150°C −65°C to +150°C Lead Temperature Soldering (10 sec) Vapor Phase (60 sec) Infrared 260°C 215°C 220°C (Note 6) 2.95V to 5.5V 2.95V to 5.5V −40°C to +125°C Operating Ratings PVIN to GND AVIN to GND Junction Temperature Electrical Characteristics Specifications with standard typeface are for TJ = 25°C only; limits in bold face type apply over the Operating Junction Temperature Range TJ range of -40°C to 125°C. Minimum and maximum limits are guaranteed through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. AVIN = PVIN = EN = 5.0V, unless otherwise indicated in the Conditions column. Symbol Parameter Conditions Min (Note 3) 0.790 Typ (Note 4) 0.8 0.04 0.25 2.6 25 170 35 34 2 4.5 Non-switching EN = 0V 1 MHz option 500 kHz option Drange ENABLE CONTROL VIH VEN(HYS) TSD TSD-HYS θJA EN Pin Rising Threshold EN Pin Hysteresis TJ for Thermal Shutdown Hysteresis for Thermal Shutdown Junction to Ambient MXA16A 0.8 1.23 150 165 10 35 1.65 V mV °C °C °C/W PWM Duty Cycle Range 800 400 0 6.0 1.7 230 1050 525 6.7 3 500 1160 580 100 2.95 375 65 65 Max (Note 3) 0.808 0.6 Units SYSTEM PARAMETERS VREF ΔVREF/ΔAVIN ΔVREF/ΔIO VON RDS(ON)-P RDS(ON)-N ISS ICL IQ ISD PWM SECTION fSW Switching Frequency kHz kHz % Reference Voltage (Note 5) Line Regulation (Note 5) Load Regulation UVLO Threshold (AVIN) PFET On Resistance NFET On Resistance Soft-Start Current Peak Current Limit Threshold Operating Current Shut Down Quiescent Current Measured at the FB pin ΔAVIN = 2.95V to 5.50V Normal operation Rising Falling hysteresis ISW = 4A ISW = 4A V % mV/A V mV mΩ mΩ µA A mA µA THERMAL CONTROL THERMAL RESISTANCE Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics. Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Test method is per JESD22-AI14. Note 3: Min and Max limits are 100% production tested at 25°C. Limits over the operating temperature range are guaranteed through correlation using Statistical Quality Control (SQC) methods. Limits are used to calculate National’s Average Outgoing Quality Level (AOQL). Note 4: Typical numbers are at 25°C and represent the most likely parametric norm. Note 5: VREF measured in a non-switching, closed-loop configuration. Note 6: PGND and AGND are electrically connected together on the PC board and the resultant net is termed GND. www.national.com 4 LM2854 Unless otherwise specified, the following conditions apply: VIN = PVIN = AVIN = EN = 5.0V, CIN is 47 µF 10V X5R ceramic capacitor, LO is from TDK SPM6530T family; TAMBIENT = 25°C for efficiency curves, bode plots and waveforms, and TJ = 25°C for all others. Feedback Voltage vs. Temperature UVLO Threshold vs. Temperature Typical Performance Characteristics 30052804 30052803 Soft Start Current vs Temperature Enable Threshold vs. Temperature 30052805 30052806 Switching Frequency vs. Temperature PMOS RDS(ON) vs. Temperature 30052807 30052808 5 www.national.com LM2854 NMOS RDS(ON) vs. Temperature IQ (operating) vs. VIN and Temperature 30052809 30052810 Peak Current Limit vs. Temperature Feedback Voltage vs. VIN 30052811 30052812 IQ (disabled) vs. VIN and Temperature, EN = 0V Switching Frequency vs. VIN 30052813 30052814 www.national.com 6 LM2854 LM2854 1 MHz Efficiency vs. IOUT VOUT = 0.8V, LO = 0.47 µH, 3.3 mΩ DCR LM2854 1 MHz Efficiency vs. IOUT VOUT = 2.5V, LO = 1.0 µH, 7.1 mΩ DCR 30052815 30052816 LM2854 1 MHz Efficiency vs. IOUT VOUT = 1.2V, LO = 0.68 µH, 4.9 mΩ DCR LM2854 1 MHz Efficiency vs. IOUT VOUT = 3.3V, LO = 1.0 µH, 7.1 mΩ DCR 30052817 30052818 LM2854 1 MHz Efficiency vs. IOUT VOUT = 1.8V, LO = 1.0 µH, 7.1 mΩ DCR LM2854 500 kHz Efficiency vs. IOUT VOUT = 0.8V, LO = 1.0 µH, 7.1 mΩ DCR 30052819 30052820 7 www.national.com LM2854 LM2854 500 kHz Efficiency vs. IOUT VOUT = 2.5V, LO = 2.2 µH, 16 mΩ DCR LM2854 500 kHz Efficiency vs. IOUT VOUT = 1.2V, LO = 1.5 µH, 9.7 mΩ DCR 30052821 30052822 LM2854 500 kHz Efficiency vs. IOUT VOUT = 3.3V, LO = 1.5 µH, 9.7 mΩ DCR LM2854 500 kHz Efficiency vs. IOUT VOUT = 1.8V, LO = 1.5 µH, 9.7 mΩ DCR 30052823 30052824 LM2854 1 MHz Bode Plot VIN = 3.3V, VOUT = 1.8V, IOUT = 4A RFB1 = 150 kΩ, RCOMP = 1 kΩ, CCOMP = 100 pF, LOUT = 0.82 µH, COUT = 100 µF ceramic LM2854 500 kHz Bode Plot VIN = 3.3V, VOUT = 1.8V, IOUT = 4A RFB1 = 250 kΩ, RCOMP = 1 kΩ, CCOMP = 47 pF, LOUT = 1.5 µH, COUT = 100 µF ceramic 30052825 30052826 www.national.com 8 LM2854 LM2854 1 MHz Bode Plot VIN = 5.0V, VOUT = 1.8V, IOUT = 4A RFB1 = 150 kΩ, RCOMP = 1 kΩ, CCOMP = 100 pF, LOUT = 0.82 µH, COUT = 100 µF LM2854 500 kHz Bode Plot VIN = 5.0V, VOUT = 1.8V, IOUT = 4A RFB1 = 250 kΩ, RCOMP = 1 kΩ, CCOMP = 47 pF, LOUT = 1.5 µH, COUT = 100 µF ceramic 30052827 30052828 LM2854 1 MHz Bode Plot VIN = 5.0V, VOUT = 3.3V, IOUT = 4A RFB1 = 150 kΩ, RCOMP = 1 kΩ, CCOMP = 68 pF, LOUT = 0.82 µH, COUT = 100 µF ceramic LM2854 500 kHz Bode Plot VIN = 5.0V, VOUT = 3.3V, IOUT = 4A RFB1 = 250 kΩ, RCOMP = 1 kΩ, CCOMP = 33 pF, LOUT = 1.5 µH, COUT = 100 µF ceramic 30052829 30052830 LM2854 500 kHz Power On Characteristic VIN = 5.0V, VOUT = 1.8V, IOUT = 4A, CSS = 220 pF LM2854 500 kHz Power On via Enable VIN = 5.0V, VOUT = 1.8V, IOUT = 4A, CSS = 220 pF 30052831 30052832 9 www.national.com LM2854 LM2854 500 kHz Power Off Characteristic VIN = 5.0V, VOUT = 1.8V, IOUT = 4A, CSS = 220 pF LM2854 1 MHz Load Transient Response VIN = 5.0V, VOUT = 3.3V, IOUT = 0.5A to 4A to 0.5A step di/dt ≊ 4A/µs, CO = 100 µF ceramic 30052833 30052834 LM2854 500 kHz Switch Node Voltage (oscilloscope set at infinite persistence) VIN = 5.0V, VOUT = 2.5V, IOUT = 4A LM2854 500 kHz Startup Waveform VOUT = 2.5V, IOUT = 0A 30052835 30052856 LM2854 500 kHz Pre-Biased Startup Waveform (oscilloscope set at infinite persistence) VOUT = 2.5V, IOUT = 0A, VPRE-BIAS = 1.25V 30052855 www.national.com 10 LM2854 Block Diagram 30052836 Applications Information GENERAL The LM2854 PowerWise® synchronous DC-DC buck regulator belongs to the National Semiconductor SIMPLE SWITCHER® family of switching regulators. Integration of the power MOSFETs and associated drivers, compensation component network and the PWM controller reduces the number of external components necessary for a complete power supply design, without sacrificing performance. Operation Description SWITCHING FREQUENCY The LM2854 is available in two switching frequency options, 500 kHz and 1 MHz. Generally, a higher switching frequency allows for faster transient response and a reduction in the footprint area and volume of the external power stage components, while a lower switching frequency affords better efficiency. These factors should be considered when selecting the appropriate switching frequency for a given application. ENABLE The LM2854 features a enable (EN) pin and associated comparator to allow the user to easily sequence the LM2854 from an external voltage rail, or to manually set the input UVLO threshold. The turn-on or rising threshold and hysteresis for this comparator are typically 1.23V and 0.15V respectively. The precise reference for the enable comparator allows the user to guarantee that the LM2854 will be disabled when the system demands it to be. SOFT-START The LM2854 begins to operate when both the AVIN and EN voltages exceed the rising UVLO and enable thresholds, re- spectively. A controlled soft-start eliminates inrush currents during start-up and allows the user more control and flexibility when sequencing the LM2854 with other power supplies. An external soft-start capacitor is used to control the LM2854 start-up time. During soft-start, the voltage on the feedback pin is connected internally to the non-inverting input of the error amplifier. The soft-start period lasts until the voltage on the soft-start pin exceeds the LM2854 reference voltage of 0.8V. At this point, the reference voltage takes over at the noninverting amplifier input. In the event of either AVIN or EN decreasing below the falling UVLO or enable threshold respectively, the voltage on the soft-start pin is collapsed by discharging the soft-start capacitor through a 5 kΩ transistor to ground. TRACKING The LM2854 can track the output of a master power supply during soft-start by connecting a resistor divider to the SS pin. In this way, the output voltage slew rate of the LM2854 will be controlled by a master supply for loads that require precise sequencing. When the tracking function is used, a small value soft-start capacitor can be connected to the SS pin to alleviate output voltage overshoot when recovering from a current limit fault. 11 www.national.com LM2854 the negative current sensed in the low-side NFET is below approximately -0.4A, the present switching cycle is immediately terminated and both FETs are turned off. When both FETs are off, the negative inductor current originally flowing in the low-side NFET and into the SW pin commutates to the high-side PFET’s body diode and ramps back to zero. At this point, the SW pin becomes a high impedance node and ringing can be observed on the SW node as the stored energy in the inductor is dissipated while resonating with the parasitic nodal capacitance. OVER-TEMPERATURE PROTECTION When the LM2854 senses a junction temperature greater than 165°C, both switching FETs are turned off and the part enters a sleep state. Upon sensing a junction temperature below 155°C, the part will re-initiate the soft-start sequence and begin switching once again. This feature is provided to prevent catastrophic failure due to excessive thermal dissipation. LOOP COMPENSATION The LM2854 preserves flexibility by integrating the control components around the error amplifier while utilizing three small external compensation components from VOUT to FB. An integrated type II (two pole, one zero) voltage-mode compensation network is featured. To ensure stability, an external resistor and small value capacitor can be added across the upper feedback resistor as a pole-zero pair to complete a type III (three pole, two zero) compensation network. For correct selection of these components, see the design section of this datasheet. 30052857 PRE-BIASED STARTUP CAPABILITY The LM2854 is in a pre-biased state when the device starts up with an output voltage greater than zero. This often occurs in many multi-rail applications such as when powering an FPGA, ASIC, or DSP. The output can be pre-biased in these applications through parasitic conduction paths from one supply rail to another. Even though the LM2854 is a synchronous converter, it will not pull the output low when a pre-bias condition exists. The LM2854 will not sink current during start up until the soft-start voltage exceeds the voltage on the FB pin. Since the device can not sink current it protects the load from damage that might otherwise occur if current is conducted through the parasitic paths of the load. FEEDBACK VOLTAGE ACCURACY The FB pin is connected to the inverting input of the voltage loop error amplifier and during closed loop operation its reference voltage is 0.8V. The FB voltage is accurate to within -1.25% / +1.0% over temperature. Additionally, the LM2854 contains error nulling circuitry to substantially eliminate the feedback voltage over temperature drift as well as the long term aging effects of the internal amplifiers. In addition, the 1/ f noise of the bandgap amplifier and reference are dramatically reduced. The manifestation of this circuit action is that the duty cycle will have two slightly different but distinct operating points, each evident every other switching cycle. The oscilloscope plot shown previously of the SW pin with infinite persistence set shows this behavior. No discernible effect is evident on the output due to LC filter attenuation. For further information, a National Semiconductor white paper is available on this topic. POSITIVE CURRENT LIMIT The LM2854 employs lossless cycle-by-cycle high-side current limit circuitry to limit the peak current through the highside FET. The peak current limit threshold, denoted ICL, is nominally set at 6A internally. When a current greater than ICL is sensed through the PFET, its on-time is immediately terminated and the NFET is activated. The NFET stays on for the entire next four switching cycles (effectively four PFET pulses are skipped). During these skipped pulses, the voltage on the soft-start pin is reduced by discharging the soft-start capacitor by a current sink on the soft-start pin of nominally 6 µA or 14 µA for the 500 kHz or 1 MHz options, respectively. Subsequent over-current events will drain more and more charge from the soft-start capacitor, effectively decreasing the reference voltage as the output droops due to the pulse skipping. Reactivation of the soft-start circuitry ensures that when the over-current situation is removed, the part will resume normal operation smoothly. NEGATIVE CURRENT LIMIT The LM2854 implements negative current limit detection circuitry to prevent large negative current in the inductor. When www.national.com 12 Design Guidelines INPUT FILTER CAPACITOR Fast switching currents place a large strain on the input supply to a buck regulator. A capacitor placed close to the PVIN and PGND pins of the LM2854 helps to supply the instantaneous charge required when the regulator demands a pulse of current every switching cycle. In fact, the input capacitor conducts a square-wave current of peak-to-peak amplitude equal to IOUT. With this high AC current present in the input capacitor, the RMS current rating becomes an important parameter. The necessary RMS current rating of the input capacitor to a buck regulator can be estimated by where the PWM duty cycle, D, is given by Neglecting capacitor ESR, the resultant input capacitor AC ripple voltage is a triangular waveform with peak-to-peak amplitude specified as follows The maximum input capacitor ripple voltage and RMS current occur at 50% duty cycle. A 22 µF or 47 µF high quality dielectric (X5R, X7R) ceramic capacitor with adequate voltage rating is typically sufficient as an input capacitor to the LM2854 LM2854. The input capacitor should be placed as close as possible to the PVIN and PGND pins to substantially eliminate the parasitic effects of any stray inductance or resistance on the PC board and supply lines. Additional bulk capacitance with higher ESR may be required to damp any resonance effects of the input capacitance and parasitic inductance. AVIN FILTERING COMPONENTS In addition to the large input filter capacitor, a smaller ceramic capacitor such as a 0.1 µF or 1.0 µF is recommended between AVIN and AGND to filter high frequency noise present on the PVIN rail from the quiet AVIN supply. For additional filtering in noisy environments, a small RC filter can be used on the AVIN pin as shown below. The above equation includes an offset voltage to ensure that the final value of the SS pin voltage exceeds the reference voltage of the LM2854. This offset will cause the LM2854 output voltage to reach regulation slightly before the master supply. A value of 33 kΩ 1% is recommended for RT2 as a compromise between high precision and low quiescent current through the divider while minimizing the effect of the 2 µA soft-start current source. For example, If the master supply voltage VOUT1 is 3.3V and the LM2854 output voltage was 1.8V, then the value of RT1 needed to give the two supplies identical soft-start times would be 14.3 kΩ. A timing diagram for this example, the equal soft-start time case, is shown below. 30052840 In general, RF is typically selected between 1Ω and 10Ω so that the steady state voltage drop across the resistor due to the AVIN bias current does not affect the UVLO level. Recommended filter capacitor, CF, is 1.0 µF in X5R or X7R dielectric. SOFT-START CAPACITOR When the LM2854 is enabled, the output voltage will ramp up linearly in the time dictated by the following relationship 30052859 where VREF is the internal reference voltage (nominally 0.8V), ISS is the soft-start charging current (nominally 2 µA) and CSS is the external soft-start capacitance. Rearranging this equation allows for the necessary soft-start capacitor for a given startup time to be calculated as follows TRACKING - EQUAL SLEW RATES Alternatively, the tracking feature can be used to have similar output voltage ramp rates. This is referred to as simultaneous startup. In this case, the tracking resistors can be determined based on the following equation and to ensure proper overdrive of the SS pin VOUT2 < 0.8 VOUT1 Thus, the required soft start capacitor per unit output voltage startup time is given by CSS = 2.5 nF / ms For example, a 10 nF soft-start capacitor will yield a 4 ms softstart time. TRACKING - EQUAL SOFT-START TIME One way to use the tracking feature is to design the tracking resistor divider so that the master supply output voltage, VOUT1, and the LM2854 output voltage, VOUT2, both rise together and reach their target values at the same time. This is termed ratiometric startup. For this case, the equation governing the values of tracking divider resistors RT1 and RT2 is given by For the example case of VOUT1 = 5V and VOUT2 = 2.5V, with RT2 set to 33 kΩ as before, RT1 is calculated from the above equation to be 15.5 kΩ. A timing diagram for the case of equal slew rates is shown below. 13 www.national.com LM2854 RFB1 is defined based on the voltage loop requirements and RFB2 is then selected for the desired output voltage. These resistors are normally selected as 0.5% or 1% tolerance. COMPENSATION COMPONENT SELECTION The power stage transfer function of a voltage mode buck converter has a complex double pole related to the LC output filter and a left half plane zero due to the output capacitor ESR, denoted RESR. The locations of these singularities are given respectively by 30052861 ENABLE AND UVLO Using a resistor divider from VIN to EN as shown in the schematic diagram below, the input voltage at which the part begins switching can be increased above the normal input UVLO level according to where CO is the output capacitance value appropriately derated for applied voltage and operating temperature, RL is the effective load resistance and RDCR is the series damping resistance associated with the inductor and power switches. For example, suppose that the required input UVLO level is 3.69V. Choosing REN2 = 10 k Ω, then we calculate REN1 = 20 kΩ. 30052844 30052848 Alternatively, the EN pin can be driven from another voltage source to cater for system sequencing requirements commonly found in FPGA and other multi-rail applications. The following schematic shows an LM2854 that is sequenced to start based on the voltage level of a master system rail. The conventional compensation strategy employed with voltage mode control is to use two compensator zeros to offset the LC double pole, one compensator pole located to cancel the output capacitor ESR zero and one compensator pole located between one third and one half switching frequency for high frequency noise attenuation. The LM2854 internal compensation components are designed to locate a pole at the origin and a pole at high frequency as mentioned above. Furthermore, a zero is located at 8.8 kHz or 17.6 kHz for the 500 kHz or 1 MHz options, respectively, to approximately cancel the likely location of one LC filter pole. The three external compensation components, RFB1, RCOMP and CCOMP, are selected to position a zero at or below the LC pole location and a pole to cancel the ESR zero. The voltage loop crossover frequency, floop, is usually selected between one tenth to one fifth of the switching frequency 0.1fSW ≤ floop ≤ 0.2fSW A simple solution for the required external compensation capacitor, CCOMP, with type III voltage mode control can be expressed as 30052845 OUTPUT VOLTAGE SETTING A divider resistor network from VOUT to the FB pin determines the desired output voltage as follows www.national.com 14 LM2854 where the constant α is nominally 0.038 or 0.075 for the 500 kHz or 1 MHz options, respectively. This assumes a compensator pole cancels the output capacitor ESR zero. Furthermore, since the modulator gain is proportional to VIN, the loop crossover frequency increases with VIN. Thus, it is recommended to design the loop at maximum expected VIN. The upper feedback resistor, RFB1, is selected to provide adequate mid-band gain and to locate a zero at or below the LC pole frequency. The series resistor, RCOMP, is selected to locate a pole at the ESR zero frequency. Thus excellent load current transient performance and low output voltage ripple. The inductance is chosen such that the peak-to-peak inductor current ripple, ΔiL, is approximately 25 to 40% of IOUT as follows Note that the peak inductor current is the DC output current plus half the ripple current and reaches its highest level at lowest duty cycle (or highest VIN). It is recommended that the inductor should have a saturation current rating in excess of the current limit level. Table 3 lists examples of off-the-shelf powdered iron and ferrite based inductors that are suitable for use with the LM2854. The output capacitor can be of ceramic or electrolytic chemistry. The chosen output capacitor requires sufficient DC voltage rating and RMS ripple current handling capability. The output capacitor RMS current and peak-to-peak output ripple are given respectively by Note that the lower feedback resistor, RFB2, has no impact on the control loop from an AC standpoint since the FB pin is the input to an error amplifier and effectively at AC ground. Hence, the control loop can be designed irrespective of output voltage level. The only caveat here is the necessary derating of the output capacitance with applied voltage. Having chosen RFB1 as above, RFB2 is then selected for the desired output voltage. Table 1 and Table 2 list inductor and ranges of capacitor values that work well with the LM2854, along with the associated compensation components to ensure stable operation. Values different than those listed may be used, but the compensation components may need to be recalculated to avoid degradation in phase margin. Note that the capacitance ranges specified refer to in-circuit values where the nominal capacitance value is adequately derated for applied voltage. FILTER INDUCTOR AND OUTPUT CAPACITOR SELECTION In a buck regulator, selection of the filter inductor and capacitor will affect many key system parameters, including stability, transient response and efficiency The LM2854 can accommodate relatively wide ranges of output capacitor and filter inductor values in a typical application and still achieve In general, 22 µF to 100 µF of ceramic output capacitance is sufficient for both LM2854 frequency options given the optimal high frequency characteristics and low ESR of ceramic dielectric. It is advisable to consult the manufacturer’s derating curves for capacitance voltage coefficient as the in-circuit capacitance may drop significantly with applied voltage. Tantalum or organic polymer electrolytic capacitance may be suitable with the LM2854 500 kHz option, particularly in applications where substantial bulk capacitance per unit volume is required. However, the high loop bandwidth achievable with the LM2854 obviates the necessity for large bulk capacitance during transient loading conditions. Table 4 lists some examples of commercially available capacitors that can be used with the LM2854. TABLE 1. LM2854 500 kHz Compensation Component Values VIN (V) 5.0 LO (µH) Min 1.5 1.5 1.5 2.2 2.2 2.2 3.3 1.5 1.5 1.5 2.2 2.2 2.2 40 100 100 40 100 100 40 100 100 40 100 100 CO (µF) Max 100 200 220 100 200 220 100 200 220 100 200 220 2 1 15 2 1 15 2 1 15 2 1 15 15 ESR (mΩ) Min Max 10 5 25 10 5 25 10 5 25 10 5 25 RFB1 (kΩ) 150 150 150 150 150 120 150 100 100 150 100 100 CCOMP (pF) 47 100 120 68 120 120 68 150 150 100 220 220 RCOMP (kΩ) 1 1 25 1 1 15 1 1 15 1 1 10 www.national.com LM2854 TABLE 2. LM2854 1 MHz Compensation Component Values VIN (V) 5.0 LO (µH) Min 0.68 0.68 0.68 1.0 1.0 1.0 3.3 0.68 0.68 0.68 1.0 1.0 1.0 20 60 100 20 60 100 20 60 100 20 60 100 CO (µF) Max 60 150 220 60 150 220 60 150 220 60 150 220 2 1 15 2 1 15 2 1 15 2 1 15 ESR (mΩ) Min Max 10 5 25 10 5 25 10 5 25 10 5 25 120 75 100 100 75 75 75 50 50 75 50 33 33 100 100 56 150 150 56 150 150 82 220 330 1 1 20 1 1 15 1 1 12 1 1 10 RFB1 (kΩ) CCOMP (pF) RCOMP (kΩ) TABLE 3. Recommended Filter Inductors Inductance (µH) 0.47 1.0 0.47 0.47 0.68 0.82 1.0 1.5 2.2 0.47 0.68 1.0 1.5 2.2 0.56 0.47 0.68 1.0 1.2 1.5 1.8 0.47 0.68 1.0 1.5 0.47 1.0 1.5 DCR (mΩ) 14.5 24.0 8.4 6.0 8.7 10.6 13.1 18.5 15.7 3.5 4.5 17.3 10.4 16.1 10 3.3 4.8 7.5 9.4 11.5 16.5 3.3 4.9 7.1 9.7 14 9 14 Manufacturer Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Sumida Sumida Sumida Sumida Sumida Coilcraft Coilcraft Coilcraft Coilcraft Coilcraft Coilcraft Coilcraft TDK TDK TDK TDK Cyntec Cyntec Cyntec Manufacturer P/N IHLP1616BZERR47M11 IHLP1616BZER1R0M11 IHLP2525AHERR47M01 IHLP2525BDERR47M01 IHLP2525BDERR68M01 IHLP2525BDERR82M01 IHLP2525BDER1R0M01 IHLP2525BDER1R5M01 IHLP2525CZER2R2M11 CDMC6D28NP-R47M CDMC6D28NP-R68M CDMC6D28NP-1R0M CDMC6D28NP-1R5M CDMC6D28NP-2R2M DO1813H-561ML HA3619-471ALC HA3619-681ALC HA3619-102ALC HA3619-122ALC HA3619-152ALC HA3619-182ALC SPM6530T-R47M170 SPM6530T-R68M140 SPM6530T-1R0M120 SPM6530T-1R5M100 PCMC042T-0R47MN PCMC063T-1R0MN PCMC063T-1R5MN Case Size (mm) 4.06 x 4.45 x 2.00 4.06 x 4.45 x 2.00 6.47 x 6.86 x 1.80 6.47 x 6.86 x 2.40 6.47 x 6.86 x 2.40 6.47 x 6.86 x 2.40 6.47 x 6.86 x 2.40 6.47 x 6.86 x 2.40 6.47 x 6.86 x 3.00 6.50 x 7.25 x 3.00 6.50 x 7.25 x 3.00 6.50 x 7.25 x 3.00 6.50 x 7.25 x 3.00 6.50 x 7.25 x 3.00 6.10 x 8.89 x 5.00 7.0 x 7.0 x 3.0 7.0 x 7.0 x 3.0 7.0 x 7.0 x 3.0 7.0 x 7.0 x 3.0 7.0 x 7.0 x 3.0 7.0 x 7.0 x 3.0 7.1 x 6.5 x 3.0 7.1 x 6.5 x 3.0 7.1 x 6.5 x 3.0 7.1 x 6.5 x 3.0 4.0 x 4.5 x 2.0 6.5 x 6.9 x 3.0 6.5 x 6.9 x 3.0 www.national.com 16 LM2854 TABLE 4. Recommended Filter Capacitors Capacitance (µF) 22 47 47 47 100 100 100 150 330 470 Voltage (V), ESR (mΩ) 6.3, < 5 6.3, < 5 6.3, < 5 10.0, < 5 6.3, < 5 6.3, 50 6.3, 25 6.3, 18 6.3, 18 6.3, 23 Chemistry Ceramic, X5R Ceramic, X5R Ceramic, X5R Ceramic, X5R Ceramic, X5R Tantalum Organic Polymer Organic Polymer Organic Polymer Niobium Oxide Manufacturer TDK TDK TDK TDK TDK AVX Sanyo Sanyo Sanyo AVX Manufacturer P/N C3216X5R0J226M C3216X5R0J476M C3225X5R0J476M C3225X5R1A476M C3225X5R0J107M TPSD157M006#0050 6TPE100MPB2 6TPE150MIC2 6TPE330MIL NOME37M006#0023 Case Size 1206 1206 1210 1210 1210 D, 7.5 x 4.3 x 2.9 mm B2, 3.5 x 2.8 x 1.9 mm C2, 6.0 x 3.2 x 1.8 mm D3L, 7.3 x 4.3 x 2.8 mm E, 7.3 x 4.3 x 4.1 mm 17 www.national.com LM2854 PC Board Layout Guidelines PC board layout is an important part of DC-DC converter design. Poor board layout can disrupt the performance of a DCDC converter and surrounding circuitry by contributing to EMI, ground bounce and resistive voltage drop in the traces. These can send erroneous signals to the DC-DC converter resulting in poor regulation or instability. Good layout can be implemented by following a few simple design rules. 30052853 1. Minimize area of switched current loops. There are two loops where currents are switched at high di/ dt slew rates in a buck regulator. The first loop represents the path taken by AC current flowing during the high side PFET on time. This current flows from the input capacitor to the regulator PVIN pins, through the high side FET to the regulator SW pin, filter inductor, output capacitor and returning via the PCB ground plane to the input capacitor. The second loop represents the path taken by AC current flowing during the low side NFET on time. This current flows from the output capacitor ground to the regulator PGND pins, through the NFET to the inductor and output capacitor. From an EMI reduction standpoint, it is imperative to minimize this loop area during PC board layout by physically locating the input capacitor close to the LM2854. Specifically, it is advantageous to place CIN as close as possible to the LM2854 PVIN and PGND pins. Grounding for both the input and output capacitor should consist of a localized top side plane that connects to PGND and the exposed die attach pad (DAP). The inductor should be placed close to the SW pin and output capacitor. 2. Minimize the copper area of the switch node. The LM2854 has two SW pins optimally located on one side of the package. In general the SW pins should be connected to the filter inductor on the top PCB layer. The inductor should be placed close to the SW pins to minimize the copper area of the switch node. 3. Have a single point ground for all device analog grounds located under the DAP. The ground connections for the Feedback, Soft-start, Enable and AVIN components should be routed to the AGND pin of the device. The AGND pin should connect to PGND under the DAP. This prevents any switched or load currents from flowing in the analog ground traces. If not properly handled, poor grounding can result in degraded load regulation or erratic switching behavior. 4. Minimize trace length to the FB pin. Since the feedback (FB) node is high impedance, the trace from the output voltage setpoint resistor divider to FB pin should be as short as possible. This is most important as relatively high value resistors are used to set the output voltage. The FB trace should be routed away from the SW pin and inductor to avoid noise pickup from the SW pin. Both feedback resistors, RFB1 and RFB2, and the compensation components, RCOMP and CCOMP, should be located close to the FB pin. 5. Make input and output bus connections as wide as possible. This reduces any voltage drops on the input or output of the converter and maximizes efficiency. To optimize voltage accuracy at the load, ensure that a separate feedback voltage sense trace is made to the load. Doing so will correct for voltage drops and provide optimum output accuracy. 6. Provide adequate device heat-sinking. Use an array of heat-sinking vias to connect the DAP to the ground plane on the bottom PCB layer. If the PCB has a plurality of copper layers, these thermal vias can also be employed to make connection to inner layer heat-spreading ground planes. For best results use a 5 x 3 via array with minimum via diameter of 10 mils. Ensure enough copper area is used to keep the junction temperature below 125°C. www.national.com 18 LM2854 LM2854 Application Circuit Schematic and BOMs This section provides several application solutions with an associated bill of materials. All bill of materials reference the schematic below. The compensation for each solution was optimized to work over the full input range. Many applications have a fixed input voltage rail. It is possible to modify the compensation to obtain a faster transient response for a given input voltage operating point. 30052854 TABLE 5. LM2854 500kHz Bill of Materials, VIN = 5V, VOUT = 3.3V, IOUT(MAX) = 4A, Optimized for Efficiency Ref Des U1 CIN CO LO RFB1 RFB2 RCOMP RF CCOMP CSS CF Description Synchronous Buck Regulator 47 µF, X5R, 10V 100 µF, X5R, 6.3V 1.5 µH, 9.7 mΩ, 10A 249 kΩ 80.6 kΩ 1.0 kΩ 1.0Ω 33 pF, ±5%, C0G, 50V 10 nF, ±10%, X7R, 16V 1.0 µF, ±10%, X7R, 10V Case Size eTSSOP-16 1210 1210 7.1 x 6.5 x 3.0 mm 0603 0603 0603 0603 0603 0603 0603 Manufacturer National Semiconductor TDK TDK TDK Vishay Dale Vishay Dale Vishay Dale Vishay Dale TDK Murata Murata Manufacturer P/N LM2854MHX-500 C3225X5R1A476M C3225X5R0J107M SPM6530T-1R5M100 CRCW06032493F-e3 CRCW060328062F-e3 CRCW06031001F-e3 CRCW06031R0F-e3 C1608C0G1H330J GRM188R71C103KA01 GRM188R71A105KA61 19 www.national.com LM2854 TABLE 6. LM2854 1 MHz Bill of Materials, VIN = 3.3V to 5V, VOUT = 2.5V, IOUT (MAX) = 4A, Optimized for Electrolytic Input and Output Capacitance Ref Des U1 CIN CO LO RFB1 RFB2 RCOMP RF CCOMP CSS CF Description Synchronous Buck Regulator 150 µF, 6.3V, 18 mΩ 330 µF, 6.3V, 18 mΩ 2.2 µH, 16 mΩ, 7A 100 kΩ 47.5 kΩ 15 kΩ 1.0Ω 330 pF, ±5%, C0G, 50V 10 nF, ±10%, X7R, 16V 1.0 µF,±10%, X7R, 10V Case Size eTSSOP-16 C2, 6.0 x 3.2 x 1.8 mm D3L, 7.3 x 4.3 x 2.8 mm 6.47 x 6.86 x 3.00 mm 0603 0603 0603 0603 0603 0603 0603 Manufacturer National Semiconductor Sanyo Sanyo Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale TDK Murata Murata Manufacturer P/N LM2854MHX-1000 6TPE150MIC2 6TPE330MIL IHLP2525CZER2R2M11 CRCW06031003F-e3 CRCW060324752F-e3 CRCW06031502F-e3 CRCW06031R0F-e3 C1608C0G1H331J GRM188R71C103KA01 GRM188R71A105KA61 TABLE 7. LM2854 1 MHz Bill of Materials, VIN = 3.3V, VOUT = 0.8V, IOUT (MAX) = 4A, Optimized for Solution Size and Transient Response Ref Des U1 CIN CO LO RFB1 RCOMP RF CCOMP CSS CF Description Synchronous Buck Regulator 47 µF, X5R, 6.3V 47 µF, X5R, 6.3V 0.47 µH, 14.5 mΩ, 7A 110 kΩ 1.0 kΩ 1.0Ω 27 pF, ±5%, C0G, 50V 10 nF, ±10%, X7R, 16V 1.0 µF, ±10%, X7R, 10V Case Size eTSSOP-16 1206 1206 4.06 x 4.45 x 2.00 mm 0402 0402 0402 0402 0402 0402 Manufacturer National Semiconductor TDK TDK Vishay Dale Vishay Dale Vishay Dale Vishay Dale Murata Murata Murata Manufacturer P/N LM2854MHX-1000 C3216X5R0J476M C3216X5R0J476M IHLP1616BZER0R47M11 CRCW04021103F-e3 CRCW04021001F-e3 CRCW04021R0F-e3 GRM1555C1H270JZ01 GRM155R71C103KA01 GRM155R61A105KE15 www.national.com 20 LM2854 Physical Dimensions inches (millimeters) unless otherwise noted 16-Lead eTSSOP Package NS Package Number MXA16A 21 www.national.com LM2854 4A 500 kHz / 1 MHz PowerWise® Synchronous SIMPLE SWITCHER® Buck Regulator Notes For more National Semiconductor product information and proven design tools, visit the following Web sites at: Products Amplifiers Audio Clock Conditioners Data Converters Displays Ethernet Interface LVDS Power Management Switching Regulators LDOs LED Lighting PowerWise Serial Digital Interface (SDI) Temperature Sensors Wireless (PLL/VCO) www.national.com/amplifiers www.national.com/audio www.national.com/timing www.national.com/adc www.national.com/displays www.national.com/ethernet www.national.com/interface www.national.com/lvds www.national.com/power www.national.com/switchers www.national.com/ldo www.national.com/led www.national.com/powerwise www.national.com/sdi www.national.com/tempsensors www.national.com/wireless WEBENCH Analog University App Notes Distributors Green Compliance Packaging Design Support www.national.com/webench www.national.com/AU www.national.com/appnotes www.national.com/contacts www.national.com/quality/green www.national.com/packaging www.national.com/quality www.national.com/refdesigns www.national.com/feedback Quality and Reliability Reference Designs Feedback THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. 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