LM3310SQX

LM3310 Step-Up PWM DC/DC Converter with Integrated Op-Amp and Gate Pulse Modulation Switch November 26, 2007 LM3310 Step-Up PWM DC/DC Converter with Integrated Op-Amp and Gate Pulse Modulation Switch General Description The LM3310 is a step-up DC/DC converter integrated with an Operational Amplifier and a gate pulse modulation switch. The boost (step-up) converter is used to generate an adjustable output voltage and features a low RDSON internal switch for maximum efficiency. The operating frequency is selectable between 660kHz and 1.28MHz allowing for the use of small external components. An external soft-start pin enables the user to tailor the soft-start time to a specific application and limit the inrush current. The Op-Amp is capable of sourcing/sinking 135mA of current (typical). The gate pulse modulation switch can operate with a VGH voltage of 5V to 30V. The LM3310 is available in a low profile 24-lead LLP package. Features ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ Boost converter with a 2A, 0.18Ω switch Boost output voltage adjustable up to 20V Operating voltage range of 2.5V to 7V 660kHz/1.28MHz pin selectable switching frequency Adjustable soft-start function Input undervoltage protection Over temperature protection Integrated Op-Amp Integrated gate pulse modulation (GPM) switch 24-Lead LLP package Applications ■ TFT Bias Supplies ■ Portable Applications Typical Application Circuit 20133331 © 2007 National Semiconductor Corporation 201333 www.national.com LM3310 Connection Diagram 20133304 LLP-24 (Top View) θJA=37°C/W Ordering Information Order Number LM3310SQ LM3310SQX LM3310SQ LM3310SQX NOPB NOPB Spec. Package Type LLP-24 LLP-24 LLP-24 LLP-24 NSC Package Drawing SQA24A SQA24A SQA24A SQA24A Supplied As 1000 units/reel tape and reel 4500 units/reel tape and reel 1000 units/reel tape and reel 4500 units/reel tape and reel Pin Descriptions Pin 1 2 3 4 5 6 7 8 9 10 11 12 13 Name NC VGHM VFLK VDPM VDD AVIN OUT NEG POS AGND NC NC NC Not internally connected. Leave pin open. Output of GPM circuit. This output directly drives the supply for the gate driver circuits. Determines when the TFT LCD is on or off. This is controlled by the timing controller in the LCD module. VDPM pin is the enable signal for the GPM block. Pulling this pin high enables the GPM while pulling this pin low disables it. VDPM is used for timing sequence control. Reference input for gate pulse modulation (GPM) circuit. The voltage at VDD is used to set the lower VGHM voltage. If the GPM function is not used connect VDD to VIN. Op-Amp analog power input. Output of the Op-Amp. Negative input terminal of the Op-Amp. Positive input terminal of the Op-Amp. Analog ground for the step-up regulator, LDO, and Op-Amp. Connect directly to DAP and PGND beneath the device. Not internally connected. Leave pin open. Not internally connected. Leave pin open. Not internally connected. Leave pin open. Function www.national.com 2 LM3310 Pin 14 15 16 17 18 19 20 21 22 23 24 DAP Name SS VC FREQ VIN SW SHDN FB PGND CE RE VGH Boost converter soft start pin. Function Boost compensation network connection. Connected to the output of the voltage error amplifier. Switching frequency select input. Connect this pin to VIN for 1.28MHz operation and AGND for 660kHz operation. Boost converter and GPM power input. Boost power switch input. Switch connected between SW pin and PGND pin. Shutdown pin. Active low, pulling this pin low disable the LM3310. Boost output voltage feedback input. Power Ground. Source connection of the step-up regulator NMOS switch and ground for the GPM circuit. Connect AGND and PGND directly to the DAP beneath the device. Connect capacitor from this pin to AGND. Connect a resistor between RE and PGND. GPM power supply input. VGH range is 5V to 30V. Die Attach Pad. Internally connected to GND. Connect AGND and PGND pins directly to this pad beneath the device. Block Diagrams 20133357 3 www.national.com LM3310 20133358 20133359 www.national.com 4 LM3310 Absolute Maximum Ratings (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN SW Voltage FB Voltage VC Voltage (Note 2) SHDN Voltage FREQ AVIN Amplifier Inputs/Output VGH Voltage VGHM Voltage VFLK, VDPM, VDD Voltage CE Voltage (Note 2) RE Voltage 7.5V 21V VIN 1.265V ± 0.3V 7.5V VIN 14.5V Rail-to-Rail 31V VGH 7.5V 1.265 + 0.3V VGH Maximum Junction Temperature Power Dissipation(Note 3) Lead Temperature Vapor Phase (60 sec.) Infrared (15 sec.) ESD Susceptibility (Note 4) Human Body Model 150°C Internally Limited 300°C 215°C 220°C 2kV Operating Conditions Operating Junction Temperature Range (Note 5) Storage Temperature Supply Voltage Maximum SW Voltage VGH Voltage Range Op-Amp Supply, AVIN −40°C to +125°C −65°C to +150°C 2.5V to 7V 20V 5V to 30V 4V to 14V Electrical Characteristics Specifications in standard type face are for TJ = 25°C and those with boldface type apply over the full Operating Temperature Range ( TJ = −40°C to +125°C). Unless otherwise specified, VIN =2.5V and IL = 0A. Symbol IQ Parameter Quiescent Current Conditions FB = 2V (Not Switching) VSHDN = 0V 660kHz Switching 1.28MHz Switching VFB %VFB/ΔVIN ICL IB ISS VSS VIN gm AV DMAX fS ISHDN IL RDSON ThSHDN UVP IFREQ Feedback Voltage Feedback Voltage Line Regulation Switch Current Limit (Note 7) FB Pin Bias Current (Note 9) SS Pin Current SS Pin Voltage Input Voltage Range Error Amp Transconductance ΔI = 5µA Error Amp Voltage Gain Maximum Duty Cycle Switching Frequency Shutdown Pin Current Switch Leakage Current Switch RDSON SHDN Threshold Undervoltage Protection Threshold FREQ Pin Current fS = 660kHz fS = 1.28MHz FREQ = Ground FREQ = VIN VSHDN = 2.5V VSHDN = 0.3V VSW = 20V ISW = 500mA Output High, VIN = 2.5V to 7V Output Low, VIN = 2.5V to 7V On Threshold (Switch On) Off Threshold (Switch Off) FREQ = VIN = 2.5V 2.5 2.4 2.3 2.7 2.1 13.5 1.4 0.4 V µA 80 80 440 1.0 8.5 1.20 2.5 26 74 69 91 89 660 1.28 8 1 0.03 0.18 760 1.5 13.5 2 5 0.35 µA Ω V 2.5V ≤ VIN ≤ 7V (Note 8) 1.231 -0.26 2.0 Min(Note 5) Typ(Note 6) Max(Note 5) 690 0.04 2.1 3.1 1.263 0.089 2.6 27 11 1.24 160 13.5 1.28 7 133 1100 0.5 8.5 2.8 4.0 1.287 0.42 µA Units mA V %/V A nA µA V V µmho V/V % kHz MHz µA 5 www.national.com LM3310 Electrical Characteristics Specifications in standard type face are for TJ = 25°C and those with boldface type apply over the full Operating Temperature Range ( TJ = −40°C to +125°C). Unless otherwise specified VIN =2.5V and AVIN = 8V. Operational Amplifier Symbol VOS IB VOUT Swing AVIN Is+ IOUT Supply Voltage Supply Current Output Current Buffer, VO = AVIN/2, No Load Source Sink 90 105 Parameter Input Offset Voltage Input Bias Current (POS Pin) Conditions Buffer configuration, VO = AVIN/2, no load Buffer configuration, VO = AVIN/2, no load (Note 9) Buffer, RL=2kΩ, VO min. Buffer, RL=2kΩ, VO max. 7.9 4 1.5 138 135 Min(Note 5) Typ(Note 6) Max(Note 5) 5.7 200 0.001 7.97 14 7.8 195 175 15 550 0.03 Units mV nA V V mA mA Electrical Characteristics Specifications in standard type face are for TJ = 25°C and those with boldface type apply over the full Operating Temperature Range ( TJ = −40°C to +125°C). Unless otherwise specified VIN =2.5V. Gate Pulse Modulation Symbol VFLK VDPM VDD(TH) IVFLK IVDPM IVGH RVGH-VGHM RVGHM-RE RVGHM(OFF) ICE VCE(TH) Parameter VFLK Voltage Levels VDPM Voltage Levels VDD Threshold VFLK Current VDPM Current VGH Bias Current VGH to VGHM Resistance VGHM to RE Resistance VGH Resistance CE Current CE Voltage Threshold Conditions Rising edge threshold Falling edge threshold Rising edge threshold Falling edge threshold VGHM = 30V VGHM = 5V VFLK = 1.5V VFLK = 0.3V VDPM = 1.5V VDPM = 0.3V VGH = 30V, VFLK High VGH = 30V, VFLK Low 20mA Current, VGH = 30V 20mA Current, VGH = VGHM = 30V VDPM is Low, VGHM = 2V CE = 0V 7 1.16 0.4 2.8 0.4 3 0.5 4.8 1.1 4.8 1.1 59 11 14 27 1.2 11 1.22 3.3 0.7 11 2.5 11 2.5 300 35.5 28.5 55 1.7 16 1.34 Ω kΩ µA V 0.4 1.4 Min(Note 5) Typ(Note 6) Max(Note 5) 1.4 Units V V V µA µA µA Note 1: Absolute maximum ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions for which the device is intended to be functional, but device parameter specifications may not be guaranteed. For guaranteed specifications and test conditions, see the Electrical Characteristics. Note 2: Under normal operation the VC and CE pins may go to voltages above this value. The maximum rating is for the possibility of a voltage being applied to the pin, however the VC and CE pins should never have a voltage directly applied to them. Note 3: The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(MAX), the junction-to-ambient thermal resistance, θJA, and the ambient temperature, TA. See the Electrical Characteristics table for the thermal resistance of various layouts. The maximum allowable power dissipation at any ambient temperature is calculated using: PD (MAX) = (TJ(MAX) − TA)/θJA. Exceeding the maximum allowable power dissipation will cause excessive die temperature, and the regulator will go into thermal shutdown. Note 4: The human body model is a 100pF capacitor discharged through a 1.5kΩ resistor into each pin per JEDEC standard JESD22-A114. Note 5: All limits guaranteed at room temperature (standard typeface) and at temperature extremes (bold typeface). All room temperature limits are 100% production tested. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL). Note 6: Typical numbers are at 25°C and represent the most likely norm. Note 7: Duty cycle affects current limit due to ramp generator. Note 8: Current limit at 0% duty cycle. See TYPICAL PERFORMANCE section for Switch Current Limit vs. VIN www.national.com 6 LM3310 Note 9: Bias current flows into pin. Typical Performance Characteristics SHDN Pin Current vs. SHDN Pin Voltage SS Pin Current vs. Input Voltage 20133361 20133362 FREQ Pin Current vs. Input Voltage FB Pin Current vs. Temperature 20133363 20133364 CE Pin Current vs. Input Voltage VDPM Pin Current vs. VDPM Pin Voltage 20133360 20133365 7 www.national.com LM3310 VFLK Pin Current vs. VFLK Pin Voltage 660kHz Switching Quiescent Current vs. Input Voltage 20133366 20133387 1.28MHz Switching Quiescent Current vs. Input Voltage 660kHz Switching Quiescent Current vs. Temperature 20133367 20133388 1.28MHz Switching Quiescent Current vs. Temperature 660kHz Switching Frequency vs. Temperature 20133368 20133389 www.national.com 8 LM3310 1.28MHz Switching Frequency vs. Temperature Switch Current Limit vs. Input Voltage 20133369 20133319 Non-Switching Quiescent Current vs. Input Voltage GPM Disabled Non-Switching Quiescent Current vs. Input Voltage GPM Enabled 20133371 20133372 Non-Switching Quiescent Current vs. Temperature GPM Disabled Non-Switching Quiescent Current vs. Temperature GPM Enabled 20133373 20133374 9 www.national.com LM3310 Power NMOS RDSON vs. Input Voltage 660kHz Max. Duty Cycle vs. Input Voltage 20133375 20133391 1.28MHz Max. Duty Cycle vs. Input Voltage 660kHz Max. Duty Cycle vs. Temperature 20133383 20133390 1.28MHz Max. Duty Cycle vs. Temperature 1.28MHz Application Efficiency 20133376 20133382 www.national.com 10 LM3310 1.28MHz Application Efficiency VGH Pin Bias Current vs. VGH Pin Voltage 20133326 20133377 VGH Pin Bias Current vs. VGH Pin Voltage VGH-VGHM PMOS RDSON vs. VGH Pin Voltage 20133378 20133379 VGHM-RE PMOS RDSON vs. VGHM Pin Voltage VGHM OFF Resistance vs. Temperature 20133380 20133381 11 www.national.com LM3310 Op-Amp Source Current vs. AVIN Op-Amp Sink Current vs. AVIN 20133396 20133397 Op-Amp Quiescent Current vs. AVIN Op-Amp Offset Voltage vs. AVIN (No Load) 20133316 20133317 Op-Amp Offset Voltage vs. Load Current 1.28MHz, 8.5V Application Boost Load Step 20133351 VOUT = 8.5V, VIN = 3.3V, COUT = 20µF 1) VOUT, 200mV/div, AC 3) ILOAD, 200mA/div, DC T = 200µs/div 20133318 www.national.com 12 LM3310 1.28MHz, 8.5V Application Boost Startup Waveform 1.28MHz, 8.5V Application Boost Startup Waveform 20133352 20133353 VOUT = 8.5V, VIN = 3.3V, COUT = 20µF, RLOAD = 20Ω, CSS = 10nF 1) VSHDN, 2V/div, DC 2) VOUT, 5V/div, DC 3) IIN, 500mA/div, DC T = 200µs/div VOUT = 8.5V, VIN = 3.3V, COUT = 20µF, RLOAD = 20Ω, CSS = 100nF 1) VSHDN, 2V/div, DC 2) VOUT, 5V/div, DC 3) IIN, 500mA/div, DC T = 1ms/div 1.28MHz, 8.5V Application Boost Startup Waveform 20133354 VOUT = 8.5V, VIN = 3.3V, COUT = 20µF, RLOAD = 20Ω, CSS = open 1) VSHDN, 2V/div, DC 2) VOUT, 5V/div, DC 3) IIN, 1A/div, DC T = 40µs/div 13 www.national.com LM3310 Operation 20133302 FIGURE 1. Simplified Boost Converter Diagram (a) First Cycle of Operation (b) Second Cycle Of Operation CONTINUOUS CONDUCTION MODE The LM3310 contains a current-mode, PWM boost regulator. A boost regulator steps the input voltage up to a higher output voltage. In continuous conduction mode (when the inductor current never reaches zero at steady state), the boost regulator operates in two cycles. In the first cycle of operation, shown in Figure 1 (a), the transistor is closed and the diode is reverse biased. Energy is collected in the inductor and the load current is supplied by COUT. The second cycle is shown in Figure 1 (b). During this cycle, the transistor is open and the diode is forward biased. The energy stored in the inductor is transferred to the load and output capacitor. The ratio of these two cycles determines the output voltage. The output voltage is defined approximately as: SOFT-START CAPACITOR The LM3310 has a soft-start pin that can be used to limit the inductor inrush current on start-up. The external SS pin is used to tailor the soft-start for a specific application but is not required for all applications and can be left open when not needed. When used a current source charges the external soft-start capacitor CSS until it reaches its typical clamp voltage, VSS. The soft-start time can be estimated as: TSS = CSS*VSS/ISS THERMAL SHUTDOWN The LM3310 includes thermal shutdown. If the die temperature reaches 145°C the device will shut down until it cools to a safe temperature at which point the device will resume operation. If the adverse condition that is heating the device is not removed (ambient temperature too high, short circuit conditions, etc...) the device will continue to cycle on and off to keep the die temperature below 145°C. The thermal shutdown has approximately 20°C of hysteresis. When in thermal shutdown the boost regulator, Op-Amp, and GPM blocks will all be disabled. INPUT UNDER-VOLTAGE PROTECTION The LM3310 includes input under-voltage protection (UVP). The purpose of the UVP is to protect the device both during start-up and during normal operation from trying to operate with insufficient input voltage. During start-up using a ramping input voltage the UVP circuitry ensures that the device does not begin switching until the input voltage reaches the UVP On threshold. If the input voltage is present and the shutdown 14 where D is the duty cycle of the switch, D and D′ will be required for design calculations. SETTING THE OUTPUT VOLTAGE (BOOST CONVERTER) The output voltage is set using the feedback pin and a resistor divider connected to the output as shown in the typical operating circuit. The feedback pin voltage is 1.263V, so the ratio of the feedback resistors sets the output voltage according to the following equation: www.national.com LM3310 pin is pulled high the UVP circuitry will prevent the device from switching if the input voltage present is lower than the UVP On threshold. During normal operation the UVP circuitry will disable the device if the input voltage falls below the UVP Off threshold for any reason. In this case the device will not turn back on until the UVP On threshold voltage is exceeded. OPERATIONAL AMPLIFIER Compensation: The architecture used for the amplifier in the LM3310 requires external compensation on the output. Depending on the equivalent resistive and capacitive distributed load of the TFT-LCD panel, external components at the amplifier outputs may or may not be necessary. If the capacitance presented by the load is equal to or greater than an equivalent distibutive load of 50Ω in series with 4.7nF no external components are needed as the TFT-LCD panel will act as compensation itself. Distributed resistive and capacitive loads enhance stability and increase performance of the amplifiers. If the capacitance and resistance presented by the load is less than 50Ω in series with 4.7nF, external components will be required as the load itself will not ensure stability. No external compensation in this case will lead to oscillation of the amplifier and an increase in power consumption. A good choice for compensation in this case is to add a 50Ω in series with a 4.7nF capacitor from the output of the amplifier to ground. This allows for driving zero to infinite capacitance loads with no oscillations, minimal overshoot, and a higher slew rate than using a single large capacitor. The high phase margin created by the external compensation will guarantee stability and good performance for all conditions. Layout and Filtering considerations: When the power supply for the amplifiers (AVIN) is connected to the output of the switching regulator, the output ripple of the regulator will produce ripple at the output of the amplifiers. This can be minimized by directly bypassing the AVIN pin to ground with a low ESR ceramic capacitor. For best noise reduction a resistor on the order of 5Ω to 20Ω from the supply being used to the AVIN pin will create and RC filter and give you a cleaner supply to the amplifier. The bypass capacitor should be placed as close to the AVIN pin as possible and connected directly to the AGND plane. For best noise immunity all bias and feedback resistors should be in the low kΩ range due to the high input impedance of the amplifier. It is good practice to use a small capacitance at the high impedance input terminals as well to reduce noise susceptibility. All resistors and capacitors should be placed as close to the input pins as possible. Special care should also be taken in routing of the PCB traces. All traces should be as short and direct as possible. The output pin trace must never be routed near any trace going to the positive input. If this happens cross talk from the output trace to the positive input trace will cause the circuit to oscillate. The op-amp is not a three terminal device it has 5 terminals: positive voltage power pin, AGND, positive input, negative input, and the output. The op-amp "routes" current from the power input pin and AGND to the output pin. So in effect an opamp has not two inputs but four, all of which must be kept noise free relative to the external circuits which are being driven by the op-amp. The current from the power pins goes through the output pin and into the load and feedback loop. The current exiting the load and feedback loops then must have a return path back to the op-amp power supply pins. Ideally this return path must follow the same path as the output pin trace to the load. Any deviation that makes the loop area larger between the output current path and the return current path adds to the probability of noise pick up. GATE PULSE MODULATION The Gate Pulse Modulation (GPM) block is designed to provide a modulated voltage to the gate driver circuitry of a TFT LCD display. Operation is best understood by referring to the GPM block diagram in the Block Diagrams section, the drawing in Figure 2 and the transient waveforms in Figure 3 and Figure 4. There are two control signals in the GPM block, VDPM and VFLK. VDPM is the enable pin for the GPM block. If VDPM is high, the GPM block is active and will respond to the VFLK drive signal from the timing controller. However, if VDPM is low, the GPM block will be disabled and both PMOS switches P2 and P3 will be turned off. The VGHM node will be discharged through a 1kΩ resistor and the NMOS switch N2. When VDPM is high, typical waveforms for the GPM block can be seen in Figure 2. The pin VGH is typically driven by a 2x or 3x charge pump. In most cases, the 2x or 3x charge pump is a discrete solution driven from the SW pin and the output of the boost switching regulator. When VFLK is high, the PMOS switch P2 is turned on and the PMOS switch P3 is turned off. With P2 on, the VGHM pin is pulled to the same voltage applied to the VGH pin. This provides a high gate drive voltage, VGHMMAX, and can source current to the gate drive circuitry. When VFLK is high, NMOS switch N3 is on which discharges the capacitor CE. 20133384 FIGURE 2. When VFLK is low, the NMOS switch N3 is turned off which allows current to charge the CE capacitor. This creates a delay, tDELAY, given by the following equations: tDELAY ≊ 1.265V(CE + 7pF)/ICE When the voltage on CE reaches about 1.265V and the VFLK signal is low, the PMOS switch P2 will turn off and the PMOS switch P3 will turn on connecting resistor R3 to the VGHM pin through P3. This will discharge the voltage at VGHM at some rate determined by R3 creating a slope, MR, as shown in Figure 2. The VGHM pin is no longer a current source, it is now sinking current from the gate drive circuitry. As VGHM is discharged through R3, the comparator connected to the pin VDD monitors the VGHM voltage. PMOS switch P3 will turn off when the following is true: VGHMMIN ≊ 10VXR2/(R1 + R2) 15 www.national.com LM3310 where VX is some voltage connected to the resistor divider on pin VDD. VX is typically connected to the output of the boost switching regulator. When PMOS switch P3 turns off, VGHM will be high impedance until the VFLK pin is high again. Figure 3 and Figure 4 give typical transient waveforms for the GPM block. Waveform (1) is the VGHM pin, (2) is the VFLK and (3) is the VDPM. The output of the boost switching regulator is operating at 8.5V and there is a 3x discrete charge pump (~23.5V) supplying the VGH pin. In Figure 3 and Figure 4, the VGHM pin is driving a purely capacitive load, 4.7nF. The value of resistor R1 is 15kohm, R2 is 1.1kΩ and R3 is 750Ω. In both transient plots, there is no CE delay capacitor. abled and the VGHM pin will discharge through NMOS switch N2 and the 1kΩ resistor. This applies also if the junction temperature of the device exceeds 145°C or if the SHDN signal is low. As shown in the block diagram, both VDPM and VFLK have internal 350kΩ pull down resistors. This puts both VDPM and VFLK in normally “off” states. Typical VDPM and VFLK pin currents can be found in the Typical Performance Characteristics section. INTRODUCTION TO COMPENSATION (BOOST CONVERTER) 20133305 20133385 FIGURE 5. (a) Inductor current. (b) Diode current. The LM3310 is a current mode PWM boost converter. The signal flow of this control scheme has two feedback loops, one that senses switch current and one that senses output voltage. To keep a current programmed control converter stable above duty cycles of 50%, the inductor must meet certain criteria. The inductor, along with input and output voltage, will determine the slope of the current through the inductor (see Figure 5 (a)). If the slope of the inductor current is too great, the circuit will be unstable above duty cycles of 50%. A 10µH inductor is recommended for most 660 kHz applications, while a 4.7µH inductor may be used for most 1.28 MHz applications. If the duty cycle is approaching the maximum of 85%, it may be necessary to increase the inductance by as much as 2X. See Inductor and Diode Selection for more detailed inductor sizing. The LM3310 provides a compensation pin (VC) to customize the voltage loop feedback. It is recommended that a series combination of RC and CC be used for the compensation network, as shown in the typical application circuit. For any given application, there exists a unique combination of RC and CC that will optimize the performance of the LM3310 circuit in terms of its transient response. The series combination of RC and CC introduces a pole-zero pair according to the following equations: FIGURE 3. 20133386 FIGURE 4. In the GPM block diagram, a signal called “Reset” is shown. This signal is generated from the VIN under-voltage lockout, thermal shutdown, or the SHDN pin. If the VIN supply voltage drops below 2.3V, typically, then the GPM block will be dis- www.national.com 16 LM3310 where RO is the output impedance of the error amplifier, approximately 900k Ω. For most applications, performance can be optimized by choosing values within the range 5kΩ ≤ RC ≤ 100kΩ (RC can be up to 200kΩ if CC2 is used, see High Output Capacitor ESR Compensation) and 68pF ≤ CC ≤ 4.7nF. Refer to the Applications Information section for recommended values for specific circuits and conditions. Refer to the Compensation section for other design requirement. COMPENSATION This section will present a general design procedure to help insure a stable and operational circuit. The designs in this datasheet are optimized for particular requirements. If different conversions are required, some of the components may need to be changed to ensure stability. Below is a set of general guidelines in designing a stable circuit for continuous conduction operation, in most all cases this will provide for stability during discontinuous operation as well. The power components and their effects will be determined first, then the compensation components will be chosen to produce stability. INDUCTOR AND DIODE SELECTION Although the inductor sizes mentioned earlier are fine for most applications, a more exact value can be calculated. To ensure stability at duty cycles above 50%, the inductor must have some minimum value determined by the minimum input voltage and the maximum output voltage. This equation is: also be sized accordingly. It should have a saturation current rating higher than the peak inductor current expected. The output voltage ripple is also affected by the total ripple current. The output diode for a boost regulator must be chosen correctly depending on the output voltage and the output current. The typical current waveform for the diode in continuous conduction mode is shown in Figure 5 (b). The diode must be rated for a reverse voltage equal to or greater than the output voltage used. The average current rating must be greater than the maximum load current expected, and the peak current rating must be greater than the peak inductor current. During short circuit testing, or if short circuit conditions are possible in the application, the diode current rating must exceed the switch current limit. Using Schottky diodes with lower forward voltage drop will decrease power dissipation and increase efficiency. DC GAIN AND OPEN-LOOP GAIN Since the control stage of the converter forms a complete feedback loop with the power components, it forms a closedloop system that must be stabilized to avoid positive feedback and instability. A value for open-loop DC gain will be required, from which you can calculate, or place, poles and zeros to determine the crossover frequency and the phase margin. A high phase margin (greater than 45°) is desired for the best stability and transient response. For the purpose of stabilizing the LM3310, choosing a crossover point well below where the right half plane zero is located will ensure sufficient phase margin. To ensure a bandwidth of ½ or less of the frequency of the RHP zero, calculate the open-loop DC gain, ADC. After this value is known, you can calculate the crossover visually by placing a −20dB/decade slope at each pole, and a +20dB/ decade slope for each zero. The point at which the gain plot crosses unity gain, or 0dB, is the crossover frequency. If the crossover frequency is less than ½ the RHP zero, the phase margin should be high enough for stability. The phase margin can also be improved by adding CC2 as discussed later in this section. The equation for ADC is given below with additional equations required for the calculation: where fs is the switching frequency, D is the duty cycle, and RDSON is the ON resistance of the internal switch taken from the graph "RDSON vs. VIN" in the Typical Performance Characteristics section. This equation is only good for duty cycles greater than 50% (D>0.5), for duty cycles less than 50% the recommended values may be used. The value given by this equation is the inductance necessary to supress sub-harmonic oscillations. In some cases the value given by this equation may be too small for a given application. In this case the average inductor current and the inductor current ripple must be considered. The corresponding inductor current ripple, average inductor current, and peak inductor current as shown in Figure 5 (a) is given by: mc ≊ 0.072fs (in V/s) Continuous conduction mode occurs when ΔiL is less than the average inductor current and discontinuous conduction mode occurs when ΔiL is greater than the average inductor current. Care must be taken to make sure that the switch will not reach its current limit during normal operation. The inductor must 17 where RL is the minimum load resistance, VIN is the minimum input voltage, gm is the error amplifier transconductance found in the Electrical Characteristics table, and RDSON is the value chosen from the graph "NMOS RDSON vs. Input Voltage" in the Typical Performance Characteristics section. www.national.com LM3310 INPUT AND OUTPUT CAPACITOR SELECTION The switching action of a boost regulator causes a triangular voltage waveform at the input. A capacitor is required to reduce the input ripple and noise for proper operation of the regulator. The size used is dependant on the application and board layout. If the regulator will be loaded uniformly, with very little load changes, and at lower current outputs, the input capacitor size can often be reduced. The size can also be reduced if the input of the regulator is very close to the source output. The size will generally need to be larger for applications where the regulator is supplying nearly the maximum rated output or if large load steps are expected. A minimum value of 10µF should be used for the less stressful condtions while a 22µF to 47µF capacitor may be required for higher power and dynamic loads. Larger values and/or lower ESR may be needed if the application requires very low ripple on the input source voltage. The choice of output capacitors is also somewhat arbitrary and depends on the design requirements for output voltage ripple. It is recommended that low ESR (Equivalent Series Resistance, denoted RESR) capacitors be used such as ceramic, polymer electrolytic, or low ESR tantalum. Higher ESR capacitors may be used but will require more compensation which will be explained later on in the section. The ESR is also important because it determines the peak to peak output voltage ripple according to the approximate equation: ΔVOUT ≊ 2ΔiLRESR (in Volts) A minimum value of 10µF is recommended and may be increased to a larger value. After choosing the output capacitor you can determine a pole-zero pair introduced into the control loop by the following equations: SELECTING THE COMPENSATION COMPONENTS The first step in selecting the compensation components RC and CC is to set a dominant low frequency pole in the control loop. Simply choose values for RC and CC within the ranges given in the Introduction to Compensation section to set this pole in the area of 10Hz to 500Hz. The frequency of the pole created is determined by the equation: where RO is the output impedance of the error amplifier, approximately 900kΩ. Since RC is generally much less than RO, it does not have much effect on the above equation and can be neglected until a value is chosen to set the zero fZC. fZC is created to cancel out the pole created by the output capacitor, fP1. The output capacitor pole will shift with different load currents as shown by the equation, so setting the zero is not exact. Determine the range of fP1 over the expected loads and then set the zero fZC to a point approximately in the middle. The frequency of this zero is determined by: Now RC can be chosen with the selected value for CC. Check to make sure that the pole fPC is still in the 10Hz to 500Hz range, change each value slightly if needed to ensure both component values are in the recommended range. HIGH OUTPUT CAPACITOR ESR COMPENSATION When using an output capacitor with a high ESR value, or just to improve the overall phase margin of the control loop, another pole may be introduced to cancel the zero created by the ESR. This is accomplished by adding another capacitor, CC2, directly from the compensation pin VC to ground, in parallel with the series combination of RC and CC. The pole should be placed at the same frequency as fZ1, the ESR zero. The equation for this pole follows: Where RL is the minimum load resistance corresponding to the maximum load current. The zero created by the ESR of the output capacitor is generally very high frequency if the ESR is small. If low ESR capacitors are used it can be neglected. If higher ESR capacitors are used see the High Output Capacitor ESR Compensation section. Some suitable capacitor vendors include Vishay, Taiyo-Yuden, and TDK. RIGHT HALF PLANE ZERO A current mode control boost regulator has an inherent right half plane zero (RHP zero). This zero has the effect of a zero in the gain plot, causing an imposed +20dB/decade on the rolloff, but has the effect of a pole in the phase, subtracting another 90° in the phase plot. This can cause undesirable effects if the control loop is influenced by this zero. To ensure the RHP zero does not cause instability issues, the control loop should be designed to have a bandwidth of less than ½ the frequency of the RHP zero. This zero occurs at a frequency of: To ensure this equation is valid, and that CC2 can be used without negatively impacting the effects of RC and CC, fPC2 must be greater than 10fZC. CHECKING THE DESIGN With all the poles and zeros calculated the crossover frequency can be checked as described in the section DC Gain and Open-loop Gain. The compensation values can be changed a little more to optimize performance if desired. This is best done in the lab on a bench, checking the load step response with different values until the ringing and overshoot on the output voltage at the edge of the load steps is minimal. This should produce a stable, high performance circuit. For improved transient response, higher values of RC should be chosen. This will improve the overall bandwidth which makes the regulator respond more quickly to transients. If more detail is required, or the most optimum performance is desired, refer to a more in depth discussion of compensating current mode DC/DC switching regulators. where ILOAD is the maximum load current. www.national.com 18 LM3310 POWER DISSIPATION The output power of the LM3310 is limited by its maximum power dissipation. The maximum power dissipation is determined by the formula PD = (Tjmax - TA)/θJA where Tjmax is the maximum specified junction temperature (125°C), TA is the ambient temperature, and θJA is the thermal resistance of the package. LAYOUT CONSIDERATIONS The input bypass capacitor CIN, as shown in the typical operating circuit, must be placed close to the IC. This will reduce copper trace resistance which effects input voltage ripple of the IC. For additional input voltage filtering, a 100nF bypass capacitor can be placed in parallel with CIN, close to the VIN pin, to shunt any high frequency noise to ground. The output capacitor, COUT, should also be placed close to the IC. Any copper trace connections for the COUT capacitor can increase the series resistance, which directly effects output voltage ripple. The feedback network, resistors RFB1 and RFB2, should be kept close to the FB pin, and away from the inductor, to minimize copper trace connections that can inject noise into the system. RE and CE should also be close to the RE and CE pins to minimize noise in the GPM circuitry. Trace connections made to the inductor and schottky diode should be minimized to reduce power dissipation and increase overall efficiency. For more detail on switching power supply layout considerations see Application Note AN-1149: Layout Guidelines for Switching Power Supplies. For Op-Amp layout please refer to the Operational Amplifier section. Figure 6, Figure 7, and Figure 8 in the Application Information section following show the schematic and an example of a good layout as used in the LM3310/11 evaluation board. Application Information 20133323 FIGURE 6. Evaluation Board Schematic 19 www.national.com LM3310 20133324 FIGURE 7. Evaluation Board Layout (top layer) www.national.com 20 LM3310 20133325 FIGURE 8. Evaluation Board Layout (bottom layer) 21 www.national.com LM3310 20133329 FIGURE 9. Li-Ion to 8V, 1.28MHz Application www.national.com 22 LM3310 20133330 FIGURE 10. 5V to 10.5V, 1.28MHz Application Some Recommended Inductors (Others May Be Used) Manufacturer Coilcraft TDK Pulse Sumida Inductor DO3316 and DT3316 series SLF10145 series P0751 and P0762 series CDRH8D28 and CDRH8D43 series Some Recommended Input And Output Capacitors (Others May Be Used) Manufacturer Vishay Sprague Taiyo Yuden Capacitor 293D, 592D, and 595D series tantalum High capacitance MLCC ceramic ESRD seriec Polymer Aluminum Electrolytic SPV and AFK series V-chip series High capacitance MLCC ceramic EEJ-L series tantalum Contact Information www.vishay.com 407-324-4140 www.t-yuden.com 408-573-4150 www.cde.com www.panasonic.com Contact Information www.coilcraft.com 800-3222645 www.component.tdk.com 847-803-6100 www.pulseeng.com www.sumida.com Cornell Dubilier Panasonic 23 www.national.com LM3310 Physical Dimensions inches (millimeters) unless otherwise noted LLP-24 Pin Package (SQA) For Ordering, Refer to Ordering Information Table NS Package Number SQA24A www.national.com 24 LM3310 Notes 25 www.national.com LM3310 Step-Up PWM DC/DC Converter with Integrated Op-Amp and Gate Pulse Modulation Switch Notes For more National Semiconductor product information and proven design tools, visit the following Web sites at: Products Amplifiers Audio Clock Conditioners Data Converters Displays Ethernet Interface LVDS Power Management Switching Regulators LDOs LED Lighting PowerWise Serial Digital Interface (SDI) Temperature Sensors Wireless (PLL/VCO) www.national.com/amplifiers www.national.com/audio www.national.com/timing www.national.com/adc www.national.com/displays www.national.com/ethernet www.national.com/interface www.national.com/lvds www.national.com/power www.national.com/switchers www.national.com/ldo www.national.com/led www.national.com/powerwise www.national.com/sdi www.national.com/tempsensors www.national.com/wireless WEBENCH Analog University App Notes Distributors Green Compliance Packaging Design Support www.national.com/webench www.national.com/AU www.national.com/appnotes www.national.com/contacts www.national.com/quality/green www.national.com/packaging www.national.com/quality www.national.com/refdesigns www.national.com/feedback Quality and Reliability Reference Designs Feedback THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. NATIONAL MAKES NO REPRESENTATIONS OR WARRANTIES WITH RESPECT TO THE ACCURACY OR COMPLETENESS OF THE CONTENTS OF THIS PUBLICATION AND RESERVES THE RIGHT TO MAKE CHANGES TO SPECIFICATIONS AND PRODUCT DESCRIPTIONS AT ANY TIME WITHOUT NOTICE. NO LICENSE, WHETHER EXPRESS, IMPLIED, ARISING BY ESTOPPEL OR OTHERWISE, TO ANY INTELLECTUAL PROPERTY RIGHTS IS GRANTED BY THIS DOCUMENT. TESTING AND OTHER QUALITY CONTROLS ARE USED TO THE EXTENT NATIONAL DEEMS NECESSARY TO SUPPORT NATIONAL’S PRODUCT WARRANTY. EXCEPT WHERE MANDATED BY GOVERNMENT REQUIREMENTS, TESTING OF ALL PARAMETERS OF EACH PRODUCT IS NOT NECESSARILY PERFORMED. NATIONAL ASSUMES NO LIABILITY FOR APPLICATIONS ASSISTANCE OR BUYER PRODUCT DESIGN. BUYERS ARE RESPONSIBLE FOR THEIR PRODUCTS AND APPLICATIONS USING NATIONAL COMPONENTS. PRIOR TO USING OR DISTRIBUTING ANY PRODUCTS THAT INCLUDE NATIONAL COMPONENTS, BUYERS SHOULD PROVIDE ADEQUATE DESIGN, TESTING AND OPERATING SAFEGUARDS. EXCEPT AS PROVIDED IN NATIONAL’S TERMS AND CONDITIONS OF SALE FOR SUCH PRODUCTS, NATIONAL ASSUMES NO LIABILITY WHATSOEVER, AND NATIONAL DISCLAIMS ANY EXPRESS OR IMPLIED WARRANTY RELATING TO THE SALE AND/OR USE OF NATIONAL PRODUCTS INCLUDING LIABILITY OR WARRANTIES RELATING TO FITNESS FOR A PARTICULAR PURPOSE, MERCHANTABILITY, OR INFRINGEMENT OF ANY PATENT, COPYRIGHT OR OTHER INTELLECTUAL PROPERTY RIGHT. LIFE SUPPORT POLICY NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS PRIOR WRITTEN APPROVAL OF THE CHIEF EXECUTIVE OFFICER AND GENERAL COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein: Life support devices or systems are devices which (a) are intended for surgical implant into the body, or (b) support or sustain life and whose failure to perform when properly used in accordance with instructions for use provided in the labeling can be reasonably expected to result in a significant injury to the user. A critical component is any component in a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system or to affect its safety or effectiveness. National Semiconductor and the National Semiconductor logo are registered trademarks of National Semiconductor Corporation. All other brand or product names may be trademarks or registered trademarks of their respective holders. Copyright© 2007 National Semiconductor Corporation For the most current product information visit us at www.national.com National Semiconductor Americas Customer Support Center Email: new.feedback@nsc.com Tel: 1-800-272-9959 National Semiconductor Europe Customer Support Center Fax: +49 (0) 180-530-85-86 Email: europe.support@nsc.com Deutsch Tel: +49 (0) 69 9508 6208 English Tel: +49 (0) 870 24 0 2171 Français Tel: +33 (0) 1 41 91 8790 National Semiconductor Asia Pacific Customer Support Center Email: ap.support@nsc.com National Semiconductor Japan Customer Support Center Fax: 81-3-5639-7507 Email: jpn.feedback@nsc.com Tel: 81-3-5639-7560 www.national.com