LM3402

LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs June 2007 LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs General Description The LM3402/02HV are monolithic switching regulators designed to deliver constant currents to high power LEDs. Ideal for automotive, industrial, and general lighting applications, they contain a high-side N-channel MOSFET switch with a current limit of 735 mA (typical) for step-down (Buck) regulators. Hysteretic control with controlled on-time coupled with an external resistor allow the converter output voltage to adjust as needed to deliver a constant current to series and series - parallel connected arrays of LEDs of varying number and type, LED dimming by pulse width modulation (PWM), broken/open LED protection, low-power shutdown and thermal shutdown complete the feature set. Features ■ ■ ■ ■ ■ ■ ■ ■ Integrated 0.5A N-channel MOSFET VIN Range from 6V to 42V (LM3402) VIN Range from 6V to 75V (LM3402HV) 500 mA Output Current Over Temperature Cycle-by-Cycle Current Limit No Control Loop Compensation Required Separate PWM Dimming and Low Power Shutdown Supports all-ceramic output capacitors and capacitor-less outputs ■ Thermal shutdown protection ■ MSOP-8, PSOP-8 Packages Applications ■ ■ ■ ■ ■ LED Driver Constant Current Source Automotive Lighting General Illumination Industrial Lighting Typical Application 20192101 © 2007 National Semiconductor Corporation 201921 www.national.com LM3402/LM3402HV Connection Diagrams 20192145 20192102 8-Lead Plastic PSOP-8 Package NS Package Number MRA08B 8-Lead Plastic MSOP-8 Package NS Package Number MUA08A Ordering Information Order Number LM3402MM LM3402MMX LM3402HVMM LM3402HVMMX LM3402MR LM3402MRX LM3402HVMR LM3402HVMRX PSOP-8 MRA08B MSOP-8 MUA08A Package Type NSC Package Drawing Supplied As 1000 units on tape and reel 3500 units on tape and reel 1000 units on tape and reel 3500 units on tape and reel 95 units in anti-static rails 2500 units on tape and reel 95 units in anti-static rails 2500 units on tape and reel Pin Descriptions Pin(s) 1 2 3 4 5 6 7 8 DAP Name SW BOOT DIM GND CS RON VCC VIN GND Description Switch pin MOSFET drive bootstrap pin Input for PWM dimming Ground pin Current sense feedback pin On-time control pin Output of the internal 7V linear regulator Input voltage pin Thermal Pad Application Information Connect this pin to the output inductor and Schottky diode. Connect a 10 nF ceramic capacitor from this pin to SW. Connect a logic-level PWM signal to this pin to enable/disable the power FET and reduce the average light output of the LED array. Connect this pin to system ground. Set the current through the LED array by connecting a resistor from this pin to ground. A resistor connected from this pin to VIN sets the regulator controlled on-time. Bypass this pin to ground with a minimum 0.1 µF ceramic capacitor with X5R or X7R dielectric. Nominal operating input range is 6V to 42V (LM3402) or 6V to 75V (LM3402HV). Connect to ground. Place 4 to 6 vias from DAP to bottom layer ground plane. www.national.com 2 LM3402/LM3402HV Absolute Maximum Ratings (LM3402) (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN to GND BOOT to GND SW to GND BOOT to VCC BOOT to SW VCC to GND DIM to GND CS to GND RON to GND Junction Temperature Storage Temp. Range ESD Rating (Note 2) Soldering Information Lead Temperature (Soldering, 10sec) Infrared/Convection Reflow (15sec) -0.3V to 45V -0.3V to 59V -1.5V -0.3V to 45V -0.3V to 14V -0.3V to 14V -0.3V to 7V -0.3V to 7V -0.3V to 7V 150°C -65°C to 125°C 2kV 260°C 235°C Operating Ratings (LM3402) (Note 1) VIN Junction Temperature Range 6V to 42V −40°C to +125°C 200°C/W 50°C/W Thermal Resistance θJA (MSOP-8 Package) (Note 3) Thermal Resistance θJA (PSOP-8 Package) (Note 5) 3 www.national.com LM3402/LM3402HV Absolute Maximum Ratings (LM3402HV) (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN to GND BOOT to GND SW to GND BOOT to VCC BOOT to SW VCC to GND DIM to GND CS to GND RON to GND Junction Temperature -0.3V to 76V -0.3V to 90V -1.5V -0.3V to 76V -0.3V to 14V -0.3V to 14V -0.3V to 7V -0.3V to 7V -0.3V to 7V 150°C Storage Temp. Range ESD Rating (Note 2) Soldering Information Lead Temperature (Soldering, 10sec) Infrared/Convection Reflow (15sec) -65°C to 125°C 2kV 260°C 235°C Operating Ratings (LM3402HV) (Note 1) VIN Junction Temperature Range 6V to 75V −40°C to +125°C 200°C/W 50°C/W Thermal Resistance θJA (MSOP-8 Package) (Note 3) Thermal Resistance θJA (PSOP-8 Package) (Note 5) www.national.com 4 LM3402/LM3402HV Electrical Characteristics LM3402 Symbol SYSTEM PARAMETERS tON-1 tON-2 On-time 1 On-time 2 Parameter VIN = 24V unless otherwise indicated. Typicals and limits appearing in plain type apply for TA = TJ = +25°C. (Note 4) Limits appearing in boldface type apply over full Operating Temperature Range. Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis. Conditions VIN = 10V, RON = 200 kΩ VIN = 40V, RON = 200 kΩ Min 2.1 490 Typ 2.75 650 Max 3.4 810 Units µs ns LM3402HV Symbol SYSTEM PARAMETERS tON-1 tON-2 On-time 1 On-time 2 VIN = 10V, RON = 200 kΩ VIN = 70V, RON = 200 kΩ 2.1 290 2.75 380 3.4 470 µs ns Parameter Conditions Min Typ Max Units LM3402/LM3402HV Symbol VREF-REG VREF-0V ICS SHUTDOWN VSD-TH VSD-HYS OFF TIMER tOFF-MIN VCC-REG VIN-DO VCC-BP-TH VCC-BP-HYS VCC-Z-6 VCC-Z-8 VCC-Z-24 VCC-LIM VCC-UV-TH VCC-UV-HYS VCC-UV-DLY IIN-OP IIN-SD CURRENT LIMIT ILIM VIH VIL Current Limit Threshold Logic High Logic Low DIM Increasing DIM Decreasing 530 2.2 0.8 735 940 mA V V DIM COMPARATOR VCC Current Limit (Note 3) VCC Under-voltage Lock-out Threshold VCC Under-voltage Lock-out Hysteresis VCC Under-voltage Lock-out Filter Delay IIN Operating Current IIN Shutdown Current Minimum Off-time VCC Regulated Output VIN - VCC Dropout VCC Bypass Threshold VCC Bypass Hysteresis VCC Output Impedance (0 mA < ICC < 5 mA) ICC = 5 mA, 6.0V < VIN < 8.0V VIN Increasing VIN Decreasing VIN = 6V VIN = 8V VIN = 24V VIN = 24V, VCC = 0V VCC Increasing VCC Decreasing 100 mV Overdrive Non-switching, CS = 0V RON / SD = 0V CS = 0V 6.6 300 7 300 8.8 225 55 50 0.4 16 5.25 150 3 600 90 900 180 mA V mV µs µA µA 7.4 ns V mV V mV Ω INTERNAL REGULATOR Shutdown Threshold Shutdown Hysteresis RON / SD Increasing RON / SD Decreasing 0.3 0.7 40 1.05 V mV Parameter CS Regulation Threshold CS Over-voltage Threshold CS Bias Current Conditions CS Decreasing, SW turns on CS Increasing, SW turns off CS = 0V Min 194 Typ 200 300 0.1 Max 206 Units mV mV µA REGULATION AND OVER-VOLTAGE COMPARATORS 5 www.national.com LM3402/LM3402HV Symbol IDIM-PU RDS-ON VDR-UVLO VDR-HYS N-MOSFET AND DRIVER Parameter DIM Pull-up Current Buck Switch On Resistance DIM = 1.5V Conditions Min Typ 75 0.7 Max Units µA ISW = 200mA, BOOT-SW = 6.3V 1.7 1.5 4 Ω V mV BOOT Under-voltage Lock-out BOOT–SW Increasing Threshold BOOT Under-voltage Lock-out BOOT–SW Decreasing Hysteresis Thermal Shutdown Threshold Thermal Shutdown Hysteresis Junction to Ambient MSOP-8 Package PSOP-8 Package 3 400 THERMAL SHUTDOWN TSD TSD-HYS THERMAL RESISTANCE 165 25 200 50 °C °C °C/W θJA Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see Electrical Characteristics. Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Note 3: VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading. Note 4: Typical specifications represent the most likely parametric norm at 25°C operation. Note 5: θJA of 50°C/W with DAP soldered to a minimum of 2 square inches of 1 oz. copper on the top or bottom PCB layer. www.national.com 6 LM3402/LM3402HV Typical Performance Characteristics VREF vs Temperature (VIN = 24V) VREF vs VIN, LM3402 (TA = 25°C) 20192129 20192130 VREF vs VIN, LM3402HV (TA = 25°C) Current Limit vs Temperature (VIN = 24V) 20192131 20192132 Current Limit vs VIN, LM3402 (TA = 25°C) Current Limit vs VIN, LM3402HV (TA = 25°C) 20192133 20192134 7 www.national.com LM3402/LM3402HV TON vs VIN, RON = 100 kΩ (TA = 25°C) TON vs VIN, (TA = 25°C) 20192135 20192136 TON vs VIN, (TA = 25°C) TON vs RON, LM3402 (TA = 25°C) 20192137 20192144 TON vs RON, LM3402HV (TA = 25°C) VCC vs VIN (TA = 25°C) 20192138 20192139 www.national.com 8 LM3402/LM3402HV VO-MAX vs fSW, LM3402 (TA = 25°C) VO-MIN vs fSW, LM3402 (TA = 25°C) 20192140 20192141 VO-MAX vs fSW, LM3402HV (TA = 25°C) VO-MIN vs fSW, LM3402HV (TA = 25°C) 20192142 20192143 9 www.national.com LM3402/LM3402HV Block Diagram 20192103 Application Information THEORY OF OPERATION The LM3402 and LM3402HV are buck regulators with a wide input voltage range, low voltage reference, and a fast output enable/disable function. These features combine to make them ideal for use as a constant current source for LEDs with forward currents as high as 500 mA. The controlled on-time (COT) architecture is a combination of hysteretic mode control and a one-shot on-timer that varies inversely with input voltage. Hysteretic operation eliminates the need for smallsignal control loop compensation. When the converter runs in continuous conduction mode (CCM) the controlled on-time maintains a constant switching frequency over the range of input voltage. Fast transient response, PWM dimming, a low power shutdown mode, and simple output overvoltage protection round out the functions of the LM3402/02HV. CONTROLLED ON-TIME OVERVIEW Figure 1 shows the feedback system used to control the current through an array of LEDs. A voltage signal, VSNS, is created as the LED current flows through the current setting resistor, RSNS, to ground. VSNS is fed back to the CS pin, where it is compared against a 200 mV reference, VREF. The on-comparator turns on the power MOSFET when VSNS falls below VREF. The power MOSFET conducts for a controlled on-time, tON, set by an external resistor, RON, and by the input voltage, VIN. On-time is governed by the following equation: At the conclusion of tON the power MOSFET turns off for a minimum off-time, tOFF-MIN, of 300 ns. Once tOFF-MIN is complete the CS comparator compares VSNS and VREF again, waiting to begin the next cycle. www.national.com 10 LM3402/LM3402HV 20192105 FIGURE 1. Comparator and One-Shot The LM3402/02HV regulators should be operated in continuous conduction mode (CCM), where inductor current stays positive throughout the switching cycle. During steady-state operationin the CCM, the converter maintains a constant switching frequency, which can be selected using the following equation: The maximum number of LEDs, nMAX, that can be placed in a single series string is governed by VO(MAX) and the maximum forward voltage of the LEDs used, VF(MAX), using the expression: VF = forward voltage of each LED, n = number of LEDs in series AVERAGE LED CURRENT ACCURACY The COT architecture regulates the valley of ΔVSNS, the AC portion of VSNS. To determine the average LED current (which is also the average inductor current) the valley inductor current is calculated using the following expression: At low switching frequency the maximum duty cycle and output voltage are higher, allowing the LM3402/02HV to regulate output voltages that are nearly equal to input voltage. The following equation relates switching frequency to maximum output voltage. In this equation tSNS represents the propagation delay of the CS comparator, and is approximately 220 ns. The average inductor/LED current is equal to IL-MIN plus one-half of the inductor current ripple, ΔiL: IF = IL = IL-MIN + ΔiL / 2 Detailed information for the calculation of ΔiL is given in the Design Considerations section. MAXIMUM OUTPUT VOLTAGE The 300 ns minimum off-time limits on the maximum duty cycle of the converter, DMAX, and in turn ,the maximum output voltage VO(MAX) is determined by the following equations: 11 MINIMUM OUTPUT VOLTAGE The minimum recommended on-time for the LM3402/02HV is 300 ns. This lower limit for tON determines the minimum duty cycle and output voltage that can be regulated based on input voltage and switching frequency. The relationship is determined by the following equation: www.national.com LM3402/LM3402HV HIGH VOLTAGE BIAS REGULATOR The LM3402/02HV contains an internal linear regulator with a 7V output, connected between the VIN and the VCC pins. The VCC pin should be bypassed to the GND pin with a 0.1 µF ceramic capacitor connected as close as possible to the pins of the IC. VCC tracks VIN until VIN reaches 8.8V (typical) and then regulates at 7V as VIN increases. Operation begins when VCC crosses 5.25V. INTERNAL MOSFET AND DRIVER The LM3402/02HV features an internal power MOSFET as well as a floating driver connected from the SW pin to the BOOT pin. Both rise time and fall time are 20 ns each (typical) and the approximate gate charge is 3 nC. The high-side rail for the driver circuitry uses a bootstrap circuit consisting of an internal high-voltage diode and an external 10 nF capacitor, CB. VCC charges CB through the internal diode while the power MOSFET is off. When the MOSFET turns on, the internal diode reverse biases. This creates a floating supply equal to the VCC voltage minus the diode drop to drive the MOSFET when its source voltage is equal to VIN. FAST SHUTDOWN FOR PWM DIMMING The DIM pin of the LM3402/02HV is a TTL logic compatible input for low frequency PWM dimming of the LED. A logic low (below 0.8V) at DIM will disable the internal MOSFET and shut off the current flow to the LED array. While the DIM pin is in a logic low state the support circuitry (driver, bandgap, VCC) remains active in order to minimize the time needed to turn the LED array back on when the DIM pin sees a logic high (above 2.2V). A 75 µA (typical) pull-up current ensures that the LM3402/02HV is on when DIM pin is open circuited, eliminating the need for a pull-up resistor. Dimming frequency, fDIM, and duty cycle, DDIM, are limited by the LED current rise time and fall time and the delay from activation of the DIM pin to the response of the internal power MOSFET. In general, fDIM should be at least one order of magnitude lower than the steady state switching frequency in order to prevent aliasing. PEAK CURRENT LIMIT The current limit comparator of the LM3402/02HV will engage whenever the power MOSFET current (equal to the inductor current while the MOSFET is on) exceeds 735 mA (typical). The power MOSFET is disabled for a cool-down time that is 10x the steady-state on-time. At the conclusion of this cooldown time the system re-starts. If the current limit condition persists the cycle of cool-down time and restarting will continue, creating a low-power hiccup mode, minimizing thermal stress on the LM3402/02HV and the external circuit components. OVER-VOLTAGE/OVER-CURRENT COMPARATOR The CS pin includes an output over-voltage/over-current comparator that will disable the power MOSFET whenever VSNS exceeds 300 mV. This threshold provides a hard limit for the output current. Output current overshoot is limited to 300 mV / RSNS by this comparator during transients. The OVP/OCP comparator can also be used to prevent the output voltage from rising to VO(MAX) in the event of an output open-circuit. This is the most common failure mode for LEDs, due to breaking of the bond wires. In a current regulator an output open circuit causes VSNS to fall to zero, commanding maximum duty cycle. Figure 2 shows a method using a zener diode, Z1, and zener limiting resistor, RZ, to limit output voltage to the reverse breakdown voltage of Z1 plus 200 mV. The zener diode reverse breakdown voltage, VZ, must be greater than the maximum combined VF of all LEDs in the array. The maximum recommended value for RZ is 1 kΩ. As discussed in the Maximum Output Voltage section, there is a limit to how high VO can rise during an output open-circuit that is always less than VIN. If no output capacitor is used, the output stage of the LM3402/02HV is capable of withstanding VO(MAX) indefinitely, however the voltage at the output end of the inductor will oscillate and can go above VIN or below 0V. A small (typically 10 nF) capacitor across the LED array dampens this oscillation. For circuits that use an output capacitor, the system can still withstand VO(MAX) indefinitely as long as CO is rated to handle VIN. The high current paths are blocked in output open-circuit and the risk of thermal stress is minimal, hence the user may opt to allow the output voltage to rise in the case of an open-circuit LED failure. 20192112 FIGURE 2. Output Open Circuit Protection www.national.com 12 LM3402/LM3402HV LOW POWER SHUTDOWN The LM3402/02HV can be switched to a low power state (IINSD = 90 µA) by grounding the RON pin with a signal-level MOSFET as shown in Figure 3. Low power MOSFETs like the 2N7000, 2N3904, or equivalent are recommended devices for putting the LM3402/02HV into low power shutdown. Logic gates can also be used to shut down the LM3402/02HV as long as the logic low voltage is below the over temperature minimum threshold of 0.3V. Noise filter circuitry on the RON pin can cause a few pulses with a longer on-time than normal after RON is grounded or released. In these cases the OVP/ OCP comparator will ensure that the peak inductor or LED current does not exceed 300 mV / RSNS. 20192113 FIGURE 3. Low Power Shutdown THERMAL SHUTDOWN Internal thermal shutdown circuitry is provided to protect the IC in the event that the maximum junction temperature is exceeded. The threshold for thermal shutdown is 165°C with a 25°C hysteresis (both values typical). During thermal shutdown the MOSFET and driver are disabled. 13 www.national.com LM3402/LM3402HV Design Considerations SWITCHING FREQUENCY Switching frequency is selected based on the tradeoffs between efficiency (better at low frequency), solution size/cost (smaller at high frequency), and the range of output voltage that can be regulated (wider at lower frequency.) Many applications place limits on switching frequency due to EMI sensitivity. The on-time of the LM3402/02HV can be programmed for switching frequencies ranging from the 10’s of kHz to over 1 MHz. The maximum switching frequency is limited only by the minimum on-time requirement. LED RIPPLE CURRENT Selection of the ripple current, ΔiF, through the LED array is analogous to the selection of output ripple voltage in a standard voltage regulator. Where the output ripple in a voltage regulator is commonly ±1% to ±5% of the DC output voltage, LED manufacturers generally recommend values for ΔiF ranging from ±5% to ±20% of IF. Higher LED ripple current allows the use of smaller inductors, smaller output capacitors, or no output capacitors at all. The advantages of higher ripple current are reduction in the solution size and cost. Lower ripple current requires more output inductance, higher switching frequency, or additional output capacitance. The advantages of lower ripple current are a reduction in heating in the LED itself and greater range of the average LED current before the current limit of the LED or the driving circuitry is reached. BUCK CONVERTERS WITHOUT OUTPUT CAPACITORS The buck converter is unique among non-isolated topologies because of the direct connection of the inductor to the load during the entire switching cycle. By definition an inductor will control the rate of change of current that flows through it, and this control over current ripple forms the basis for component selection in both voltage regulators and current regulators. A current regulator such as the LED driver for which the LM3402/02HV was designed focuses on the control of the current through the load, not the voltage across it. A constant current regulator is free of load current transients, and has no need of output capacitance to supply the load and maintain output voltage. Referring to the Typical Application circuit on the front page of this datasheet, the inductor and LED can form a single series chain, sharing the same current. When no output capacitor is used, the same equations that govern inductor ripple current, ΔiL, also apply to the LED ripple current, ΔiF. For a controlled on-time converter such as LM3402/02HV the ripple current is described by the following expression: ered, making the magnetics smaller and less expensive. Alternatively, the circuit could be run at lower frequency but keep the same inductor value, improving the efficiency and expanding the range of output voltage that can be regulated. Both the peak current limit and the OVP/OCP comparator still monitor peak inductor current, placing a limit on how large ΔiL can be even if ΔiF is made very small. A parallel output capacitor is also useful in applications where the inductor or input voltage tolerance is poor. Adding a capacitor that reduces ΔiF to well below the target provides headroom for changes in inductance or VIN that might otherwise push the peak LED ripple current too high. Figure 4 shows the equivalent impedances presented to the inductor current ripple when an output capacitor, CO, and its equivalent series resistance (ESR) are placed in parallel with the LED array. The entire inductor ripple current flows through RSNS to provide the required 25 mV of ripple voltage for proper operation of the CS comparator. 20192115 FIGURE 4. LED and CO Ripple Current To calculate the respective ripple currents the LED array is represented as a dynamic resistance, rD. LED dynamic resistance is not always specified on the manufacturer’s datasheet, but it can be calculated as the inverse slope of the LED’s VF vs. IF curve. Note that dividing VF by IF will give an incorrect value that is 5x to 10x too high. Total dynamic resistance for a string of n LEDs connected in series can be calculated as the rD of one device multiplied by n. Inductor ripple current is still calculated with the expression from Buck Regulators without Output Capacitors. The following equations can then be used to estimate ΔiF when using a parallel capacitor: A minimum ripple voltage of 25 mV is recommended at the CS pin to provide good signal-to-noise ratio (SNR). The CS pin ripple voltage, ΔVSNS, is described by the following: ΔVSNS = ΔiF x RSNS BUCK CONVERTERS WITH OUTPUT CAPACITORS A capacitor placed in parallel with the LED or array of LEDs can be used to reduce the LED current ripple while keeping the same average current through both the inductor and the LED array. This technique is demonstrated in Design Example 1. With this topology the output inductance can be lowwww.national.com 14 The calculation for ZC assumes that the shape of the inductor ripple current is approximately sinusoidal. Small values of CO that do not significantly reduce ΔiF can also be used to control EMI generated by the switching action of the LM3402/02HV. EMI reduction becomes more important as the length of the connections between the LED and the rest of the circuit increase. LM3402/LM3402HV INPUT CAPACITORS Input capacitors at the VIN pin of the LM3402/02HV are selected using requirements for minimum capacitance and rms ripple current. The input capacitors supply pulses of current approximately equal to IF while the power MOSFET is on, and are charged up by the input voltage while the power MOSFET is off. Switching converters such as the LM3402/02HV have a negative input impedance due to the decrease in input current as input voltage increases. This inverse proportionality of input current to input voltage can cause oscillations (sometimes called ‘power supply interaction’) if the magnitude of the negative input impedance is greater the the input filter impedance. Minimum capacitance can be selected by comparing the input impedance to the converter’s negative resistance; however this requires accurate calculation of the input voltage source inductance and resistance, quantities which can be difficult to determine. An alternative method to select the minimum input capacitance, CIN(MIN), is to select the maximum voltage ripple which can be tolerated. This value,ΔvIN (MAX), is equal to the change in voltage across C IN during the converter on-time, when CIN supplies the load current. CIN (MIN) can be selected with the following: RECIRCULATING DIODE The LM3402/02HV is a non-synchronous buck regulator that requires a recirculating diode D1 (see the Typical Application circuit) to carrying the inductor current during the MOSFET off-time. The most efficient choice for D1 is a Schottky diode due to low forward drop and near-zero reverse recovery time. D1 must be rated to handle the maximum input voltage plus any switching node ringing when the MOSFET is on. In practice all switching converters have some ringing at the switching node due to the diode parasitic capacitance and the lead inductance. D1 must also be rated to handle the average current, ID, calculated as: ID = (1 – D) x IF This calculation should be done at the maximum expected input voltage. The overall converter efficiency becomes more dependent on the selection of D1 at low duty cycles, where the recirculating diode carries the load current for an increasing percentage of the time. This power dissipation can be calculated by checking the typical diode forward voltage, VD, from the I-V curve on the product datasheet and then multiplying it by ID. Diode datasheets will also provide a typical junction-to-ambient thermal resistance, θJA, which can be used to estimate the operating die temperature of the Schottky. Multiplying the power dissipation (PD = ID x VD) by θJA gives the temperature rise. The diode case size can then be selected to maintain the Schottky diode temperature below the operational maximum. A good starting point for selection of CIN is to use an input voltage ripple of 5% to 10% of VIN. A minimum input capacitance of 2x the CIN(MIN) value is recommended for all LM3402/02HV circuits. To determine the rms current rating, the following formula can be used: Design Example 1: LM3402 The first example circuit will guide the user through component selection for an architectural accent lighting application. A regulated DC voltage input of 24V ±10% will power a single 1W white LED at a forward current of 350 mA ±5%. The typical forward voltage of a 1W InGaN LED is 3.5V, hence the estimated average output voltage will be 3.7V. The objective of this application is to place the complete current regulator and LED in the compact space formerly occupied by an MR16 halogen light bulb. (The LED will be on a separate metal-core PCB.) Switching frequency will be as fast as the 300 ns tON limit allows, with the emphasis on space savings over efficiency. Efficiency cannot be ignored, however, as the confined space with little air-flow requires a maximum temperature rise of 40°C in each circuit component. A complete bill of materials can be found in Table 1 at the end of this datasheet. Ceramic capacitors are the best choice for the input to the LM3402/02HV due to their high ripple current rating, low ESR, low cost, and small size compared to other types. When selecting a ceramic capacitor, special attention must be paid to the operating conditions of the application. Ceramic capacitors can lose one-half or more of their capacitance at their rated DC voltage bias and also lose capacitance with extremes in temperature. A DC voltage rating equal to twice the expected maximum input voltage is recommended. In addition, the minimum quality dielectric which is suitable for switching power supply inputs is X5R, while X7R or better is preferred. 15 www.national.com LM3402/LM3402HV 20192119 FIGURE 5. Schematic for Design Example 1 RON and tON To select RON the expression relating tON to input voltage from the Controlled On-time Overview section can be re-written as: Minimum on-time occurs at the maximum VIN, which is 24V x 110% = 26.4V. RON is therefore calculated as: RON = (300 x 10-9 x 26.4) / 1.34 x 10-10 = 59105 Ω The closest 1% tolerance resistor is 59.0 kΩ. The switching frequency of the circuit can then be found using the equation relating RON to fSW: fSW = 3.7 / (59000 x 1.34 x 10-10) = 468 kHz USING AN OUTPUT CAPACITOR The inductor will be the largest component used in this design. Because the application does not require any PWM dimming, an output capacitor can be used to greatly reduce the inductance needed without worry of slowing the potential PWM dimming frequency. The total solution size will be reduced by using an output capacitor and small inductor as opposed to one large inductor. The closest standard inductor value is 33 µH. Off-the-shelf inductors rated at 33 µH are available from many magnetics manufacturers. Inductor datasheets should contain three specifications which are used to select the inductor. The first of these is the average current rating, which for a buck regulator is equal to the average load current, or IF. The average current rating is given by a specified temperature rise in the inductor, normally 40° C. For this example, the average current rating should be greater than 350 mA to ensure that heat from the inductor does not reduce the lifetime of the LED or cause the LM3402 to enter thermal shutdown. The second specification is the tolerance of the inductance itself, typically ±10% to ±30% of the rated inductance. In this example an inductor with a tolerance of ±20% will be used. With this tolerance the typical, minimum, and maximum inductor current ripples can be calculated: ΔiL(TYP) = [(26.4 – 3.7) x 300 x 10-9] / 33 x 10-6 = 206 mAP-P ΔiL(MIN) = [(26.4 – 3.7) x 300 x 10-9] / 39.6 x 10-6 = 172 mAP-P ΔiL(MAX) = [(26.4 – 3.7) x 300 x 10-9] / 26.4 x 10-6 = 258 mAP-P The third specification for an inductor is the peak current rating, normally given as the point at which the inductance drops off by a given percentage due to saturation of the core. The worst-case peak current occurs at maximum input voltage and at minimum inductance, and can be determined with the equation from the Design Considerations section: OUTPUT INDUCTOR Knowing that an output capacitor will be used, the inductor can be selected for a larger current ripple. The desired maximum value for ΔiL is ±30%, or 0.6 x 350 mA = 210 mAP-P. Minimum inductance is selected at the maximum input voltage. Re-arranging the equation for current ripple selection yields the following: LMIN = [(26.4 – 3.7) x 300 x 10-9] / (0.6 x 0.35) = 32.4 µH www.national.com 16 IL(PEAK) = 0.35 + 0.258 / 2 = 479 mA LM3402/LM3402HV For this example the peak current rating of the inductor should be greater than 479 mA. In the case of a short circuit across the LED array, the LM3402 will continue to deliver rated current through the short but will reduce the output voltage to equal the CS pin voltage of 200 mV. Worst-case peak current in this condition is equal to: ΔiL(LED-SHORT) = [(26.4 – 0.2) x 300 x 10-9] / 26.4 x 10-6 = 298 mAP-P IL(PEAK) = 0.35 + 0.149 = 499 mA In the case of a short at the switch node, the output, or from the CS pin to ground the short circuit current limit will engage at a typical peak current of 735 mA. In order to prevent inductor saturation during these short circuits the inductor’s peak current rating must be above 735 mA. The device selected is an off-the-shelf inductor rated 33 µH ±20% with a DCR of 96 mΩ and a peak current rating of 0.82A. The physical dimensions of this inductor are 7.0 x 7.0 x 4.5 mm. RSNS The current sensing resistor value can be determined by rearranging the expression for average LED current from the LED Current Accuracy section: CO = 1/(2 x π x 0.157 x 4.68 x 105) = 2.18 µF This calculation assumes that impedance due to the equivalent series resistance (ESR) and equivalent series inductance (ESL) of CO is negligible. The closest 10% tolerance capacitor value is 2.2 µF. The capacitor used should be rated to 10V or more and have an X7R dielectric. Several manufacturers produce ceramic capacitors with these specifications in the 0805 case size. A typical value for ESR of 1 mΩ can be read from the curve of impedance vs. frequency in the product datasheet. INPUT CAPACITOR Following the calculations from the Input Capacitor section, ΔvIN(MAX) will be 1%P-P = 240 mV. The minimum required capacitance is: CIN(MIN) = (0.35 x 300 x 10-9) / 0.24 = 438 nF In expectation that more capacitance will be needed to prevent power supply interaction a 1.0 µF ceramic capacitor rated to 50V with X7R dielectric in a 1206 case size will be used. From the Design Considerations section, input rms current is: IIN-RMS = 0.35 x Sqrt(0.154 x 0.846) = 126 mA Ripple current ratings for 1206 size ceramic capacitors are typically higher than 1A, more than enough for this design. RSNS = 0.74Ω, tSNS = 220 ns Sub-1Ω resistors are available in both 1% and 5% tolerance. A 1%, 0.75Ω resistor will give the best accuracy of the average LED current. To determine the resistor size the power dissipation can be calculated as: PSNS = (IF)2 x RSNS PSNS = 0.352 x 0.75 = 92 mW Standard 0805 size resistors are rated to 125 mW and will be suitable for this application. To select the proper output capacitor the equation from Buck Regulators with Output Capacitors is re-arranged to yield the following: RECIRCULATING DIODE The first parameter for D1 which must be determined is the reverse voltage rating. Schottky diodes are available at reverse ratings of 30V and 40V, often in the same package, with the same forward current rating. To account for ringing a 40V Schottky will be used. The next parameters to be determined are the forward current rating and case size. In this example the low duty cycle (D = 3.7 / 24 = 15%) requires the recirculating diode D1 to carry the load current much longer than the internal power MOSFET of the LM3402. The estimated average diode current is: ID = 0.35 x 0.85 = 298 mA Schottky diodes are available at forward current ratings of 0.5A, however the current rating often assumes a 25°C ambient temperature and does not take into account the application restrictions on temperature rise. A diode rated for higher current may be needed to keep the temperature rise below 40°C.To determine the proper case size, the dissipation and temperature rise in D1 can be calculated as shown in the Design Considerations section. VD for a small case size such as SOD-123 in a 40V, 0.5A Schottky diode at 350 mA is approximately 0.4V and the θJA is 206°C/W. Power dissipation and temperature rise can be calculated as: PD = 0.298 x 0.4 = 119 mW TRISE = 0.119 x 206 = 24.5°C According to these calculations the SOD-123 diode will meet the requirements. Heating and dissipation are among the fac17 www.national.com The target tolerance for LED ripple current is ±5% or 10%PP = 35 mAP-P, and the LED datasheet gives a typical value for rD of 1.0Ω at 350 mA. The required capacitor impedance to reduce the worst-case inductor ripple current of 258 mAP-P is therefore: ZC = [0.035 / (0.258 - 0.035] x 1.0 = 0.157Ω A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 468 kHz: LM3402/LM3402HV tors most difficult to predict in converter design. If possible, a footprint should be used that is capable of accepting both SOD-123 and a larger case size, such as SMA. A larger diode with a higher forward current rating will generally have a lower forward voltage, reducing dissipation, as well as having a lower θJA, reducing temperature rise. CB and CF The bootstrap capacitor CB should always be a 10 nF ceramic capacitor with X7R dielectric. A 25V rating is appropriate for all application circuits. The linear regulator filter capacitor CF should always be a 100 nF ceramic capacitor, also with X7R dielectric and a 25V rating. EFFICIENCY To estimate the electrical efficiency of this example the power dissipation in each current carrying element can be calculated and summed. This term should not be confused with the optical efficacy of the circuit, which depends upon the LEDs themselves. Total output power, PO, is calculated as: PO = IF x VO = 0.35 x 3.7 = 1.295W Conduction loss, PC, in the internal MOSFET: PC = (IF2 x RDSON) x D = (0.352 x 1.5) x 0.154 = 28 mW Gate charging and VCC loss, PG, in the gate drive and linear regulator: PG = (IIN-OP + fSW x QG) x VIN PG = (600 x 10-6 + 468000 x 3 x 10-9) x 24 = 48 mW Switching loss, PS, in the internal MOSFET: PS = 0.5 x VIN x IF x (tR + tF) x fSW PS = 0.5 x 24 x 0.35 x (40 x 10-9) x 468000 = 78 mW AC rms current loss, PCIN, in the input capacitor: PCIN = IIN(rms)2 x ESR = (0.126)2 x 0.006 = 0.1 mW (negligible) DCR loss, PL, in the inductor PL = IF2 x DCR = 0.352 x 0.096 = 11.8 mW Recirculating diode loss, PD = 119 mW Current Sense Resistor Loss, PSNS = 92 mW Electrical efficiency, η = PO / (PO + Sum of all loss terms) = 1.295 / (1.295 + 0.377) = 77% DIE TEMPERATURE TLM3402 = (PC + PG + PS) x θJA TLM3402 = (0.028 + 0.05 + 0.078) x 200 = 31°C ±5% DC input rail that feeds three LM3402HV current regulators to drive one series array each of red, green, and blue 1W LEDs. The target for average LED current is 350 mA ±5% in each string. The monitor will adjust the color temperature dynamically, requiring fast PWM dimming of each string with external, parallel MOSFETs. 1W green and blue InGaN LEDs have a typical forward voltage of 3.5V, however red LEDs use AlInGaP technology with a typical forward voltage of 2.9V. In order to match color properly the design requires 14 green LEDs, twice as many as needed for the red and blue LEDs. This example will follow the design for the green LED array, providing the necessary information to repeat the exercise for the blue and red LED arrays. The circuit schematic for Design Example 2 is the same as the Typical Application on the front page. The bill of materials (green array only) can be found in Table 2 at the end of this datasheet. OUTPUT VOLTAGE Green Array: VO(G) = 14 x 3.5 + 0.2 = 49.2V Blue Array: VO(B) = 7 x 3.5 + 0.2 = 24.7V Red Array: VO(R) = 7 x 2.9 + 0.2 = 20.5V RON and tON A compromise in switching frequency is needed in this application to balance the requirements of magnetics size and efficiency. The high duty cycle translates into large conduction losses and high temperature rise in the IC. For best response to a PWM dimming signal this circuit will not use an output capacitor; hence a moderate switching frequency of 300 kHz will keep the inductance from becoming so large that a custom-wound inductor is needed. This design will use only surface mount components, and the selection of off-the-shelf SMT inductors for switching regulators is poor at 1000 µH and above. RON is selected from the equation for switching frequency as follows: RON = 49.2 / (1.34 x 10-10 x 3 x 105) = 1224 kΩ The closest 1% tolerance resistor is 1.21 MΩ. The switching frequency and on-time of the circuit can then be found using the equations relating RON and tON to fSW: fSW = 49.2 / (1210000 x 1.34 x 10-10) = 303 kHz tON = (1.34 x 10-10 x 1210000) / 60 = 2.7 µs USING AN OUTPUT CAPACITOR This application is dominated by the need for fast PWM dimming, requiring a circuit without any output capacitance. Design Example 2: LM3402HV The second example application is an RGB backlight for a flat screen monitor. A separate boost regulator provides a 60V www.national.com 18 OUTPUT INDUCTOR In this example the ripple current through the LED array and the inductor are equal. Inductance is selected to give the smallest ripple current possible while still providing enough ΔvSNS signal for the CS comparator to operate correctly. De- LM3402/LM3402HV signing to a desired ΔvSNS of 25 mV and assuming that the average inductor current will equal the desired average LED current of 350 mA yields the target current ripple in the inductor and LEDs: ΔiF = ΔiL = ΔvSNS / RSNS, RSNS = VSNS / IF ΔiF = 0.025 / 0.57 = 43.8 mA With the target ripple current determined the inductance can be chosen: shelf inductor rated to 1.2A (peak) and 0.72A (average) with a DCR of 1.1Ω will be used for the green LED array. RSNS A preliminary value for RSNS was determined in selecting ΔiL. This value should be re-evaluated based on the calculations for ΔiF: LMIN = [(60 – 49.2) x 2.7 x 10-6] / (0.044) = 663 µH The closest standard inductor value is 680 µH. As with the previous example, the average current rating should be greater than 350 mA. Separation between the LM3402HV drivers and the LED arrays mean that heat from the inductor will not threaten the lifetime of the LEDs, but an overheated inductor could still cause the LM3402HV to enter thermal shutdown. The inductance itself of the standard part chosen is ±20%. With this tolerance the typical, minimum, and maximum inductor current ripples can be calculated: ΔiF(TYP) = [(60 - 49.2) x 2.7 x 10-6] / 680 x 10-6 = 43 mAP-P ΔiF(MIN) = [(60 - 49.2) x 2.7 x 10-6] / 816 x 10-6 = 36 mAP-P ΔiF(MAX) = [(60 - 49.2) x 2.7 x 10-6] / 544 x 10-6 = 54 mAP-P The peak LED/inductor current is then estimated: IL(PEAK) = IL + [ΔiL(MAX)] / 2 IL(PEAK) = 0.35 + 0.027 = 377 mA In the case of a short circuit across the LED array, the LM3402HV will continue to deliver rated current through the short but will reduce the output voltage to equal the CS pin voltage of 200 mV. Worst-case peak current in this condition would be equal to: ΔiF(LED-SHORT) = [(63 – 0.2) x 2.7 x 10-6] / 544 x 10-6 = 314 mAP-P IF(PEAK) = 0.35 + 0.156 = 506 mA In the case of a short at the switch node, the output, or from the CS pin to ground the short circuit current limit will engage at a typical peak current of 735 mA. In order to prevent inductor saturation during these fault conditions the inductor’s peak current rating must be above 735 mA. A 680 µH off-the 19 Sub-1Ω resistors are available in both 1% and 5% tolerance. A 1%, 0.56Ω device is the closest value, and a 0.125W, 0805 size device will handle the power dissipation of 69 mW. With the resistance selected, the average value of LED current is re-calculated to ensure that current is within the ±5% tolerance requirement. From the expression for LED current accuracy: IF = 0.19 / 0.56 + 0.043 / 2 = 361 mA, 3% above 350 mA INPUT CAPACITOR Following the calculations from the Input Capacitor section, ΔvIN(MAX) will be 1%P-P = 600 mV. The minimum required capacitance is: CIN(MIN) = (0.35 x 2.7 x 10-6) / 0.6 = 1.6 µF In expectation that more capacitance will be needed to prevent power supply interaction a 2.2 µF ceramic capacitor rated to 100V with X7R dielectric in an 1812 case size will be used. From the Design Considerations section, input rms current is: IIN-RMS = 0.35 x Sqrt(0.82 x 0.18) = 134 mA Ripple current ratings for 1812 size ceramic capacitors are typically higher than 2A, more than enough for this design. RECIRCULATING DIODE The input voltage of 60V ±5% requires Schottky diodes with a reverse voltage rating greater than 60V. Some manufacturers provide Schottky diodes with ratings of 70, 80 or 90V; however the next highest standard voltage rating is 100V. Selecting a 100V rated diode provides a large safety margin for the ringing of the switch node and also makes cross-referencing of diodes from different vendors easier. The next parameters to be determined are the forward current rating and case size. In this example the high duty cycle (D = 49.2 / 60 = 82%) places less thermals stress on D1 and more on the internal power MOSFET of the LM3402. The estimated average diode current is: ID = 0.361 x 0.18 = 65 mA A Schottky with a forward current rating of 0.5A would be adequate, however at 100V the majority of diodes have a minimum forward current rating of 1A. To determine the proper case size, the dissipation and temperature rise in D1 can be calculated as shown in the Design Considerations section. VD for a small case size such as SOD-123F in a 100V, 1A www.national.com LM3402/LM3402HV Schottky diode at 350 mA is approximately 0.65V and the θJA is 88°C/W. Power dissipation and temperature rise can be calculated as: PD = 0.065 x 0.65 = 42 mW TRISE = 0.042 x 88 = 4°C CB AND CF The bootstrap capacitor CB should always be a 10 nF ceramic capacitor with X7R dielectric. A 25V rating is appropriate for all application circuits. The linear regulator filter capacitor CF should always be a 100 nF ceramic capacitor, also with X7R dielectric and a 25V rating. EFFICIENCY To estimate the electrical efficiency of this example the power dissipation in each current carrying element can be calculated and summed. Electrical efficiency, η, should not be confused with the optical efficacy of the circuit, which depends upon the LEDs themselves. Total output power, PO, is calculated as: PO = IF x VO = 0.361 x 49.2 = 17.76W Conduction loss, PC, in the internal MOSFET: PC = (IF2 x RDSON) x D = (0.3612 x 1.5) x 0.82 = 160 mW Gate charging and VCC loss, PG, in the gate drive and linear regulator: PG = (IIN-OP + fSW x QG) x VIN PG = (600 x 10-6 + 3 x 105 x 3 x 10-9) x 60 = 90 mW Switching loss, PS, in the internal MOSFET: PS = 0.5 x VIN x IF x (tR + tF) x fSW PS = 0.5 x 60 x 0.361 x 40 x 10-9 x 3 x 105 = 130 mW AC rms current loss, PCIN, in the input capacitor: PCIN = IIN(rms)2 x ESR = (0.134)2 x 0.006 = 0.1 mW (negligible) DCR loss, PL, in the inductor PL = IF2 x DCR = 0.352 x 1.1 = 135 mW Recirculating diode loss, PD = 42 mW Current Sense Resistor Loss, PSNS = 69 mW Electrical efficiency, η = PO / (PO + Sum of all loss terms) = 17.76 / (17.76 + 0.62) = 96% Temperature Rise in the LM3402HV IC is calculated as: TLM3402 = (PC + PG + PS) x θJA = (0.16 + 0.084 + 0.13) x 200 = 74.8°C Layout Considerations The performance of any switching converter depends as much upon the layout of the PCB as the component selection. The following guidelines will help the user design a circuit with maximum rejection of outside EMI and minimum generation of unwanted EMI. COMPACT LAYOUT Parasitic inductance can be reduced by keeping the power path components close together and keeping the area of the loops that high currents travel small. Short, thick traces or copper pours (shapes) are best. In particular, the switch node (where L1, D1, and the SW pin connect) should be just large enough to connect all three components without excessive heating from the current it carries. The LM3402/02HV operates in two distinct cycles whose high current paths are shown in Figure 6: 20192128 FIGURE 6. Buck Converter Current Loops The dark grey, inner loop represents the high current path during the MOSFET on-time. The light grey, outer loop represents the high current path during the off-time. GROUND PLANE AND SHAPE ROUTING The diagram of Figure 6 is also useful for analyzing the flow of continuous current vs. the flow of pulsating currents. The circuit paths with current flow during both the on-time and offtime are considered to be continuous current, while those that carry current during the on-time or off-time only are pulsating currents. Preference in routing should be given to the pulsating current paths, as these are the portions of the circuit most likely to emit EMI. The ground plane of a PCB is a conductor and return path, and it is susceptible to noise injection just as any other circuit path. The continuous current paths on the ground net can be routed on the system ground plane with www.national.com 20 LM3402/LM3402HV less risk of injecting noise into other circuits. The path between the input source and the input capacitor and the path between the recirculating diode and the LEDs/current sense resistor are examples of continuous current paths. In contrast, the path between the recirculating diode and the input capacitor carries a large pulsating current. This path should be routed with a short, thick shape, preferably on the component side of the PCB. Multiple vias in parallel should be used right at the pad of the input capacitor to connect the component side shapes to the ground plane. A second pulsating current loop that is often ignored is the gate drive loop formed by the SW and BOOT pins and capacitor CB. To minimize this loop at the EMI it generates, keep CB close to the SW and BOOT pins. CURRENT SENSING The CS pin is a high-impedance input, and the loop created by RSNS, RZ (if used), the CS pin and ground should be made as small as possible to maximize noise rejection. RSNS should therefore be placed as close as possible to the CS and GND pins of the IC. REMOTE LED ARRAYS In some applications the LED or LED array can be far away (several inches or more) from the LM3402/02HV, or on a separate PCB connected by a wiring harness. When an output capacitor is used and the LED array is large or separated from the rest of the converter, the output capacitor should be placed close to the LEDs to reduce the effects of parasitic inductance on the AC impedance of the capacitor. The current sense resistor should remain on the same PCB, close to the LM3402/02HV. 21 www.national.com LM3402/LM3402HV TABLE 1. BOM for Design Example 1 ID U1 L1 D1 Cf Cb Cin Co Rsns Ron Part Number LM3402 SLF7045T-330MR82 CMHSH5-4 VJ0805Y104KXXAT VJ0805Y103KXXAT C3216X7R1H105M C2012X7R1A225M ERJ6BQFR75V CRCW08055902F Type LED Driver Inductor Schottky Diode Capacitor Capacitor Capacitor Capacitor Resistor Resistor Size MSOP-8 7.0x7.0 x4.5mm SOD-123 0805 0805 1206 0805 0805 0805 Parameters 40V, 0.5A 33µH, 0.82A, 96mΩ 40V, 0.5A 100nF 10% 10nF 10% 1µF 50V 2.2 µF 10V 0.75Ω 1% 59.0 kΩ 1% Qty 1 1 1 1 1 1 1 1 1 Vendor NSC TDK Central Semi Vishay Vishay TDK TDK Panasonic Vishay TABLE 2. BOM for Design Example 2 ID U1 L1 D1 Cf Cb Cin Rsns Ron Part Number LM3402HV DO5022P-684 CMMSH1-100 VJ0805Y104KXXAT VJ0805Y103KXXAT C4532X7R2A225M ERJ6BQFR56V CRCW08051214F Type LED Driver Inductor Schottky Diode Capacitor Capacitor Capacitor Resistor Resistor Size MSOP-8 18.5x15.2 x7.1mm SOD-123F 0805 0805 1812 0805 0805 Parameters 75V, 0.5A 680µH, 1.2A, 1.1Ω 100V, 1A 100nF 10% 10nF 10% 2.2µF 100V 0.56Ω 1% 1.21MΩ 1% Qty 1 1 1 1 1 1 1 1 Vendor NSC Coilcraft Central Semi Vishay Vishay TDK Panasonic Vishay www.national.com 22 LM3402/LM3402HV Physical Dimensions inches (millimeters) unless otherwise noted 8-Lead MSOP Package NS Package Number MUA08A 8-Lead PSOP Package NS Package Number MRA08B 23 www.national.com LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs Notes THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. NATIONAL MAKES NO REPRESENTATIONS OR WARRANTIES WITH RESPECT TO THE ACCURACY OR COMPLETENESS OF THE CONTENTS OF THIS PUBLICATION AND RESERVES THE RIGHT TO MAKE CHANGES TO SPECIFICATIONS AND PRODUCT DESCRIPTIONS AT ANY TIME WITHOUT NOTICE. NO LICENSE, WHETHER EXPRESS, IMPLIED, ARISING BY ESTOPPEL OR OTHERWISE, TO ANY INTELLECTUAL PROPERTY RIGHTS IS GRANTED BY THIS DOCUMENT. TESTING AND OTHER QUALITY CONTROLS ARE USED TO THE EXTENT NATIONAL DEEMS NECESSARY TO SUPPORT NATIONAL’S PRODUCT WARRANTY. EXCEPT WHERE MANDATED BY GOVERNMENT REQUIREMENTS, TESTING OF ALL PARAMETERS OF EACH PRODUCT IS NOT NECESSARILY PERFORMED. NATIONAL ASSUMES NO LIABILITY FOR APPLICATIONS ASSISTANCE OR BUYER PRODUCT DESIGN. BUYERS ARE RESPONSIBLE FOR THEIR PRODUCTS AND APPLICATIONS USING NATIONAL COMPONENTS. PRIOR TO USING OR DISTRIBUTING ANY PRODUCTS THAT INCLUDE NATIONAL COMPONENTS, BUYERS SHOULD PROVIDE ADEQUATE DESIGN, TESTING AND OPERATING SAFEGUARDS. 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Copyright© 2007 National Semiconductor Corporation For the most current product information visit us at www.national.com National Semiconductor Americas Customer Support Center Email: new.feedback@nsc.com Tel: 1-800-272-9959 National Semiconductor Europe Customer Support Center Fax: +49 (0) 180-530-85-86 Email: europe.support@nsc.com Deutsch Tel: +49 (0) 69 9508 6208 English Tel: +49 (0) 870 24 0 2171 Français Tel: +33 (0) 1 41 91 8790 National Semiconductor Asia Pacific Customer Support Center Email: ap.support@nsc.com National Semiconductor Japan Customer Support Center Fax: 81-3-5639-7507 Email: jpn.feedback@nsc.com Tel: 81-3-5639-7560 www.national.com