LM3404HV

LM3404/04HV 1.0A Constant Current Buck Regulator for Driving High Power LEDs February 8, 2010 LM3404/04HV 1.0A Constant Current Buck Regulator for Driving High Power LEDs General Description The LM3404/04HV are monolithic switching regulators designed to deliver constant currents to high power LEDs. Ideal for automotive, industrial, and general lighting applications, they contain a high-side N-channel MOSFET switch with a current limit of 1.5A (typical) for step-down (Buck) regulators. Hysteretic controlled on-time and an external resistor allow the converter output voltage to adjust as needed to deliver a constant current to series and series-parallel connected LED arrays of varying number and type. LED dimming via pulse width modulation (PWM), broken/open LED protection, lowpower shutdown and thermal shutdown complete the feature set. Features ■ ■ ■ ■ ■ ■ ■ ■ Integrated 1.0A MOSFET VIN Range 6V to 42V (LM3404) VIN Range 6V to 75V (LM3404HV) 1.2A Output Current Over Temperature Cycle-by-Cycle Current Limit No Control Loop Compensation Required Separate PWM Dimming and Low Power Shutdown Supports all-ceramic output capacitors and capacitor-less outputs ■ Thermal shutdown protection ■ SO-8 Package, PSOP-8 Package Applications ■ ■ ■ ■ ■ LED Driver Constant Current Source Automotive Lighting General Illumination Industrial Lighting Typical Application 20205401 © 2010 National Semiconductor Corporation 202054 www.national.com LM3404/LM3404HV Connection Diagrams 20205456 20205402 8-Lead Plastic PSOP-8 Package NS Package Number MRA08B 8-Lead Plastic SO-8 Package NS Package Number M08A Ordering Information Order Number LM3404MA LM3404MAX LM3404HVMA LM3404HVMAX LM3404MR LM3404MRX LM3404HVMR LM3404HVMRX PSOP-8 MRA08B SO-8 M08A Package Type NSC Package Drawing Supplied As 95 units in anti-static rails 2500 units on tape and reel 95 units in anti-static rails 2500 units on tape and reel 95 units in anti-static rails 2500 units on tape and reel 95 units in anti-static rails 2500 units on tape and reel Pin Descriptions Pin(s) 1 2 3 4 5 6 7 8 DAP Name SW BOOT DIM GND CS RON VCC VIN GND Description Switch pin MOSFET drive bootstrap pin Input for PWM dimming Ground pin Current sense feedback pin On-time control pin Output of the internal 7V linear regulator Input voltage pin Thermal Pad Application Information Connect this pin to the output inductor and Schottky diode. Connect a 10 nF ceramic capacitor from this pin to SW. Connect a logic-level PWM signal to this pin to enable/disable the power MOSFET and reduce the average light output of the LED array. Connect this pin to system ground. Set the current through the LED array by connecting a resistor from this pin to ground. A resistor connected from this pin to VIN sets the regulator controlled on-time. Bypass this pin to ground with a minimum 0.1 µF ceramic capacitor with X5R or X7R dielectric. Nominal operating input range for this pin is 6V to 42V (LM3404) or 6V to 75V (LM3404HV). Connect to ground. Place 4-6 vias from DAP to bottom layer ground plane. www.national.com 2 LM3404/LM3404HV Absolute Maximum Ratings (LM3404) (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN to GND BOOT to GND SW to GND BOOT to VCC BOOT to SW VCC to GND DIM to GND CS to GND RON to GND Junction Temperature -0.3V to 45V -0.3V to 59V -1.5V to 45V -0.3V to 45V -0.3V to 14V -0.3V to 14V -0.3V to 7V -0.3V to 7V -0.3V to 7V 150°C Storage Temp. Range ESD Rating (Note 2) Soldering Information Lead Temperature (Soldering, 10sec) Infrared/Convection Reflow (15sec) -65°C to 125°C 2kV 260°C 235°C Operating Ratings (LM3404) (Note 1) VIN Junction Temperature Range Thermal Resistance θJA (SO-8 Package) Thermal Resistance θJA (PSOP-8 Package) (Note 5) 6V to 42V −40°C to +125°C 155°C/W 50°C/W 3 www.national.com LM3404/LM3404HV Absolute Maximum Ratings (LM3404HV) (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN to GND BOOT to GND SW to GND BOOT to VCC BOOT to SW VCC to GND DIM to GND CS to GND RON to GND Junction Temperature -0.3V to 76V -0.3V to 90V -1.5V to 76V -0.3V to 76V -0.3V to 14V -0.3V to 14V -0.3V to 7V -0.3V to 7V -0.3V to 7V 150°C Storage Temp. Range ESD Rating (Note 2) Soldering Information Lead Temperature (Soldering, 10sec) Infrared/Convection Reflow (15sec) -65°C to 125°C 2kV 260°C 235°C Operating Ratings (LM3404HV) (Note 1) VIN Junction Temperature Range Thermal Resistance θJA (SO-8 Package) Thermal Resistance θJA (PSOP-8 Package) (Note 5) 6V to 75V −40°C to +125°C 155°C/W 50°C/W www.national.com 4 LM3404/LM3404HV Electrical Characteristics LM3404 Symbol SYSTEM PARAMETERS tON-1 tON-2 On-time 1 On-time 2 Parameter VIN = 24V unless otherwise indicated. Typicals and limits appearing in plain type apply for TA = TJ = +25°C. (Note 4) Limits appearing in boldface type apply over full Operating Temperature Range. Datasheet min/ max specification limits are guaranteed by design, test, or statistical analysis. Conditions VIN = 10V, RON = 200 kΩ VIN = 40V, RON = 200 kΩ Min 2.1 515 Typ 2.75 675 Max 3.4 835 Units µs ns LM3404HV Symbol SYSTEM PARAMETERS tON-1 tON-2 On-time 1 On-time 2 VIN = 10V, RON = 200 kΩ VIN = 70V, RON = 200 kΩ 2.1 325 2.75 415 3.4 505 µs ns Parameter Conditions Min Typ Max Units LM3404/LM3404HV Symbol VREF-REG VREF-0V ICS SHUTDOWN VSD-TH VSD-HYS OFF TIMER tOFF-MIN VCC-REG VIN-DO VCC-BP-TH VCC-BP-HYS VCC-Z-6 VCC-Z-8 VCC-Z-24 VCC-LIM VCC-UV-TH VCC-UV-HYS VCC-UV-DLY IIN-OP IIN-SD CURRENT LIMIT ILIM Current Limit Threshold 1.2 1.5 1.8 A VCC Current Limit (Note 3) VCC Under-voltage Lock-out Threshold VCC Under-voltage Lock-out Hysteresis VCC Under-voltage Lock-out Filter Delay IIN Operating Current IIN Shutdown Current Minimum Off-time VCC Regulated Output VIN - VCC VCC Bypass Threshold VCC Bypass Hysteresis VCC Output Impedance (0 mA < ICC < 5 mA) ICC = 5 mA, 6.0V < VIN < 8.0V VIN Increasing VIN Decreasing VIN = 6V VIN = 8V VIN = 24V VIN = 24V, VCC = 0V VCC Increasing VCC Decreasing 100 mV Overdrive Non-switching, CS = 0.5V RON / SD = 0V CS = 0V 6.4 270 7 300 8.8 230 55 50 0.4 16 5.3 150 3 625 95 900 180 mA V mV µs µA µA 7.4 ns V mV V mV Ω INTERNAL REGULATOR Shutdown Threshold Shutdown Hysteresis RON / SD Increasing RON / SD Decreasing 0.3 0.7 40 1.05 V mV Parameter CS Regulation Threshold CS Over-voltage Threshold CS Bias Current Conditions CS Decreasing, SW turns on CS Increasing, SW turns off CS = 0V Min 194 Typ 200 300 0.1 Max 206 Units mV mV µA REGULATION AND OVER-VOLTAGE COMPARATORS 5 www.national.com LM3404/LM3404HV Symbol DIM COMPARATOR VIH VIL IDIM-PU RDS-ON VDR-UVLO VDR-HYS Parameter Logic High Logic Low DIM Pull-up Current Buck Switch On Resistance BST Under-voltage Lock-out Threshold BST Under-voltage Lock-out Hysteresis Thermal Shutdown Threshold Thermal Shutdown Hysteresis Junction to Ambient Conditions DIM Increasing DIM Decreasing DIM = 1.5V ISW = 200mA, BST-SW = 6.3V BST–SW Increasing BST–SW Decreasing Min 2.2 Typ Max Units V 0.8 80 0.37 1.7 3 400 0.75 4 V µA Ω V mV MOSFET AND DRIVER THERMAL SHUTDOWN TSD TSD-HYS THERMAL RESISTANCE 165 25 SOIC-8 Package PSOP-8 Package (Note 5) 155 50 °C °C °C/W θJA Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see Electrical Characteristics. Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Note 3: VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading. Note 4: Typical specifications represent the most likely parametric norm at 25°C operation. Note 5: θJA of 50°C/W with DAP soldered to a minimum of 2 square inches of 1oz. copper on the top or bottom PCB layer. www.national.com 6 LM3404/LM3404HV Typical Performance Characteristics VREF vs Temperature (VIN = 24V) VREF vs VIN, LM3404 (TA = 25°C) 20205450 20205451 VREF vs VIN, LM3404HV (TA = 25°C) Current Limit vs Temperature (VIN = 24V) 20205452 20205453 Current Limit vs VIN, LM3404 (TA = 25°C) Current Limit vs VIN, LM3404HV (TA = 25°C) 20205454 20205455 7 www.national.com LM3404/LM3404HV TON vs VIN, RON = 100 kΩ (TA = 25°C) TON vs VIN, (TA = 25°C) 20205435 20205436 TON vs VIN, (TA = 25°C) TON vs RON, LM3404 (TA = 25°C) 20205437 20205444 TON vs RON, LM3404HV (TA = 25°C) VCC vs VIN (TA = 25°C) 20205438 20205439 www.national.com 8 LM3404/LM3404HV VO-MAX vs fSW, LM3404 (TA = 25°C) VO-MIN vs fSW, LM3404 (TA = 25°C) 20205440 20205441 VO-MAX vs fSW, LM3404HV (TA = 25°C) VO-MIN vs fSW, LM3404HV (TA = 25°C) 20205442 20205443 9 www.national.com LM3404/LM3404HV Block Diagram 20205403 Application Information THEORY OF OPERATION The LM3404 and LM3404HV are buck regulators with a wide input voltage range, low voltage reference, and a fast output enable/disable function. These features combine to make them ideal for use as a constant current source for LEDs with forward currents as high as 1.2A. The controlled on-time (COT) architecture is a combination of hysteretic mode control and a one-shot on-timer that varies inversely with input voltage. Hysteretic operation eliminates the need for smallsignal control loop compensation. When the converter runs in continuous conduction mode (CCM) the controlled on-time maintains a constant switching frequency over the range of input voltage. Fast transient response, PWM dimming, a low power shutdown mode, and simple output overvoltage protection round out the functions of the LM3404/04HV. CONTROLLED ON-TIME OVERVIEW Figure 1 shows the feedback system used to control the current through an array of LEDs. A voltage signal, VSNS, is created as the LED current flows through the current setting resistor, RSNS, to ground. VSNS is fed back to the CS pin, where it is compared against a 200 mV reference, VREF. The on-comparator turns on the power MOSFET when VSNS falls below VREF. The power MOSFET conducts for a controlled on-time, tON, set by an external resistor, RON, and by the input voltage, VIN. On-time is governed by the following equation: At the conclusion of tON the power MOSFET turns off for a minimum off-time, tOFF-MIN, of 300 ns. Once tOFF-MIN is complete the CS comparator compares VSNS and VREF again, waiting to begin the next cycle. 20205405 FIGURE 1. Comparator and One-Shot The LM3404/04HV regulators should be operated in continuous conduction mode (CCM), where inductor current stays positive throughout the switching cycle. During steady-state 10 www.national.com LM3404/LM3404HV CCM operation, the converter maintains a constant switching frequency that can be selected using the following equation: MINIMUM OUTPUT VOLTAGE The minimum recommended on-time for the LM3404/04HV is 300 ns. This lower limit for tON determines the minimum duty cycle and output voltage that can be regulated based on input voltage and switching frequency. The relationship is determined by the following equation, shown on the same graphs as maximum output voltage in the Typical Performance Characteristics section: VF = forward voltage of each LED, n = number of LEDs in series AVERAGE LED CURRENT ACCURACY The COT architecture regulates the valley of ΔVSNS, the AC portion of VSNS. To determine the average LED current (which is also the average inductor current) the valley inductor current is calculated using the following expression: HIGH VOLTAGE BIAS REGULATOR The LM3404/04HV contains an internal linear regulator with a 7V output, connected between the VIN and the VCC pins. The VCC pin should be bypassed to the GND pin with a 0.1 µF ceramic capacitor connected as close as possible to the pins of the IC. VCC tracks VIN until VIN reaches 8.8V (typical) and then regulates at 7V as VIN increases. Operation begins when VCC crosses 5.25V. INTERNAL MOSFET AND DRIVER The LM3404/04HV features an internal power MOSFET as well as a floating driver connected from the SW pin to the BOOT pin. Both rise time and fall time are 20 ns each (typical) and the approximate gate charge is 6 nC. The high-side rail for the driver circuitry uses a bootstrap circuit consisting of an internal high-voltage diode and an external 10 nF capacitor, CB. VCC charges CB through the internal diode while the power MOSFET is off. When the MOSFET turns on, the internal diode reverse biases. This creates a floating supply equal to the VCC voltage minus the diode drop to drive the MOSFET when its source voltage is equal to VIN. FAST SHUTDOWN FOR PWM DIMMING The DIM pin of the LM3404/04HV is a TTL compatible input for low frequency PWM dimming of the LED. A logic low (below 0.8V) at DIM will disable the internal MOSFET and shut off the current flow to the LED array. While the DIM pin is in a logic low state the support circuitry (driver, bandgap, VCC) remains active in order to minimize the time needed to turn the LED array back on when the DIM pin sees a logic high (above 2.2V). A 75 µA (typical) pull-up current ensures that the LM3404/04HV is on when DIM pin is open circuited, eliminating the need for a pull-up resistor. Dimming frequency, fDIM, and duty cycle, DDIM, are limited by the LED current rise time and fall time and the delay from activation of the DIM pin to the response of the internal power MOSFET. In general, fDIM should be at least one order of magnitude lower than the steady state switching frequency in order to prevent aliasing. PEAK CURRENT LIMIT The current limit comparator of the LM3404/04HV will engage whenever the power MOSFET current (equal to the inductor current while the MOSFET is on) exceeds 1.5A (typical). The power MOSFET is disabled for a cool-down time that is approximately 75x the steady-state on-time. At the conclusion of this cool-down time the system re-starts. If the current limit condition persists the cycle of cool-down time and restarting will continue, creating a low-power hiccup mode, minimizing thermal stress on the LM3404/04HV and the external circuit components. OVER-VOLTAGE/OVER-CURRENT COMPARATOR The CS pin includes an output over-voltage/over-current comparator that will disable the power MOSFET whenever 11 www.national.com In this equation tSNS represents the propagation delay of the CS comparator, and is approximately 220 ns. The average inductor/LED current is equal to IL-MIN plus one-half of the inductor current ripple, ΔiL: IF = IL = IL-MIN + ΔiL / 2 Detailed information for the calculation of ΔiL is given in the Design Considerations section. MAXIMUM OUTPUT VOLTAGE The 300 ns minimum off-time limits the maximum duty cycle of the converter, DMAX, and in turn the maximum output voltage, VO(MAX), determined by the following equations: The maximum number of LEDs, nMAX, that can be placed in a single series string is governed by VO(MAX) and the maximum forward voltage of the LEDs used, VF(MAX), using the expression: At low switching frequency the maximum duty cycle and output voltage are higher, allowing the LM3404/04HV to regulate output voltages that are nearly equal to input voltage. The following equation relates switching frequency to maximum output voltage, and is also shown graphically in the Typical Performance Characteristics section: LM3404/LM3404HV VSNS exceeds 300 mV. This threshold provides a hard limit for the output current. Output current overshoot is limited to 300 mV / RSNS by this comparator during transients. The OVP/OCP comparator can also be used to prevent the output voltage from rising to VO(MAX) in the event of an output open-circuit. This is the most common failure mode for LEDs, due to breaking of the bond wires. In a current regulator an output open circuit causes VSNS to fall to zero, commanding maximum duty cycle. Figure 2 shows a method using a zener diode, Z1, and zener limiting resistor, RZ, to limit output voltage to the reverse breakdown voltage of Z1 plus 200 mV. The zener diode reverse breakdown voltage, VZ, must be greater than the maximum combined VF of all LEDs in the array. The maximum recommended value for RZ is 1 kΩ. As discussed in the Maximum Output Voltage section, there is a limit to how high VO can rise during an output open-circuit that is always less than VIN. If no output capacitor is used, the output stage of the LM3404/04HV is capable of withstanding VO(MAX) indefinitely, however the voltage at the output end of the inductor will oscillate and can go above VIN or below 0V. A small (typically 10 nF) capacitor across the LED array dampens this oscillation. For circuits that use an output capacitor, the system can still withstand VO(MAX) indefinitely as long as CO is rated to handle VIN. The high current paths are blocked in output open-circuit and the risk of thermal stress is minimal, hence the user may opt to allow the output voltage to rise in the case of an open-circuit LED failure. 20205412 FIGURE 2. Output Open Circuit Protection LOW POWER SHUTDOWN The LM3404/04HV can be placed into a low power state (IINSD = 90 µA) by grounding the RON pin with a signal-level MOSFET as shown in Figure 3. Low power MOSFETs like the 2N7000, 2N3904, or equivalent are recommended devices for putting the LM3404/04HV into low power shutdown. Logic gates can also be used to shut down the LM3404/04HV as long as the logic low voltage is below the over temperature minimum threshold of 0.3V. Noise filter circuitry on the RON pin can cause a few pulses with longer on-times than normal after RON is grounded or released. In these cases the OVP/ OCP comparator will ensure that the peak inductor or LED current does not exceed 300 mV / RSNS. 20205413 FIGURE 3. Low Power Shutdown www.national.com 12 LM3404/LM3404HV THERMAL SHUTDOWN Internal thermal shutdown circuitry is provided to protect the IC in the event that the maximum junction temperature is exceeded. The threshold for thermal shutdown is 165°C with a 25°C hysteresis (both values typical). During thermal shutdown the MOSFET and driver are disabled. Design Considerations SWITCHING FREQUENCY Switching frequency is selected based on the trade-offs between efficiency (better at low frequency), solution size/cost (smaller at high frequency), and the range of output voltage that can be regulated (wider at lower frequency.) Many applications place limits on switching frequency due to EMI sensitivity. The on-time of the LM3404/04HV can be programmed for switching frequencies ranging from the 10’s of kHz to over 1 MHz. The maximum switching frequency is limited only by the minimum on-time and minimum off-time requirements. LED RIPPLE CURRENT Selection of the ripple current, ΔiF, through the LED array is analogous to the selection of output ripple voltage in a standard voltage regulator. Where the output ripple in a voltage regulator is commonly ±1% to ±5% of the DC output voltage, LED manufacturers generally recommend values for ΔiF ranging from ±5% to ±20% of IF. Higher LED ripple current allows the use of smaller inductors, smaller output capacitors, or no output capacitors at all. The advantages of higher ripple current are reduction in the solution size and cost. Lower ripple current requires more output inductance, higher switching frequency, or additional output capacitance. The advantages of lower ripple current are a reduction in heating in the LED itself and greater tolerance in the average LED current before the current limit of the LED or the driving circuitry is reached. BUCK CONVERTERS WITHOUT OUTPUT CAPACITORS The buck converter is unique among non-isolated topologies because of the direct connection of the inductor to the load during the entire switching cycle. By definition an inductor will control the rate of change of current that flows through it, and this control over current ripple forms the basis for component selection in both voltage regulators and current regulators. A current regulator such as the LED driver for which the LM3404/04HV was designed focuses on the control of the current through the load, not the voltage across it. A constant current regulator is free of load current transients, and has no need of output capacitance to supply the load and maintain output voltage. Referring to the Typical Application circuit on the front page of this datasheet, the inductor and LED can form a single series chain, sharing the same current. When no output capacitor is used, the same equations that govern inductor ripple current, ΔiL, also apply to the LED ripple current, ΔiF. For a controlled on-time converter such as LM3404/04HV the ripple current is described by the following expression: BUCK CONVERTERS WITH OUTPUT CAPACITORS A capacitor placed in parallel with the LED or array of LEDs can be used to reduce the LED current ripple while keeping the same average current through both the inductor and the LED array. This technique is demonstrated in Design Examples 1 and 2. With this topology the output inductance can be lowered, making the magnetics smaller and less expensive. Alternatively, the circuit could be run at lower frequency but keep the same inductor value, improving the efficiency and expanding the range of output voltage that can be regulated. Both the peak current limit and the OVP/OCP comparator still monitor peak inductor current, placing a limit on how large ΔiL can be even if ΔiF is made very small. A parallel output capacitor is also useful in applications where the inductor or input voltage tolerance is poor. Adding a capacitor that reduces ΔiF to well below the target provides headroom for changes in inductance or VIN that might otherwise push the peak LED ripple current too high. Figure 4 shows the equivalent impedances presented to the inductor current ripple when an output capacitor, CO, and its equivalent series resistance (ESR) are placed in parallel with the LED array. The entire inductor ripple current flows through RSNS to provide the required 25 mV of ripple voltage for proper operation of the CS comparator. 20205415 FIGURE 4. LED and CO Ripple Current To calculate the respective ripple currents the LED array is represented as a dynamic resistance, rD. LED dynamic resistance is not always specified on the manufacturer’s datasheet, but it can be calculated as the inverse slope of the LED’s VF vs. IF curve. Note that dividing VF by IF will give an incorrect value that is 5x to 10x too high. Total dynamic resistance for a string of n LEDs connected in series can be calculated as the rD of one device multiplied by n. Inductor ripple current is still calculated with the expression from Buck Regulators without Output Capacitors. The following equations can then be used to estimate ΔiF when using a parallel capacitor: A minimum ripple voltage of 25 mV is recommended at the CS pin to provide good signal to noise ratio (SNR). The CS pin ripple voltage, ΔvSNS, is described by the following: ΔvSNS = ΔiF x RSNS 13 The calculation for ZC assumes that the shape of the inductor ripple current is approximately sinusoidal. www.national.com LM3404/LM3404HV Small values of CO that do not significantly reduce ΔiF can also be used to control EMI generated by the switching action of the LM3404/04HV. EMI reduction becomes more important as the length of the connections between the LED and the rest of the circuit increase. INPUT CAPACITORS Input capacitors at the VIN pin of the LM3404/04HV are selected using requirements for minimum capacitance and rms ripple current. The input capacitors supply pulses of current approximately equal to IF while the power MOSFET is on, and are charged up by the input voltage while the power MOSFET is off. Switching converters such as the LM3404/04HV have a negative input impedance due to the decrease in input current as input voltage increases. This inverse proportionality of input current to input voltage can cause oscillations (sometimes called ‘power supply interaction’) if the magnitude of the negative input impedance is greater the the input filter impedance. Minimum capacitance can be selected by comparing the input impedance to the converter’s negative resistance; however this requires accurate calculation of the input voltage source inductance and resistance, quantities which can be difficult to determine. An alternative method to select the minimum input capacitance, CIN(MIN), is to select the maximum input voltage ripple which can be tolerated. This value, ΔvIN(MAX), is equal to the change in voltage across CIN during the converter on-time, when CIN supplies the load current. CIN(MIN) can be selected with the following: ID = (1 – D) x IF This calculation should be done at the maximum expected input voltage. The overall converter efficiency becomes more dependent on the selection of D1 at low duty cycles, where the recirculating diode carries the load current for an increasing percentage of the time. This power dissipation can be calculating by checking the typical diode forward voltage, VD, from the I-V curve on the product datasheet and then multiplying it by ID. Diode datasheets will also provide a typical junction-to-ambient thermal resistance, θJA, which can be used to estimate the operating die temperature of the device. Multiplying the power dissipation (PD = ID x VD) by θJA gives the temperature rise. The diode case size can then be selected to maintain the Schottky diode temperature below the operational maximum. LED CURRENT DURING DIM MODE The LM3402 contains high speed MOSFET gate drive circuitry that switches the main internal power MOSFET between “on” and “off” states. This circuitry uses current derived from the VCC regulator to charge the MOSFET during turnon, then dumps current from the MOSFET gate to the source (the SW pin) during turn-off. As shown in the block diagram, the MOSFET drive circuitry contains a gate drive under-voltage lockout (UVLO) circuit that ensures the MOSFET remains off when there is inadequate VCC voltage for proper operation of the driver. This watchdog circuitry is always running including during DIM and shutdown modes, and supplies a small amount of current from VCC to SW. Because the SW pin is connected directly to the LEDs through the buck inductor, this current returns to ground through the LEDs. The amount of current sourced is a function of the SW voltage, as shown in . A good starting point for selection of CIN is to use an input voltage ripple of 5% to 10% of VIN. A minimum input capacitance of 2x the CIN(MIN) value is recommended for all LM3404/04HV circuits. To determine the rms current rating, the following formula can be used: Ceramic capacitors are the best choice for the input to the LM3404/04HV due to their high ripple current rating, low ESR, low cost, and small size compared to other types. When selecting a ceramic capacitor, special attention must be paid to the operating conditions of the application. Ceramic capacitors can lose one-half or more of their capacitance at their rated DC voltage bias and also lose capacitance with extremes in temperature. A DC voltage rating equal to twice the expected maximum input voltage is recommended. In addition, the minimum quality dielectric which is suitable for switching power supply inputs is X5R, while X7R or better is preferred. RECIRCULATING DIODE The LM3404/04HV is a non-synchronous buck regulator that requires a recirculating diode D1 (see the Typical Application circuit) to carrying the inductor current during the MOSFET off-time. The most efficient choice for D1 is a Schottky diode due to low forward drop and near-zero reverse recovery time. D1 must be rated to handle the maximum input voltage plus any switching node ringing when the MOSFET is on. In practice all switching converters have some ringing at the switching node due to the diode parasitic capacitance and the lead inductance. D1 must also be rated to handle the average current, ID, calculated as: www.national.com 14 20205457 FIGURE 5. LED Current From SW Pin Though most power LEDs are designed to run at several hundred milliamps, some can be seen to glow with a faint light at extremely low current levels, as low as a couple microamps in some instances. In lab testing, the forward voltage was found to be approximately 2V for LEDs that exhibited visible light at these low current levels. For LEDs that did not show light emission at very low current levels, the forward voltage was found to be around 900mV. It is important to remember that the forward voltage is also temperature dependent, de- LM3404/LM3404HV creasing at higher temperatures. Consequently, with a maximum Vcc voltage of 7.4V, current will be observed in the LEDs if the total stack voltage is less than about 6V at a forward current of several microamps. No current is observed if the stack voltage is above 6V, as shown in . The need for absolute darkness during DIM mode is also application dependent. It will not affect regular PWM dimming operation. The fix for this issue is extremely simple. Place a resistor from the SW pin to ground according to the chart below. Number of LEDs 1 2 3 4 5 >5 Resistor Value (kΩ) 20 50 90 150 200 300 than 5µJ (microjoules). Any event that transfers more energy than this may damage the ESD structure. Damage is typically represented as a short from the pin to ground as the extreme localized heat of the ESD / EOS event causes the aluminum metal on the chip to melt, causing the short. This situation is common to all integrated circuits and not just unique to the LM340X device. CS PIN PROTECTION When hot swapping in a load (e.g. test points, load boards, LED stack), any residual charge on the load will be immediately transferred through the output capacitor to the CS pin, which is then damaged as shown in Figure 6 below. The EOS event due to the residual charge from the load is represented as VTRANSIENT. From measurements, we know that the 8V ESD structure on the CS pin can typically withstand 25mA of direct current (DC). Adding a 1kΩ resistor in series with the CS pin, shown in Figure 7, results in the majority of the transient energy to pass through the discrete sense resistor rather than the device. The series resistor limits the peak current that can flow during a transient event, thus protecting the CS pin. With the 1kΩ resistor shown, a 33V, 49A transient on the LED return connector terminal could be absorbed as calculated by: V = 25mA * 1kΩ + 8V = 33V I = 33V / 0.67Ω = 49A This is an extremely high energy event, so the protection measures previously described should be adequate to solve this issue. The luminaire designer should ensure that the suggested resistor is effective in eliminating the off-state light output. A combination of calculations based on LED manufacturer data and lab measurements over temperature will ensure the best design. Transient Protection Considerations Considerations need to be made when external sources, loads or connections are made to the switching converter circuit due to the possibility of Electrostatic Discharge (ESD) or Electric Over Stress (EOS) events occurring and damaging the integrated circuit (IC) device. All IC device pins contain zener based clamping structures that are meant to clamp ESD. ESD events are very low energy events, typically less 20205463 FIGURE 6. CS Pin, Transient Path 15 www.national.com LM3404/LM3404HV 20205458 FIGURE 7. CS Pin, Transient Path with Protection Adding a resistor in series with the CS pin causes the observed output LED current to shift very slightly. The reason for this is twofold: (1) the CS pin has about 20pF of inherent capacitance inside it which causes a slight delay (20ns for a 1kΩ series resistor), and (2) the comparator that is watching the voltage at the CS pin uses a pnp bipolar transistor at its input. The base current of this pnp transistor is approximately 100nA which will cause a 0.1mV change in the 200mV threshold. These are both very minor changes and are well understood. The shift in current can either be neglected or taken into consideration by changing the current sense resistance slightly. CS PIN PROTECTION WITH OVP When designing output overvoltage protection into the switching converter circuit using a zener diode, transient protection on the CS pin requires additional consideration. As shown in Figure 8, adding a zener diode from the output to the CS pin (with the series resistor) for output overvoltage protection will now again allow the transient energy to be passed Adding an additional series resistor to the CS pin as shown in Figure 9 will result in the majority of the transient energy to pass through the sense resistor thereby protecting the LM340X device. www.national.com 16 LM3404/LM3404HV 20205459 FIGURE 8. CS Pin with OVP, Transient Path 20205460 FIGURE 9. CS Pin with OVP, Transient Path with Protection 17 www.national.com LM3404/LM3404HV VIN PIN PROTECTION The VIN pin also has an ESD structure from the pin to GND with a breakdown voltage of approximately 80V. Any transient that exceeds this voltage may damage the device. Although transient absorption is usually present at the front end of a switching converter circuit, damage to the VIN pin can still occur. When VIN is hot swapped in, the current that rushes in to charge CIN up to the VIN value also charges (energizes) the circuit board trace inductance as shown in Figure 10. The excited trace inductance then resonates with the input capacitance (similar to an under-damped LC tank circuit) and causes voltages at the VIN pin to rise well in excess of both VIN and the voltage at the module input connector as clamped by the input TVS. If the resonating voltage at the VIN pin exceeds the 80V breakdown voltage of the ESD structure, the ESD structure will activate and then “snap-back” to a lower voltage due to its inherent design. If this lower snap-back voltage is less than the applied nominal VIN voltage, then significant current will flow through the ESD structure resulting in the IC being damaged. An additional TVS or small zener diode should be placed as close as possible to the VIN pins of each IC on the board, in parallel with the input capacitor as shown in Figure 11. A minor amount of series resistance in the input line would also help, but would lower overall conversion efficiency. For this reason, NTC resistors are often used as inrush limiters instead. 20205461 FIGURE 10. VIN Pin with Typical Input Protection www.national.com 18 LM3404/LM3404HV 20205462 FIGURE 11. VIN Pin with Additional Input Protection GENERAL COMMENTS REGARDING OTHER PINS Any pin that goes “off-board” through a connector should have series resistance of at least 1kΩ to 10kΩ in series with it to protect it from ESD or other transients. These series resistors limit the peak current that can flow (or cause a voltage drop) during a transient event, thus protecting the pin and the device. Pins that are not used should not be left floating. They should instead be tied to GND or to an appropriate voltage through resistance. "warm white" LED module that consists of four LEDs in a 2 x 2 series-parallel configuration. The module will be treated as a two-terminal element and driven with a forward current of 700 mA ±5%. The typical forward voltage of the LED module in thermal steady state is 6.9V, hence the average output voltage will be 7.1V. The objective of this application is to place the complete current regulator and LED module in a compact space formerly occupied by a halogen light source. (The LED will be on a separate metal-core PCB and heatsink.) Switching frequency will be 400 kHz to keep switching loss low, as the confined space with no air-flow requires a maximum temperature rise of 50°C in each circuit component. A small solution size is also important, as the regulator must fit on a circular PCB with a 1.5" diameter. A complete bill of materials can be found in Table 1 at the end of this datasheet. Design Example 1: LM3404 The first example circuit will guide the user through component selection for an architectural accent lighting application. A regulated DC voltage input of 24V ±10% will power a 5.4W 20205419 FIGURE 12. Schematic for Design Example 1 19 www.national.com LM3404/LM3404HV RON and tON A moderate switching frequency is needed in this application to balance the requirements of magnetics size and efficiency. RON is selected from the equation for switching frequency as follows: = 330 mAP-P The peak LED/inductor current is then estimated: IL(PEAK) = IL + 0.5 x ΔiL(MAX) IL(PEAK) = 0.7 + 0.5 x 0.330 = 866 mA In the case of a short circuit across the LED array, the LM3404 will continue to deliver rated current through the short but will reduce the output voltage to equal the CS pin voltage of 200 mV. The inductor ripple current and peak current in this condition would be equal to: ΔiL(LED-SHORT) = [(24 – 0.2) x 7.43 x 10-7] / 38 x 10-6 = 465 mAP-P IL(PEAK) = 0.7 + 0.5 x 0.465 = 933 mA In the case of a short at the switch node, the output, or from the CS pin to ground the short circuit current limit will engage at a typical peak current of 1.5A. In order to prevent inductor saturation during these fault conditions the inductor’s peak current rating must be above 1.5A. A 47 µH off-the shelf inductor rated to 1.4A (peak) and 1.5A (average) with a DCR of 0.1Ω will be used. USING AN OUTPUT CAPACITOR This application does not require high frequency PWM dimming, allowing the use of an output capacitor to reduce the size and cost of the output inductor. To select the proper output capacitor the equation from Buck Regulators with Output Capacitors is re-arranged to yield the following: RON = 7.1 / (1.34 x 10-10 x 4 x 105) = 132.5 kΩ The closest 1% tolerance resistor is 133 kΩ. The switching frequency and on-time of the circuit can then be found using the equations relating RON and tON to fSW: fSW = 7.1 / (1.33 x 105 x 1.34 x 10-10) = 398 kHz tON = (1.34 x 10-10 x 1.33 x 105) / 24 = 743 ns OUTPUT INDUCTOR Since an output capacitor will be used to filter some of the AC ripple current, the inductor ripple current can be set higher than the LED ripple current. A value of 40%P-P is typical in many buck converters: ΔiL = 0.4 x 0.7 = 0.28A With the target ripple current determined the inductance can be chosen: LMIN = [(24 – 7.1) x 7.43 x 10-7] / (0.28) = 44.8 µH The closest standard inductor value is 47 µH. The average current rating should be greater than 700 mA to prevent overheating in the inductor. Separation between the LM3404 drivers and the LED arrays means that heat from the inductor will not threaten the lifetime of the LEDs, but an overheated inductor could still cause the LM3404 to enter thermal shutdown. The inductance of the standard part chosen is ±20%. With this tolerance the typical, minimum, and maximum inductor current ripples can be calculated: ΔiL(TYP) = [(24 - 7.1) x 7.43 x 10-7] / 47 x 10-6 = 266 mAP-P ΔiL(MIN) = [(24 - 7.1) x 7.43 x 10-7] / 56 x 10-6 = 223 mAP-P ΔiL(MAX) = [(24 - 7.1) x 7.43 x 10-7] / 38 x 10-6 The target tolerance for LED ripple current is 100 mAP-P, and a typical value for rD of 1.8Ω at 700 mA can be read from the LED datasheet. The required capacitor impedance to reduce the worst-case inductor ripple current of 333 mAP-P is therefore: ZC = [0.1 / (0.333 - 0.1] x 1.8 = 0.77Ω A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 400 kHz: CO = 1/(2 x π x 0.77 x 4 x 105) = 0.51 µF This calculation assumes that impedance due to the equivalent series resistance (ESR) and equivalent series inductance (ESL) of CO is negligible. The closest 10% tolerance capacitor value is 1.0 µF. The capacitor used should be rated to 25V or more and have an X7R dielectric. Several manufacturers produce ceramic capacitors with these specifications in the 0805 case size. A typical value for ESR of 3 mΩ can be read from the curve of impedance vs. frequency in the product datasheet. www.national.com 20 LM3404/LM3404HV RSNS A preliminary value for RSNS was determined in selecting ΔiL. This value should be re-evaluated based on the calculations for ΔiF: can be calculated as shown in the Design Considerations section. VD for a case size such as SMB in a 40V, 2A Schottky diode at 700 mA is approximately 0.3V and the θJA is 75°C/ W. Power dissipation and temperature rise can be calculated as: PD = 0.509 x 0.3 = 153 mW TRISE = 0.153 x 75 = 11.5°C tSNS = 220 ns, RSNS = 0.33Ω Sub-1Ω resistors are available in both 1% and 5% tolerance. A 1%, 0.33Ω device is the closest value, and a 0.33W, 1206 size device will handle the power dissipation of 162 mW. With the resistance selected, the average value of LED current is re-calculated to ensure that current is within the ±5% tolerance requirement. From the expression for average LED current: IF = 0.2 / 0.33 - (7.1 x 2.2 x 10-7) / 47 x 10-6 + 0.266 / 2 = 706 mA, 1% above 700 mA INPUT CAPACITOR Following the calculations from the Input Capacitor section, ΔvIN(MAX) will be 24V x 2%P-P = 480 mV. The minimum required capacitance is: CIN(MIN) = (0.7 x 7.4 x 10-7) / 0.48 = 1.1 µF To provide additional safety margin the a higher value of 3.3 µF ceramic capacitor rated to 50V with X7R dielectric in an 1210 case size will be used. From the Design Considerations section, input rms current is: IIN-RMS = 0.7 x Sqrt(0.28 x 0.72) = 314 mA CB AND CF The bootstrap capacitor CB should always be a 10 nF ceramic capacitor with X7R dielectric. A 25V rating is appropriate for all application circuits. The linear regulator filter capacitor CF should always be a 100 nF ceramic capacitor, also with X7R dielectric and a 25V rating. EFFICIENCY To estimate the electrical efficiency of this example the power dissipation in each current carrying element can be calculated and summed. Electrical efficiency, η, should not be confused with the optical efficacy of the circuit, which depends upon the LEDs themselves. Total output power, PO, is calculated as: PO = IF x VO = 0.706 x 7.1 = 5W Conduction loss, PC, in the internal MOSFET: PC = (IF2 x RDSON) x D = (0.7062 x 0.8) x 0.28 = 112 mW Gate charging and VCC loss, PG, in the gate drive and linear regulator: PG = (IIN-OP + fSW x QG) x VIN PG = (600 x 10-6 + 4 x 105 x 6 x 10-9) x 24 = 72 mW Switching loss, PS, in the internal MOSFET: Ripple current ratings for 1210 size ceramic capacitors are typically higher than 2A, more than enough for this design. RECIRCULATING DIODE The input voltage of 24V ±5% requires Schottky diodes with a reverse voltage rating greater than 30V. The next highest standard voltage rating is 40V. Selecting a 40V rated diode provides a large safety margin for the ringing of the switch node and also makes cross-referencing of diodes from different vendors easier. The next parameters to be determined are the forward current rating and case size. In this example the low duty cycle (D = 7.1 / 24 = 28%) places a greater thermal stress on D1 than on the internal power MOSFET of the LM3404. The estimated average diode current is: ID = 0.706 x 0.72 = 509 mA A Schottky with a forward current rating of 1A would be adequate, however reducing the power dissipation is critical in this example. Higher current diodes have lower forward voltages, hence a 2A-rated diode will be used. To determine the proper case size, the dissipation and temperature rise in D1 PS = 0.5 x VIN x IF x (tR + tF) x fSW PS = 0.5 x 24 x 0.706 x 40 x 10-9 x 4 x 105 = 136 mW AC rms current loss, PCIN, in the input capacitor: PCIN = IIN(rms)2 x ESR = 0.3172 0.003 = 0.3 mW (negligible) DCR loss, PL, in the inductor PL = IF2 x DCR = 0.7062 x 0.1 = 50 mW Recirculating diode loss, PD = 153 mW Current Sense Resistor Loss, PSNS = 164 mW Electrical efficiency, η = PO / (PO + Sum of all loss terms) = 5 / (5 + 0.687) = 88% Temperature Rise in the LM3404 IC is calculated as: TLM3404 = (PC + PG + PS) x θJA = (0.112 + 0.072 + 0.136) x 155 = 49.2°C 21 www.national.com LM3404/LM3404HV Design Example 2: LM3404HV The second example circuit will guide the user through component selection for an outdoor general lighting application. A regulated DC voltage input of 48V ±10% will power ten se- ries-connected LEDs at 500 mA ±10% with a ripple current of 50 mAP-P or less. The typical forward voltage of the LED module in thermal steady state is 35V, hence the average output voltage will be 35.2V. A complete bill of materials can be found in Table 2 at the end of this datasheet. 20205432 FIGURE 13. Schematic for Design Example 2 RON and tON A low switching frequency, 225 kHz, is needed in this application, as high efficiency and low power dissipation take precedence over the solution size. RON is selected from the equation for switching frequency as follows: LMIN = [(48 – 35.2) x 3.3 x 10-6] / (0.15) = 281 µH The closest standard inductor value above 281 is 330 µH. The average current rating should be greater than 0.5A to prevent overheating in the inductor. In this example the LM3404HV driver and the LED array share the same metal-core PCB, meaning that heat from the inductor could threaten the lifetime of the LEDs. For this reason the average current rating of the inductor used should have a de-rating of about 50%, or 1A. The inductance of the standard part chosen is ±20%. With this tolerance the typical, minimum, and maximum inductor current ripples can be calculated: ΔiL(TYP) = [(48 - 35.2) x 3.3 x 10-6] / 330 x 10-6 = 128 mAP-P ΔiL(MIN) = [(48 - 35.2) x 3.3 x 10-6] / 396 x 10-6 = 107 mAP-P ΔiL(MAX) = [(48 - 35.2) x 3.3 x 10-6] / 264 x 10-6 = 160 mAP-P The peak inductor current is then estimated: IL(PEAK) = IL + 0.5 x ΔiL(MAX) IL(PEAK) = 0.5 + 0.5 x 0.16 = 0.58A In the case of a short circuit across the LED array, the LM3404HV will continue to deliver rated current through the short but will reduce the output voltage to equal the CS pin RON = 35.2 / (1.34 x 10-10 x 2.25 x 105) = 1.16 MΩ The next highest 1% tolerance resistor is 1.18 MΩ. The switching frequency and on-time of the circuit can then be found using the equations relating RON and tON to fSW: fSW = 35.2 / (1.18 x 106 x 1.34 x 10-10) = 223 kHz tON = (1.34 x 10-10 x 1.18 x 106) / 48 = 3.3 µs OUTPUT INDUCTOR Since an output capacitor will be used to filter some of the AC ripple current, the inductor ripple current can be set higher than the LED ripple current. A value of 30%P-P makes a good trade-off between the current ripple and the size of the inductor: ΔiL = 0.3 x 0.5 = 0.15A With the target ripple current determined the inductance can be chosen: www.national.com 22 LM3404/LM3404HV voltage of 200 mV. The inductor ripple current and peak current in this condition would be equal to: ΔiL(LED-SHORT) = [(48 – 0.2) x 3.3 x 10-6] / 264 x 10-6 = 0.598AP-P IL(PEAK) = 0.5 + 0.5 x 0.598 = 0.8A In the case of a short at the switch node, the output, or from the CS pin to ground the short circuit current limit will engage at a typical peak current of 1.5A. In order to prevent inductor saturation during these fault conditions the inductor’s peak current rating must be above 1.5A. A 330 µH off-the shelf inductor rated to 1.9A (peak) and 1.0A (average) with a DCR of 0.56Ω will be used. USING AN OUTPUT CAPACITOR This application uses sub-1 kHz frequency PWM dimming, allowing the use of a small output capacitor to reduce the size and cost of the output inductor. To select the proper output capacitor the equation from Buck Regulators with Output Capacitors is re-arranged to yield the following: tSNS = 220 ns, RSNS = 0.43Ω Sub-1Ω resistors are available in both 1% and 5% tolerance. A 1%, 0.43Ω device is the closest value, and a 0.25W, 0805 size device will handle the power dissipation of 110 mW. With the resistance selected, the average value of LED current is re-calculated to ensure that current is within the ±10% tolerance requirement. From the expression for average LED current: IF = 0.2 / 0.43 - (35.2 x 2.2 x 10-7) / 330 x 10-6 + 0.128 / 2 = 505 mA INPUT CAPACITOR Following the calculations from the Input Capacitor section, ΔvIN(MAX) will be 48V x 2%P-P = 960 mV. The minimum required capacitance is: CIN(MIN) = (0.5 x 3.3 x 10-6) / 0.96 = 1.7 µF To provide additional safety margin a 2.2 µF ceramic capacitor rated to 100V with X7R dielectric in an 1812 case size will be used. From the Design Considerations section, input rms current is: The target tolerance for LED ripple current is 50 mAP-P, and the typical value for rD is 10Ω with ten LEDs in series. The required capacitor impedance to reduce the worst-case steady-state inductor ripple current of 160 mAP-P is therefore: ZC = [0.05 / (0.16 - 0.05] x 10 = 4.5Ω A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 223 kHz: CO = 1/(2 x π x 4.5 x 2.23 x 105) = 0.16 µF This calculation assumes that impedance due to the equivalent series resistance (ESR) and equivalent series inductance (ESL) of CO is negligible. The closest 10% tolerance capacitor value is 0.15 µF. The capacitor used should be rated to 50V or more and have an X7R dielectric. Several manufacturers produce ceramic capacitors with these specifications in the 0805 case size. ESR values are not typically provided for such low value capacitors, however is can be assumed to be under 100 mΩ, leaving plenty of margin to meet to LED ripple current requirement. The low capacitance required allows the use of a 100V rated, 1206-size capacitor. The rating of 100V ensures that the capacitance will not decrease significantly when the DC output voltage is applied across the capacitor. RSNS A preliminary value for RSNS was determined in selecting ΔiL. This value should be re-evaluated based on the calculations for ΔiF: IIN-RMS = 0.5 x Sqrt(0.73 x 0.27) = 222 mA Ripple current ratings for 1812 size ceramic capacitors are typically higher than 2A, more than enough for this design, and the ESR is approximately 3 mΩ. RECIRCULATING DIODE The input voltage of 48V requires Schottky diodes with a reverse voltage rating greater than 50V. The next highest standard voltage rating is 60V. Selecting a 60V rated diode provides a large safety margin for the ringing of the switch node and also makes cross-referencing of diodes from different vendors easier. The next parameters to be determined are the forward current rating and case size. In this example the high duty cycle (D = 35.2 / 48 = 73%) places a greater thermal stress on the internal power MOSFET than on D1. The estimated average diode current is: ID = 0.5 x 0.27 = 135 mA A Schottky with a forward current rating of 0.5A would be adequate, however reducing the power dissipation is critical in this example. Higher current diodes have lower forward voltages, hence a 1A-rated diode will be used. To determine the proper case size, the dissipation and temperature rise in D1 can be calculated as shown in the Design Considerations section. VD for a case size such as SMA in a 60V, 1A Schottky diode at 0.5A is approximately 0.35V and the θJA is 75°C/W. Power dissipation and temperature rise can be calculated as: PD = 0.135 x 0.35 = 47 mW TRISE = 0.047 x 75 = 3.5°C 23 www.national.com LM3404/LM3404HV CB AND CF The bootstrap capacitor CB should always be a 10 nF ceramic capacitor with X7R dielectric. A 25V rating is appropriate for all application circuits. The linear regulator filter capacitor CF should always be a 100 nF ceramic capacitor, also with X7R dielectric and a 25V rating. EFFICIENCY To estimate the electrical efficiency of this example the power dissipation in each current carrying element can be calculated and summed. Electrical efficiency, η, should not be confused with the optical efficacy of the circuit, which depends upon the LEDs themselves. Total output power, PO, is calculated as: PO = IF x VO = 0.5 x 35.2 = 17.6W Conduction loss, PC, in the internal MOSFET: PC = (IF2 x RDSON) x D = (0.52 x 0.8) x 0.73 = 146 mW PCIN = IIN(rms)2 x ESR = 0.2222 0.003 = 0.1 mW (negligible) DCR loss, PL, in the inductor PL = IF2 x DCR = 0.52 x 0.56 = 140 mW Recirculating diode loss, PD = 47 mW Current Sense Resistor Loss, PSNS = 110 mW Electrical efficiency, η = PO / (PO + Sum of all loss terms) = 17.6 / (17.6 + 0.644) = 96% Temperature Rise in the LM3404HV IC is calculated as: TLM3404 = (PC + PG + PS) x θJA = (0.146 + 0.094 + 0.107) x 155 = 54°C Layout Considerations The performance of any switching converter depends as much upon the layout of the PCB as the component selection. The following guidelines will help the user design a circuit with maximum rejection of outside EMI and minimum generation of unwanted EMI. COMPACT LAYOUT Parasitic inductance can be reduced by keeping the power path components close together and keeping the area of the loops that high currents travel small. Short, thick traces or copper pours (shapes) are best. In particular, the switch node (where L1, D1, and the SW pin connect) should be just large enough to connect all three components without excessive heating from the current it carries. The LM3404/04HV operates in two distinct cycles whose high current paths are shown in Figure 14: Gate charging and VCC loss, PG, in the gate drive and linear regulator: PG = (IIN-OP + fSW x QG) x VIN PG = (600 x 10-6 + 2.23 x 105 x 6 x 10-9) x 48 = 94 mW Switching loss, PS, in the internal MOSFET: PS = 0.5 x VIN x IF x (tR + tF) x fSW PS = 0.5 x 48 x 0.5 x 40 x 10-9 x 2.23 x 105 = 107 mW AC rms current loss, PCIN, in the input capacitor: 20205428 FIGURE 14. Buck Converter Current Loops The dark grey, inner loop represents the high current path during the MOSFET on-time. The light grey, outer loop represents the high current path during the off-time. GROUND PLANE AND SHAPE ROUTING The diagram of Figure 14 is also useful for analyzing the flow of continuous current vs. the flow of pulsating currents. The circuit paths with current flow during both the on-time and offtime are considered to be continuous current, while those that carry current during the on-time or off-time only are pulsating currents. Preference in routing should be given to the pulsating current paths, as these are the portions of the circuit most likely to emit EMI. The ground plane of a PCB is a conductor and return path, and it is susceptible to noise injection just as any other circuit path. The continuous current paths on the ground net can be routed on the system ground plane with less risk of injecting noise into other circuits. The path between the input source and the input capacitor and the path between the recirculating diode and the LEDs/current sense resistor are examples of continuous current paths. In contrast, the path between the recirculating diode and the input capacitor carries a large pulsating current. This path should be www.national.com 24 LM3404/LM3404HV routed with a short, thick shape, preferably on the component side of the PCB. Multiple vias in parallel should be used right at the pad of the input capacitor to connect the component side shapes to the ground plane. A second pulsating current loop that is often ignored is the gate drive loop formed by the SW and BOOT pins and capacitor CB. To minimize this loop at the EMI it generates, keep CB close to the SW and BOOT pins. CURRENT SENSING The CS pin is a high-impedance input, and the loop created by RSNS, RZ (if used), the CS pin and ground should be made as small as possible to maximize noise rejection. RSNS should therefore be placed as close as possible to the CS and GND pins of the IC. REMOTE LED ARRAYS In some applications the LED or LED array can be far away (several inches or more) from the LM3404/04HV, or on a separate PCB connected by a wiring harness. When an output capacitor is used and the LED array is large or separated from the rest of the converter, the output capacitor should be placed close to the LEDs to reduce the effects of parasitic inductance on the AC impedance of the capacitor. The current sense resistor should remain on the same PCB, close to the LM3404/04HV. 25 www.national.com LM3404/LM3404HV TABLE 1. BOM for Design Example 1 ID U1 L1 D1 Cf Cb Cin Co Rsns Ron Part Number LM3404 SLF10145T-470M1R4 CMSH2-40 VJ0805Y104KXXAT VJ0805Y103KXXAT C3225X7R1H335M C2012X7R1E105M ERJ8BQFR33V CRCW08051333F Type LED Driver Inductor Schottky Diode Capacitor Capacitor Capacitor Capacitor Resistor Resistor Size SO-8 10 x 10 x 4.5mm SMB 0805 0805 1210 0805 1206 0805 Parameters 42V, 1.2A 47 µH, 1.4A, 120 mΩ 40V, 2A 100 nF 10% 10 nF 10% 3.3 µF, 50V 1.0 µF, 25V 0.33Ω 1% 133 kΩ 1% Qty 1 1 1 1 1 1 1 1 1 Vendor NSC TDK Central Semi Vishay Vishay TDK TDK Panasonic Vishay TABLE 2. BOM for Design Example 2 ID U1 L1 D1 Cf Cb Cin Co Rsns Ron Part Number LM3404HV DO5022P-334 CMSH1-60M VJ0805Y104KXXAT VJ0805Y103KXXAT C4532X7R2A225M C3216X7R2A154M ERJ6BQFR43V CRCW08051184F Type LED Driver Inductor Schottky Diode Capacitor Capacitor Capacitor Capacitor Resistor Resistor Size SO-8 18.5 x 15.4 x 7.1mm SMA 0805 0805 1812 1206 0805 0805 Parameters 75V, 1.2A 330 µH, 1.9A, 0.56Ω 60V, 1A 100 nF 10% 10 nF 10% 2.2 µF, 100V 0.15 µF, 100V 0.43Ω 1% 1.18 MΩ 1% Qty 1 1 1 1 1 1 1 1 1 Vendor NSC Coilcraft Central Semi Vishay Vishay TDK TDK Panasonic Vishay www.national.com 26 LM3404/LM3404HV Physical Dimensions inches (millimeters) unless otherwise noted SO-8 Package NS Package Number M08A PSOP-8 Package NS Package Number MRA08B 27 www.national.com LM3404/04HV 1.0A Constant Current Buck Regulator for Driving High Power LEDs Notes For more National Semiconductor product information and proven design tools, visit the following Web sites at: www.national.com Products Amplifiers Audio Clock and Timing Data Converters Interface LVDS Power Management Switching Regulators LDOs LED Lighting Voltage References PowerWise® Solutions Temperature Sensors PLL/VCO www.national.com/amplifiers www.national.com/audio www.national.com/timing www.national.com/adc www.national.com/interface www.national.com/lvds www.national.com/power www.national.com/switchers www.national.com/ldo www.national.com/led www.national.com/vref www.national.com/powerwise WEBENCH® Tools App Notes Reference Designs Samples Eval Boards Packaging Green Compliance Distributors Quality and Reliability Feedback/Support Design Made Easy Design Support www.national.com/webench www.national.com/appnotes www.national.com/refdesigns www.national.com/samples www.national.com/evalboards www.national.com/packaging www.national.com/quality/green www.national.com/contacts www.national.com/quality www.national.com/feedback www.national.com/easy www.national.com/solutions www.national.com/milaero www.national.com/solarmagic www.national.com/training Applications & Markets Mil/Aero PowerWise® Design University Serial Digital Interface (SDI) www.national.com/sdi www.national.com/wireless www.national.com/tempsensors SolarMagic™ THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. 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