LM3478MM

LM3478/LM3478Q High Efficiency Low-Side N-Channel Controller for Switching Regulator March 30, 2009 LM3478/LM3478Q High Efficiency Low-Side N-Channel Controller for Switching Regulator General Description The LM3478 is a versatile Low-Side N-Channel MOSFET controller for switching regulators. It is suitable for use in topologies requiring a low side MOSFET, such as boost, flyback, SEPIC, etc. Moreover, the LM3478 can be operated at extremely high switching frequency in order to reduce the overall solution size. The switching frequency of the LM3478 can be adjusted to any value between 100kHz and 1MHz by using a single external resistor. Current mode control provides superior bandwidth and transient response, besides cycle-by-cycle current limiting. Output current can be programmed with a single external resistor. The LM3478 has built in features such as thermal shutdown, short-circuit protection, over voltage protection, etc. Power saving shutdown mode reduces the total supply current to 5µA and allows power supply sequencing. Internal soft-start limits the inrush current at start-up. Features ■ LM3478Q is AEC-Q100 qualified and manufactured on an Automotive Grade Flow 8-lead Mini-SO8 (MSOP-8) package Internal push-pull driver with 1A peak current capability Current limit and thermal shutdown Frequency compensation optimized with a capacitor and a resistor ■ Internal softstart ■ Current Mode Operation ■ Undervoltage Lockout with hysteresis ■ ■ ■ ■ Applications ■ ■ ■ ■ ■ Distributed Power Systems Battery Chargers Offline Power Supplies Telecom Power Supplies Automotive Power Systems Key Specifications ■ ■ ■ ■ Wide supply voltage range of 2.97V to 40V 100kHz to 1MHz Adjustable clock frequency ±2.5% (over temperature) internal reference 10µA shutdown current (over temperature) Typical Application Circuit 10135501 Typical High Efficiency Step-Up (Boost) Converter © 2009 National Semiconductor Corporation 101355 www.national.com LM3478/LM3478Q Connection Diagram 10135502 8 Lead Mini SO8 Package (MSOP-8 Package) Package Marking and Ordering Information Order Number LM3478MM LM3478MMX LM3478QMM LM3478QMMX Package Type MSOP-8 MSOP-8 Package Marking S14B SSFB Supplied As: 1000 units on Tape and Reel 3500 units on Tape and Reel 1000 units on Tape and Reel 3500 units on Tape and Reel AEC-Q100 (Grade 1) qualified. Automotive Grade Production Flow* Feature * Automotive Grade (Q) product incorporates enhanced manufacturing and support processes for the automotive market, including defect detection methodologies. Reliability qualification is compliant with the requirements and temperature grades defined in the AEC-Q100 standard. Automotive grade products are identified with the letter Q. For more information go to http://www.national.com/automotive. Pin Descriptions Pin Name ISEN COMP FB AGND PGND DR FA/SD Pin Number 1 2 3 4 5 6 7 Description Current sense input pin. Voltage generated across an external sense resistor is fed into this pin. Compensation pin. A resistor, capacitor combination connected to this pin provides compensation for the control loop. Feedback pin. The output voltage should be adjusted using a resistor divider to provide 1.26V at this pin. Analog ground pin. Power ground pin. Drive pin. The gate of the external MOSFET should be connected to this pin. Frequency adjust and Shutdown pin. A resistor connected to this pin sets the oscillator frequency. A high level on this pin for longer than 30 µs will turn the device off. The device will then draw less than 10µA from the supply. Power Supply Input pin. VIN 8 www.national.com 2 LM3478/LM3478Q Absolute Maximum Ratings (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. Input Voltage FB Pin Voltage FA/SD Pin Voltage Peak Driver Output Current (0.5 and Compensation Ramp to Avoid Sub-Harmonic Oscillation 11 www.national.com LM3478/LM3478Q Sub-harmonic Oscillation can be easily understood as a geometric problem. If the control signal does not have slope, the slope representing the inductor current ramps up until the control signal is reached and then slopes down again. If the duty cycle is above 50%, any perturbation will not converge but diverge from cycle to cycle and causes sub-harmonic oscillation. It is apparent that the difference in the inductor current from one cycle to the next is a function of Sn, Sf and Se as follows: It is a good design practice to only add as much slope compensation as needed to avoid subharmonic oscillation. Additional slope compensation minimizes the influence of the sensed current in the control loop. With very large slope compensation the control loop characteristics are similar to a voltage mode regulator which compares the error voltage to a saw tooth waveform rather than the inductor current. Hence, if the quantity (Sf - Se)/(Sn + Se) is greater than 1, the inductor current diverges and subharmonic oscillation results. This counts for all current mode topologies. The LM3478 has some internal slope compensation VSL which is enough for many applications above 50% duty cycle to avoid subharmonic oscillation . For boost applications, the slopes Se, Sf and Sn can be calculated with the formulas below: Se = VSL x fs Sf = (VOUT - VIN)/L Sn = VIN/L When Se increases then the factor which determines if subharmonic oscillation will occur decreases. When the duty cycle is greater than 50%, and the inductance becomes less, the factor increases. For more flexibility slope compensation can be increased by adding one external resistor, RSL, in the Isens path. Figure 4 shows the setup. The externally generated slope compensation is then added to the internal slope compensation of the LM3478. When using external slope compensation, the formula for Se becomes: Se = (VSL + (K x RSL)) x fs A typical value for factor K is 40 µA. The factor changes with switching frequency. Figure 5 is used to determine the factor K for individual applications and the formula below gives the factor K. K = ΔVSL / RSL 10135513 FIGURE 4. Adding External Slope Compensation 10135595 FIGURE 5. External Slope Compensation ΔVSL vs RSL FREQUENCY ADJUST/SHUTDOWN The switching frequency of the LM3478 can be adjusted between 100kHz and 1MHz using a single external resistor. This resistor must be connected between FA/SD pin and ground, as shown in Figure 6. To determine the value of the resistor required for a desired switching frequency refer to the typical performance characteristics. www.national.com 12 LM3478/LM3478Q The FA/SD pin also functions as a shutdown pin. If a high signal (>1.35V) appears on the FA/SD pin, the LM3478 stops switching and goes into a low current mode. The total supply current of the IC reduces to less than 10 uA under these conditions. Figure 7 shows implementation of the shutdown function when operating in frequency adjust mode. In this mode a high signal for more than 30us shuts down the IC. However, the voltage on the FA/SD pin should be always less than the absolute maximum of 7V to avoid any damage to the device. 10135514 FIGURE 6. Frequency Adjust 10135516 FIGURE 7. Shutdown Operation in Frequency Adjust Mode SHORT-CIRCUIT PROTECTION When the voltage across the sense resistor measured on the Isen pin exceeds 343 mV, short circuit current limit protection gets activated. A comparator inside the LM3478 reduces the switching frequency by a factor of 5 and maintains this condition until the short is removed. In normal operation the sensed current will trigger the power MOSFET to turn off. During the blanking interval the PWM comparator will not react to an over current so that this additional 343 mV current limit threshold is implemented to protect the device in a short circuit or severe overload condition. Typical Applications The LM3478 may be operated in either the continuous (CCM) or the discontinuous current conduction mode (DCM). The following applications are designed for the CCM operation. This mode of operation has higher efficiency and usually lower EMI characteristics than the DCM. BOOST CONVERTER The boost converter converts a low input voltage into a higher output voltage. The basic configuration for a boost converter is shown in Figure 8. In the CCM (when the inductor current never reaches zero at steady state), the boost regulator operates in two states. In the first state of operation, MOSFET Q is turned on and energy is stored in the inductor. During this state, diode D is reverse biased and load current is supplied by the output capacitor, Cout. In the second state, MOSFET Q is off and the diode is forward biased. The energy stored in the inductor is transferred to the load and the output capacitor. The ratio of the switch on time to the total period is the duty cycle D: D = 1 - (Vin / Vout) Including the voltage drop across the MOSFET and the diode the definition for the duty cycle is: D = 1 - ((Vin - Vq)/(Vout + Vd)) Vd is the forward voltage drop of the diode and Vq is the voltage drop across the MOSFET when it is on. 13 www.national.com LM3478/LM3478Q 10135522 FIGURE 8. Simplified Boost Convertert A. First Cycle Operation B. Second Cycle of Operation POWER INDUCTOR SELECTION The inductor is one of the two energy storage elements in a boost converter. Figure 9 shows how the inductor current varies during a switching cycle. The current through an inductor is quantified by the following relationship of L, IL and VL : The important quantities in determining a proper inductance value are IL (the average inductor current) and ΔIL (the inductor current ripple). If ΔIL is larger than IL, the inductor current will drop to zero for a portion of the cycle and the converter will operate in the DCM. All the analysis in this datasheet assumes operation in the CCM. To operate in the CCM, the following condition must be met: Choose the minimum Iout to determine the minimum inductance value. A common choice is to set ΔIL to 30% of IL. Choosing an appropriate core size for the inductor involves calculating the average and peak currents expected through the inductor. In a boost converter the peak inductor current is: ILPEAK = Average IL(max) + ΔIL(max) Average IL(max) = Iout / (1-D) ΔIL(max) = D x Vin / (2 x fs x L) An inductor size with ratings higher than these values has to be selected. If the inductor is not properly rated, saturation will occur and may cause the circuit to malfunction. The LM3478 can be set to switch at very high frequencies. When the switching frequency is high, the converter can be operated with very small inductor values. The LM3478 senses the peak current through the switch which is the same as the peak inductor current as calculated above. 10135524 FIGURE 9. Inductor Current and Diode Current www.national.com 14 LM3478/LM3478Q PROGRAMMING THE OUTPUT VOLTAGE AND OUTPUT CURRENT The output voltage can be programmed using a resistor divider between the output and the FB pin. The resistors are selected such that the voltage at the FB pin is 1.26V. Pick RF1 (the resistor between the output voltage and the feedback pin) and RF2 (the resistor between the feedback pin and ground) can be selected using the following equation, RF2 = (1.26V x RF1) / (Vout - 1.26V) A 100pF capacitor may be connected between the feedback and ground pins to reduce noise. CURRENT LIMIT WITH ONLY THE INTERNAL SLOPE COMPENSATION The maximum amount of current that can be delivered at the output is controlled by the sense resistor, R SEN. Current limit occurs when the voltage that is generated across the sense resistor equals the current sense threshold voltage, VSENSE. Limits for VSENSE have been specified in the electrical characteristics section. This can be expressed as: VSENSE = ISW(PEAK) x RSEN VSENSE represents the maximum value of the control signal VCS as shown in Figure 10. This control signal, however, is not a constant value and changes over the course of a period as a result of the internal compensation ramp. Therefore the current limit will also change as a result of the internal compensation ramp. The actual VCS can be better expressed as a function of the sense voltage (VSENSE) and the internal compensation ramp: VCS = VSENSE − (D x VSL) The expression for RSEN is: The numerator of the above equation is the VCS and the denominator is the peak current. CURRENT LIMIT WITH THE INTERNAL SLOPE COMPENSATION PLUS ADDITIONAL EXTERNAL SLOPE COMPENSATION If an external slope compensation resistor RSL is used, the internal command signal will be modified and this will have an effect on the current limit. The command signal then includes the external slope ΔVSL: VCS = VSENSE – (D x (VSL + ΔVSL)) Where ΔVSL is the additional slope compensation generated as discussed in the section slope compensation ramp. This changes the equation for RSEN to: Note that RSL which defines ΔVSL is an additional way of setting the current limit. In some designs RSL can also help to filter noise to keep the ISEN pin quiet. POWER DIODE SELECTION Observation of the boost converter circuit shows that the average current through the diode is the average load current, and the peak current through the diode is the peak current through the inductor. The diode should be rated to handle more than its peak current. The peak diode current can be calculated using the formula: ID(Peak) = IOUT/ (1−D) + ΔIL Thermally the diode must be able to handle the maximum average current delivered to the output. The peak reverse voltage for boost converters is equal to the regulated output voltage. The diode must be capable of handling this voltage. To improve efficiency, a low forward drop schottky diode is recommended. 10135552 FIGURE 10. Current Sense Voltage vs Duty Cycle Figure 10 shows how VCS changes with duty cycle. The curve shows the ramp Se which is defined by the voltage VSENSE (at 0% dutycycle) and by the internally generated slope VSL which changes VCS with duty cycle. The dotted line shows VSENSE. At 100% duty cycle, the current sense voltage will be VSENSE minus VSL. The graph also shows the increased current limit of 343 mV (typical) during the 325 ns (typical) blank out time. For different frequencies this fixed blank out time obviously occupies more duty cycle, percentage wise. The peak current through the switch is equal to the peak inductor current. POWER MOSFET SELECTION The drive pin of the LM3478 must be connected to the gate of an external MOSFET. The drive pin (DR) voltage depends on the input voltage (see typical performance characteristics). In most applications, a logic level MOSFET can be used. For very low input voltages, a sub logic level MOSFET should be used. The selected MOSFET has a great influence on the system efficiency. The critical parameters for selecting a MOSFET are: 1. Minimum threshold voltage, VTH(MIN) 2. On-resistance, RDS(ON) 3. Total gate charge, Qg 4. Reverse transfer capacitance, CRSS 5. Maximum drain to source voltage, VDS(MAX) 15 www.national.com LM3478/LM3478Q The off-state voltage of the MOSFET is approximately equal to the output voltage. Vds(max) must be greater than the output voltage. The power losses in the MOSFET can be categorized into conduction losses and switching losses. Rds(on) is needed to estimate the conduction losses, Pcond: Pcond = I2 x RDS(ON) x D x fS The temperature effect on the RDS(ON) usually is quite significant. Assume 30% increase at hot. For the current I in the formula above the average inductor current may be used. Especially at high switching frequencies the switching losses may be the largest portion of the total losses. The switching losses are very difficult to calculate due to changing parasitics of a given MOSFET in operation. Often the individual MOSFETS datasheet does not give enough information to yield a useful result. The following formulas give a rough idea how the switching losses are calculated: OUTPUT CAPACITOR SELECTION The output capacitor in a boost converter provides all the output current when the inductor is charging. As a result it sees very large ripple currents. The output capacitor should be capable of handling the maximum rms current. The rms current in the output capacitor is: Where INPUT CAPACITOR SELECTION Due to the presence of an inductor at the input of a boost converter, the input current waveform is continuous and triangular as shown in figure 9. The inductor ensures that the input capacitor sees fairly low ripple currents. However, as the input capacitor gets smaller, the input ripple goes up. The rms current in the input capacitor is given by: The ESR and ESL of the capacitor directly control the output ripple. Use capacitors with low ESR and ESL at the output for high efficiency and low ripple voltage. Surface mount tantalums, surface mount polymer electrolytic, polymer tantalum, or multi-layer ceramic capacitors are recommended at the output. For applications that require very low output voltage ripple, a second stage LC filter often is a good solution. Most of the time it is lower cost to use a small second Inductor in the power path and an additional final output capacitor than to reduce the output voltage ripple by purely increasing the output capacitor without an additional LC filter. LAYOUT GUIDELINES Good board layout is critical for switching controllers. First the ground plane area must be sufficient for thermal dissipation purposes and second, appropriate guidelines must be followed to reduce the effects of switching noise. Switching converters are very fast switching devices. In such devices, the rapid increase of input current combined with the parasitic trace inductance generates unwanted Ldi/dt noise spikes. The magnitude of this noise tends to increase as the output current increases. This parasitic spike noise may turn into electromagnetic interference (EMI), and can also cause problems in device performance. Therefore, care must be taken in layout to minimize the effect of this switching noise. The current sensing circuit in current mode devices can be easily affected by switching noise. This noise can cause duty cycle jittering which leads to increased spectral noise. Although the LM3478 has 325ns blanking time at the beginning of every cycle to ignore this noise, some noise may remain after the blanking time. The most important layout rule is to keep the AC current loops as small as possible. Figure 12 shows the current flow of a boost converter. The top schematic shows a dotted line which represents the current flow during on-state and the middle schematic shows the current flow during off-state. The bottom schematic shows the currents we refer to as AC currents. They are the most critical ones since current is changing in very short time periods. The dotted lined traces of the bottom schematic are the once to make as short as possible. The input capacitor should be capable of handling the rms current. Although the input capacitor is not as critical in a boost application, low values can cause impedance interactions. Therefore a good quality capacitor should be chosen in the range of 100µF to 200µF. If a value lower than 100µF is used, then problems with impedance interactions or switching noise can affect the LM3478. To improve performance, especially with Vin below 8 volts, it is recommended to use a 20 Ohm resistor at the input to provide an RC filter. The resistor is placed in series with the VIN pin with only a bypass capacitor attached to the VIN pin directly (see figure 11). A 0.1µF or 1µF ceramic capacitor is necessary in this configuration. The bulk input capacitor and inductor will connect on the other side of the resistor at the input power supply. 10135593 FIGURE 11. Reducing IC Input Noise www.national.com 16 LM3478/LM3478Q Designing SEPIC Using the LM3478 Since the LM3478 controls a low-side N-Channel MOSFET, it can also be used in SEPIC (Single Ended Primary Inductance Converter) applications. An example of a SEPIC using the LM3478 is shown in Figure 13. Note that the output voltage can be higher or lower than the input voltage. The SEPIC uses two inductors to step-up or step-down the input voltage. The inductors L1 and L2 can be two discrete inductors or two windings of a coupled inductor since equal voltages are applied across the inductor throughout the switching cycle. Using two discrete inductors allows use of catalog magnetics, as opposed to a custom inductor. The input ripple can be reduced along with size by using the coupled windings for L1 and L2. Due to the presence of the inductor L1 at the input, the SEPIC inherits all the benefits of a boost converter. One main advantage of a SEPIC over a boost converter is the inherent input to output isolation. The capacitor CS isolates the input from the output and provides protection against a shorted or malfunctioning load. Hence, the SEPIC is useful for replacing boost circuits when true shutdown is required. This means that the output voltage falls to 0V when the switch is turned off. In a boost converter, the output can only fall to the input voltage minus a diode drop. The duty cycle of a SEPIC is given by: 10135520 FIGURE 12. Current Flow In A Boost Application The PGND and AGND pins have to be connected to the same ground very close to the IC. To avoid ground loop currents, attach all the grounds of the system only at one point. A ceramic input capacitor should be connected as close as possible to the Vin pin and grounded close to the GND pin. For a layout example please see Application Note1204. For more information about layout in switch mode power supplies please refer to Application Note 1229. COMPENSATION For detailed explanation on how to select the right compensation components to attach to the compensation pin for a boost topology please see Application Note 1286. In the above equation, VQ is the on-state voltage of the MOSFET, Q, and VDIODE is the forward voltage drop of the diode. 10135544 FIGURE 13. Typical SEPIC Converter POWER MOSFET SELECTION As in a boost converter, parameters governing the selection of the MOSFET are the minimum threshold voltage, VTH (MIN), the on-resistance, RDS(ON), the total gate charge, Qg, the reverse transfer capacitance, CRSS, and the maximum drain to source voltage, VDS(MAX). The peak switch voltage in a SEPIC is given by: VSW(PEAK) = VIN + VOUT + VDIODE The selected MOSFET should satisfy the condition: VDS(MAX) > VSW(PEAK) 17 www.national.com LM3478/LM3478Q The peak switch current is given by: The rms current through the switch is given by: IL1PK must be lower than the maximum current rating set by the current sense resistor. The value of L1 can be increased above the minimum recommended to reduce input ripple and output ripple. However, once DIL1 is less than 20% of IL1AVE, the benefit to output ripple is minimal. By increasing the value of L2 above the minimum recommended, ΔIL2 can be reduced, which in turn will reduce the output ripple voltage: POWER DIODE SELECTION The Power diode must be selected to handle the peak current and the peak reverse voltage. In a SEPIC, the diode peak current is the same as the switch peak current. The off-state voltage or peak reverse voltage of the diode is VIN + VOUT. Similar to the boost converter, the average diode current is equal to the output current. Schottky diodes are recommended. SELECTION OF INDUCTORS L1 AND L2 Proper selection of inductors L1 and L2 to maintain continuous current conduction mode requires calculations of the following parameters. Average current in the inductors: where ESR is the effective series resistance of the output capacitor. If L1 and L2 are wound on the same core, then L1 = L2 = L. All the equations above will hold true if the inductance is replaced by 2L. SENSE RESISTOR SELECTION The peak current through the switch, ISW(PEAK) can be adjusted using the current sense resistor, RSEN, to provide a certain output current. Resistor RSEN can be selected using the formula: Sepic Capacitor Selection IL2AVE = IOUT Peak to peak ripple current, to calculate core loss if necessary: The selection of the SEPIC capacitor, CS, depends on the rms current. The rms current of the SEPIC capacitor is given by: Maintaining the condition IL > ΔiL/2 to ensure continuous current conduction yields: The SEPIC capacitor must be rated for a large ACrms current relative to the output power. This property makes the SEPIC much better suited to lower power applications where the rms current through the capacitor is relatively small (relative to capacitor technology). The voltage rating of the SEPIC capacitor must be greater than the maximum input voltage. There is an energy balance between CS and L1, which can be used to determine the value of the capacitor. The basic energy balance equation is: where Peak current in the inductor, to ensure the inductor does not saturate: is the ripple voltage across the SEPIC capacitor, and www.national.com 18 LM3478/LM3478Q is the ripple current through the inductor L1. The energy balance equation can be solved to provide a minimum value for CS: boost application, low values can cause impedance interactions. Therefore a good quality capacitor should be chosen in the range of 100µF to 200µF. If a value lower than 100µF is used than problems with impedance interactions or switching noise can affect the LM3478. To improve performance, especially with VIN below 8 volts, it is recommended to use a 20Ω resistor at the input to provide a RC filter. The resistor is placed in series with the VIN pin with only a bypass capacitor attached to the VIN pin directly (see Figure 11). A 0.1µF or 1µF ceramic capacitor is necessary in this configuration. The bulk input capacitor and inductor will connect on the other side of the resistor with the input power supply. Input Capacitor Selection Similar to a boost converter, the SEPIC has an inductor at the input. Hence, the input current waveform is continuous and triangular. The inductor ensures that the input capacitor sees fairly low ripple currents. However, as the input capacitor gets smaller, the input ripple goes up. The rms current in the input capacitor is given by: Output Capacitor Selection The output capacitor of the SEPIC sees very large ripple currents (similar to the output capacitor of a boost converter). The rms current through the output capacitor is given by: The input capacitor should be capable of handling the rms current. Although the input capacitor is not as critical in a The ESR and ESL of the output capacitor directly control the output ripple. Use low capacitors with low ESR and ESL at the output for high efficiency and low ripple voltage. Surface mount tantalums, surface mount polymer electrolytic and polymer tantalum, Sanyo- OSCON, or multi-layer ceramic capacitors are recommended at the output for low ripple. Other Application Circuit 10135543 FIGURE 14. Typical Flyback Circuit 19 www.national.com LM3478/LM3478Q Physical Dimensions inches (millimeters) unless otherwise noted www.national.com 20 LM3478/LM3478Q Notes 21 www.national.com LM3478/LM3478Q High Efficiency Low-Side N-Channel Controller for Switching Regulator Notes For more National Semiconductor product information and proven design tools, visit the following Web sites at: Products Amplifiers Audio Clock and Timing Data Converters Interface LVDS Power Management Switching Regulators LDOs LED Lighting Voltage Reference PowerWise® Solutions Serial Digital Interface (SDI) Temperature Sensors Wireless (PLL/VCO) www.national.com/amplifiers www.national.com/audio www.national.com/timing www.national.com/adc www.national.com/interface www.national.com/lvds www.national.com/power www.national.com/switchers www.national.com/ldo www.national.com/led www.national.com/vref www.national.com/powerwise www.national.com/sdi www.national.com/tempsensors www.national.com/wireless WEBENCH® Tools App Notes Reference Designs Samples Eval Boards Packaging Green Compliance Distributors Design Support www.national.com/webench www.national.com/appnotes www.national.com/refdesigns www.national.com/samples www.national.com/evalboards www.national.com/packaging www.national.com/quality/green www.national.com/contacts www.national.com/quality www.national.com/feedback www.national.com/easy www.national.com/solutions www.national.com/milaero www.national.com/solarmagic www.national.com/AU Quality and Reliability Feedback/Support Design Made Easy Solutions Mil/Aero SolarMagic™ Analog University® THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. 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