LM34917A

LM34917A Ultra Small 33V, 1.25A Constant On-Time Buck Switching Regulator with Intelligent Current Limit May 8, 2008 LM34917A Ultra Small 33V, 1.25A Constant On-Time Buck Switching Regulator with Intelligent Current Limit General Description The LM34917A Step-Down Switching Regulator features all the functions needed to implement a low cost, efficient, buck bias regulator capable of supplying at least 1.25A to the load. To reduce excessive switch current due to the possibility of a saturating inductor the valley current limit threshold changes with input and output voltages, and the on-time is reduced when current limit is detected. This buck regulator contains an N-Channel Buck Switch, and is available in the 12 pin micro SMD package. The constant on-time feedback regulation scheme requires no loop compensation, results in fast load transient response, and simplifies circuit implementation. The operating frequency remains constant with line and load variations due to the inverse relationship between the input voltage and the on-time. The valley current limit results in a smooth transition from constant voltage to constant current mode when current limit is detected, reducing the frequency and output voltage, without the use of foldback. Additional features include: VCC under-voltage lock-out, input over-voltage shutdown, thermal shutdown, gate drive under-voltage lock-out, and maximum duty cycle limit. ■ Transient Capability to 50V ■ Integrated N-Channel buck switch ■ Valley current limit varies with VIN and VOUT to reduce ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ excessive inductor current On-time is reduced when in current limit Integrated start-up regulator No loop compensation required Ultra-Fast transient response Maximum switching frequency: 2 MHz Operating frequency remains nearly constant with load current and input voltage variations Programmable soft-start Precision internal reference Adjustable output voltage Thermal shutdown Typical Applications ■ High Efficiency Point-Of-Load (POL) Regulator ■ Non-Isolated Buck Regulator ■ Secondary High Voltage Post Regulator Features ■ Functional Input Voltage Range: 8V to 33V ■ Micro SMD package ■ Input Over-Voltage Shutdown at ≊35V Package ■ micro SMD Package Basic Step Down Regulator 20216601 © 2008 National Semiconductor Corporation 202166 www.national.com LM34917A Connection Diagrams 20216629 20216630 Bump Side Top View Ordering Information Order Number LM34917ATL LM34917ATLX Package Type 12-Bump micro SMD NSC Package Drawing TLA12UNA Junction Temperature Range −40°C to + 125°C Supplied As 250 Units Tape and Reel 3k Units Tape and Reel Pin Descriptions Pin Number A1 A2 A3 B1 B2 B3 C1,C2 C3 Name SGND RTN FB ISEN RON/SD SS VIN VCC Description Sense Ground Circuit Ground Feedback input from the regulated output Current sense On-time control and shutdown Softstart Input supply voltage Output from the startup regulator Application Information Re-circulating current flows into this pin to the current sense resistor. Ground for all internal circuitry other than the current limit detection. Internally connected to the regulation and over-voltage comparators. The regulation level is 2.5V. The re-circulating current flows out of this pin to the freewheeling diode. An external resistor from VIN to this pin sets the buck switch on-time. Grounding this pin shuts down the regulator. An internal current source charges an external capacitor to 2.5V, providing the softstart function. Operating input range is 8.0V to 33V, with over-voltage shutdown internally set at ≊35V. Transient capability is 50V. Nominally regulated at 7.0V. Connect a 0.1 µF capacitor from this pin to RTN. An external voltage (8V to 14V) can be applied to this pin to reduce internal dissipation. An internal diode connects VCC to VIN. Internally connected to the buck switch source. Connect to the inductor, diode, and bootstrap capacitor. Connect a 0.022 µF capacitor from SW to this pin. The capacitor is charged each off-time via an internal diode. D1,D2 D3 SW BST Switching Node Boost pin for bootstrap capacitor www.national.com 2 LM34917A Absolute Maximum Ratings (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. VIN to RTN BST to RTN SW to RTN (Steady State) BST to VCC VIN to SW BST to SW VCC to RTN SGND to RTN 50V 64V -1.5V 50V 50V 14V 14V -0.3V to +0.3V Current out of ISEN SS to RTN All Other Inputs to RTN ESD Rating (Note 2) Human Body Model Storage Temperature Range Junction Temperature See text -0.3V to 4V -0.3 to 7V 2kV -65°C to +150°C 150°C (Note 1) 8.0V to 33V −40°C to + 125°C Operating Ratings VIN Voltage Junction Temperature Electrical Characteristics Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are guaranteed through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only. Unless otherwise stated the following conditions apply: VIN = 12V, RON = 200kΩ. See (Note 4) and (Note 5). Symbol Start-Up Regulator, VCC VCCReg VCC regulated output VIN-VCC dropout voltage VCC output impedance (0 mA ≤ ICC ≤ 5 mA) UVLOVCC VCC current limit (Note 3) VCC under-voltage lockout threshold UVLOVCC hysteresis UVLOVCC filter delay IIN operating current IIN shutdown current Switch Characteristics Rds(on) UVLOGD Softstart Pin VSS ISS VRES Current Limit ILIM Threshold VIN = 8V, VFB = 2.4V VIN = 30V, VFB = 2.4V VIN = 30V, VFB = 1.0V Response time 1.15 1.05 0.95 1.35 1.2 1.15 150 1.55 1.45 1.35 ns A Pull-up voltage Internal current source Restart threshold after OVP shutdown 2.5 11.6 0.18 V µA V Buck Switch Rds(on) Gate Drive UVLO UVLOGD hysteresis ITEST = 200 mA VBST - VSW Increasing 2.65 0.33 4 450 0.7 4.62 Ω V mV Vin > 9V ICC = 0 mA, VCC = UVLOVCC + 250 mV VIN = 8V VIN = 12V VCC = 0V VCC increasing VCC decreasing 100 mV overdrive Non-switching, FB = 3V RON/SD = 0V 6.6 7.0 1.3 150 0.75 11 5.45 145 3 0.68 85 0.95 160 mA V mV µs mA µA 7.4 V V Ω Parameter Conditions Min Typ Max Units 3 www.national.com LM34917A Symbol On Timer tON - 1 tON - 2 tON - 3 Parameter On-time (normal operation) On-time (normal operation) On-time (current limit) Shutdown threshold at RON/ SD Shutdown Threshold hysteresis Conditions VIN = 10V, RON = 200 kΩ VIN = 32V, RON = 200 kΩ VIN = 10V, RON = 200 kΩ Voltage at RON/SD rising Voltage at RON/SD falling Min 2.1 Typ 2.8 860 1.13 Max 3.5 Units µs ns µs 0.3 0.65 40 1.0 V mV Off Timer tOFF VREF Minimum Off-time FB regulation threshold FB over-voltage threshold FB bias current Input Over-Voltage Shutdown VIN(OV) Shutdown voltage threshold at VIN increasing VIN Thermal shutdown temperature Junction temperature rising Thermal shutdown hysteresis Thermal Resistance θJA Junction to Ambient 0 LFPM Air Flow 58 °C/W 33.0 34.8 36.9 V FB = 3V SS pin = steady state 2.445 90 2.50 2.9 10 2.550 ns V V nA Regulation and Over-Voltage Comparators (FB Pin) Thermal Shutdown TSD 175 20 °C °C Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics. Note 2: The human body model is a 100pF capacitor discharged through a 1.5kΩ resistor into each pin. Note 3: VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading Note 4: For detailed information on soldering micro SMD packages, refer to AN-1112 available from National Semiconductor Corporation. Note 5: Typical specifications represent the most likely parametric norm at 25°C operation. www.national.com 4 LM34917A Typical Performance Characteristics Unless otherwise specified the following conditions apply: TJ = 25°C Efficiency at 1.5 MHz Efficiency at 2 MHz 20216645 20216646 VCC vs. VIN ON-Time vs. VIN and RON 20216603 20216604 Valley Current Limit Threshold vs. VFB and VIN Voltage at the RON/SD Pin 20216606 20216608 5 www.national.com LM34917A VCC vs. ICC ICC vs Externally Applied VCC 20216607 20216640 Shutdown and Operating Current Into VIN 20216644 www.national.com 6 LM34917A Typical Application Circuit and Block Diagram 7 20216609 www.national.com LM34917A 20216610 FIGURE 1. Startup Sequence Functional Description The LM34917A Step Down Switching Regulator features all the functions needed to implement a low cost, efficient buck bias power converter capable of supplying at least 1.25A to the load. This high voltage regulator contains an N-Channel buck switch, is easy to implement, and is available in the micro SMD package. The regulator’s operation is based on a constant on-time control scheme where the on-time is inversely proportional to the input voltage. This feature results in the operating frequency remaining relatively constant with load and input voltage variations. The feedback control scheme requires no loop compensation resulting in very fast load transient response. The valley current limit scheme protects against excessively high currents if the output is short circuited when VIN is high. To aid in controlling excessive switch current due to a possible saturating inductor the valley current limit threshold changes with input and output voltages, and the on-time is reduced by approximately 50% when current limit is detected. An over-voltage detection at VIN stops the circuit's switching when the input voltage exceeds 34.8V. The LM34917A can be applied in numerous applications to efficiently regulate down higher voltages. Additional features include: Thermal shutdown, VCC under-voltage lock-out, gate drive under-voltage lock-out, and maximum duty cycle limit. Control Circuit Overview The LM34917A buck DC-DC regulator employs a control scheme based on a comparator and a one-shot on-timer, with the output voltage feedback (FB) compared to an internal reference (2.5V). If the FB voltage is below the reference the buck switch is switched on for a time period determined by the input voltage and a programming resistor (RON). Following the on-time the switch remains off until the FB voltage falls below the reference, but for a time not less than the minimum off-time forced by the LM34917A. The buck switch is then switched on for another on-time period. When in regulation, the LM34917A operates in continuous conduction mode at heavy load currents and discontinuous conduction mode at light load currents. In continuous conduction mode the inductor’s current is always greater than www.national.com 8 LM34917A zero, and the operating frequency remains relatively constant with load and line variations. The minimum load current for continuous conduction mode is one-half the inductor’s ripple current amplitude. The approximate operating frequency is calculated as follows: ON-Time Timer The on-time for the LM34917A is determined by the RON resistor and the input voltage (VIN), calculated from: (4) (1) The buck switch duty cycle is equal to: The inverse relationship with VIN results in a nearly constant frequency as VIN is varied. To set a specific continuous conduction mode switching frequency (fSW), the RON resistor is determined from the following: (2) In discontinuous conduction mode, where the inductor’s current reaches zero during the off-time forcing a longer-thannormal off-time, the operating frequency is lower than in continuous conduction mode, and varies with load current. Conversion efficiency is maintained at light loads since the switching losses reduce with the reduction in load and frequency. The approximate discontinuous operating frequency can be calculated as follows: (5) Equations 1, 4 and 5 are valid only during normal operation i.e., the circuit is not in current limit. When the LM34917A operates in current limit, the on-time is reduced by approximately 50%. This feature reduces the peak inductor current which may be excessively high if the load current and the input voltage are simultaneously high. This feature operates on a cycle-by-cycle basis until the load current is reduced and the output voltage resumes its normal regulated value. Equations 1, 4 and 5 have a ±25% tolerance. (3) where RL = the load resistance, and L1 is the circuit’s inductor. The output voltage is set by the two feedback resistors (R1, R2 in the Block Diagram). The regulated output voltage is calculated as follows: VOUT = 2.5 x (R1 + R2) / R2 Output voltage regulation is based on supplying ripple voltage to the feedback input (FB pin) in phase with the SW pin. The LM34917A requires a minimum of 25 mVp-p of ripple voltage at the FB pin. The ripple is generated as a triangle wavefrom at the junction of R3 and C8 as the SW pin switches high and low, and fed to the FB pin by C7. If the voltage at FB rises above 2.9V, due to a transient at VOUT or excessive inductor current which creates higher than normal ripple at VOUT, the internal over-voltage comparator immediately shuts off the internal buck switch. The next ontime starts when the voltage at FB falls below 2.5V and the inductor current falls below the current limits threshold. Remote Shutdown The LM34917A can be remotely shut down by taking the RON/SD pin below 0.65V. See Figure 2. In this mode the SS pin is internally grounded, the on-timer is disabled, and bias currents are reduced. Releasing the RON/SD pin allows the circuit to resume operation after the SS pin voltage is below 0.18V. The voltage at the RON/SD pin is normally between 1.4V and 3.5V, depending on VIN and the RON resistor. 20216616 FIGURE 2. Remote Shutdown Input Over-Voltage Shutdown If the input voltage at VIN increases above 34.8V an internal comparator disables the buck switch and the on-timer, and grounds the soft-start pin. Normal operation resumes when the VIN voltage reduces below 34.8V, and when the soft-start voltage (at the SS pin) has reduced below 0.18V. Current Limit Current limit detection occurs during the off-time by monitoring the recirculating current flowing out of the ISEN pin. Referring to the Block Diagram, during the off-time the inductor current flows through the load, into SGND, through the internal sense resistor, out of ISEN and through D1 to the inductor. If that current exceeds the current limit threshold the current limit comparator output delays the start of the next on- 9 www.national.com LM34917A time period. The next on-time starts when the current out of ISEN is below the threshold and the voltage at FB falls below 2.5V. The operating frequency is typically lower due to longerthan-normal off-times. The valley current limit threshold is a function of the input voltage (VIN) and the output voltage sensed at FB, as shown in the graph “Valley Current Limit Threshold vs. VFB and VIN”. This feature reduces the inductor current’s peak value at high line and load. To further reduce the inductor’s peak current, the next cycle’s on-time is reduced by approximately 50% if the voltage at FB is below its threshold when the inductor current reduces to the current limit threshold (VOUT is low due to current limiting). Figure 3 illustrates the inductor current waveform during normal operation and in current limit. During the first “Normal Operation” the load current is IOUT1, the average of the ripple waveform. As the load resistance is reduced, the inductor current increases until it exceeds the current limit threshold. During the “Current Limited” portion of Figure 3, the current limit threshold lowers since the high load current causes VOUT (and the voltage at FB) to reduce. The on-time is reduced by approximately 50%, resulting in lower ripple amplitude for the inductor’s current. During this time the LM34917A is in a constant current mode, with an average load current equal to the current limit threshold + ΔI/2 (IOUT2). Normal operation resumes when the load current is reduced to IOUT3, allowing VOUT, the current limit threshold, and the on-time to return to their normal values. Note that in the second period of “Normal Operation”, even though the inductor’s peak current exceeds the current limit threshold during part of each cycle, the circuit is not in current limit since the current falls below the threshold before the feedback voltage reduces to its threshold to initiate the next on-time. The peak current allowed through the buck switch, and the ISEN pin, is 2A, and the maximum allowed average current is 1.5A. 20216617 FIGURE 3. Inductor Current - Normal and Current Limit Operation N - Channel Buck Switch and Driver The LM34917A integrates an N-Channel buck switch and associated floating high voltage gate driver. The gate driver circuit works in conjunction with an external bootstrap capacitor and an internal high voltage diode. A 0.022 µF capacitor (C4) connected between BST and SW provides the voltage to the driver during the on-time. During each off-time, the SW pin is at approximately -1V, and C4 is recharged for the next on-time from VCC through the internal diode. The minimum off-time ensures a minimum time each cycle to recharge the bootstrap capacitor. Softstart The softstart feature allows the converter to gradually reach a steady state operating point, thereby reducing start-up stresses and current surges. Upon turn-on, after VCC reaches the under-voltage threshold, an internal 11.6 µA current source charges up the external capacitor at the SS pin to 2.5V (t2 in Figure 1). The ramping voltage at SS (and the non-inverting input of the regulation comparator) ramps up the output voltage in a controlled manner. An internal switch grounds the SS pin if VCC is below the under-voltage lockout threshold, if the RON/SD pin is grounded, or if VIN exceeds the overvoltage threshold. Thermal Shutdown The LM34917A should be operated so the junction temperature does not exceed 125°C. If the junction temperature increases above that, an internal Thermal Shutdown circuit activates (typically) at 175°C, taking the controller to a low power reset state by disabling the buck switch. This feature www.national.com 10 LM34917A helps prevent catastrophic failures from accidental device overheating. When the junction temperature reduces below 155°C (typical hysteresis = 20°C), normal operation resumes. Applications Information EXTERNAL COMPONENTS The procedure for calculating the external components is illustrated with the following design example. Referring to the Block Diagram, the circuit is to be configured for the following specifications: - VOUT = 5V - VIN = 8V to 33V - Minimum load current = 200 mA - Maximum load current = 1000 mA - Switching Frequency = 1.5 MHz - Soft-start time = 5 ms - Output voltage ripple level: Minimum R1 and R2: These resistors set the output voltage. The ratio of the feedback resistors is calculated from: R1/R2 = (VOUT/2.5V) - 1 For this example, R1/R2 = 1. R1 and R2 should be chosen from standard value resistors in the range of 1.0 kΩ – 10 kΩ which satisfy the above ratio. For this example, 2.49 kΩ is chosen for R1 and R2. RON: This resistor sets the on-time, and (by default) the switching frequency. Since the maximum frequency is limited by the minimum off-time forced by the LM34917A, first check that the desired frequency is less than: A standard value 15 µH inductor is selected. The maximum ripple amplitude, which occurs at maximum VIN, calculates to 351 mA p-p, and the peak current is 1175 mA at maximum load current. Ensure the selected inductor is rated for this peak current. C2: C2 should typically be no smaller than 3.3 µF, although that is dependent on the frequency and the desired output characteristics. C2 should be a low ESR good quality ceramic capacitor. Experimentation is usually necessary to determine the minimum value for C2, as the nature of the load may require a larger value. A load which creates significant transients requires a larger value for C2 than a non-varying load. C1 and C5: C1’s purpose is to supply most of the switch current during the on-time, and limit the voltage ripple at VIN, since it is assumed the voltage source feeding VIN has some amount of source impedance. At maximum load current, when the buck switch turns on, the current into VIN suddenly increases to the lower peak of the inductor’s ripple current, ramps up to the upper peak, then drops to zero at turn-off. The average current during the ontime is the load current. For a worst case calculation, C1 must supply this average load current during the maximum on-time, without letting the voltage at VIN drop below ≊7.5V. The minimum value for C1 is calculated from: The RON resistor is calculated from equation 5 using the minimum input voltage: Equation 4 is used to verify that this value resistor does not set an on-time less than 120 ns at maximum input voltage. A standard value 22.1 kΩ resistor is used, resulting in a nominal frequency of 1.49 MHz. The minimum on-time is 188 ns at Vin = 33V, and the maximum on-time is 510 ns at Vin = 8V. L1: The main parameter affected by the inductor is the inductor current ripple amplitude (IOR). The minimum load current is used to determine the maximum allowable ripple in order to maintain continuous conduction mode, where the lower peak does not reach 0 mA. This is not a requirement of the LM34917A, but serves as a guideline for selecting L1. For this example, the maximum ripple current should be less than: IOR(MAX) = 2 x IOUT(min) = 400 mAp-p (6) For other applications, if the minimum load current is zero, use 20% of IOUT(max) for IOUT(min) in equation 6. The ripple amplitude calculated in Equation 6 is then used in the following equation: where tON is the maximum on-time, and ΔV is the allowable ripple voltage at VIN (0.5V at VIN = 8V). C5’s purpose is to minimize transients and ringing due to long lead inductance leading the VIN pin. A low ESR 0.1 µF ceramic chip capacitor must be located close to the VIN and RTN pins. C3: The capacitor at the VCC pin provides noise filtering and stability for the VCC regulator. C3 should be no smaller than 0.1 µF, and should be a good quality, low ESR ceramic capacitor. C3’s value, and the VCC current limit, determine a portion of the turn-on-time (t1 in Figure 1). C4: The recommended value for C4 is 0.022 µF. A high quality ceramic capacitor with low ESR is recommended as C4 supplies a surge current to charge the buck switch gate at each turn-on. A low ESR also helps ensure a complete recharge during each off-time. C6: The capacitor at the SS pin determines the soft-start time, i.e. the time for the output voltage to reach its final value (t2 in Figure 1). The capacitor value is determined from: R3, C7, C8: The ripple amplitude at VOUT is determined by C2’s characteristics and the inductor’s ripple current amplitude, and typically ranges from 5 mV to 30 mV over the Vin range. Since the LM34917A’s regulation comparator requires a minimum of 25 mVp-p ripple at the FB pin, these three components are added to generate and provide the necessary ripple to FB in phase with the waveform at SW. R3 and C8 are chosen to generate a sawtooth waveform at their junction, and that voltage is AC coupled to the FB pin via C7. To determine the values for R3, C7 and C8, the following procedure is used: Calculate VA = VOUT – (VSW x (1 – (VOUT/VIN(min))) (7) 11 www.national.com LM34917A where VSW is the absolute value of the voltage at the SW pin during the off-time (typically 1V). VA, the DC voltage at the R3/ C8 junction, calculates to 4.63V, and is used in the next equation. the SW pin may inadvertently affect the IC’s operation through external or internal EMI. The diode must be rated for the maximum input voltage, the maximum load current, and the peak current which occurs when the current limit and maximum ripple current are reached simultaneously. The diode’s average power dissipation is calculated from: PD1 = VF x IOUT x (1-D) where VF is the diode’s forward voltage drop, and D is the ontime duty cycle. FINAL CIRCUIT The final circuit is shown in Figure 4, and its performance is shown in Figure 5 and Figure 6. Current limit measured approximately 1.34A at Vin = 8V, and 1.27A at Vin = 33V. The output ripple amplitude measured 4 mVp-p at Vin = 8V, and 14 mVp-p at Vin = 33V. where tON is the maximum on-time (at minimum input voltage), and ΔV is the desired ripple amplitude at the R3/C8 junction, typically 100 mV. R3 and C8 are chosen from standard value components to satisfy the above product. For this example, 3300 pF is chosen for C8, and 5.23 kΩ is chosen for R3. C7 is chosen large compared to C8, typically 0.1 µF. D1: A Schottky diode is recommended. Ultra-fast recovery diodes are not recommended as the high speed transitions at 20216643 FIGURE 4. Example Circuit 20216633 FIGURE 5. Efficiency vs. Load Current and VIN (Circuit of Figure 4) www.national.com 12 LM34917A 20216634 FIGURE 6. Frequency vs. VIN (Circuit of Figure 4) ALTERNATE OUTPUT RIPPLE CONFIGURATIONS For applications which can accept higher levels of ripple at VOUT, the following configurations are simpler and a bit more economical. a) Alternate #1: In Figure 7 R3, C7 and C8 are removed, and Cff and R4 are installed, resulting in a higher ripple level than the circuit of Figure 4. Ripple is created at VOUT by the inductor’s ripple current passing through R4. That ripple voltage is AC coupled to the FB pin through Cff, allowing the minimum ripple at VOUT to be set at 25 mVp-p. The minimum ripple current amplitude (IOR(min)) is calculated by re-arranging Equation 7 using tON(max) and VIN(min). The minimum value for R4 is calculated from: circuits of Figure 4 and Figure 7. Ripple is created at VOUT by the inductor’s ripple current passing through R4. That ripple voltage is coupled to the FB pin through the feedback resistors (R1, R2). Since the LM34917A requires a minimum of 25 mVp-p ripple at the FB pin, the ripple required at VOUT is higher than 25 mVp-p by the gain of the feedback resistors. The minimum ripple current (IOR(min)) is calculated by re-arranging Equation 7 using tON(max) and VIN(min). The minimum value for R4 is calculated from: The next larger standard value resistor should be used for R4. The next larger standard value resistor should be selected for R4 to allow for tolerances. The minimum value for Cff is determined from: 20216627 The next larger standard value capacitor should be used for Cff. FIGURE 8. Maximum Ripple Configuration c) Alternate minimum ripple configuration: The circuit in Figure 9 is the same as that in Figure 8, except the output voltage is taken from the junction of R4 and C2. The ripple at VOUT is determined by the inductor’s ripple current and C2’s characteristics. However, R4 slightly degrades the load regulation. This circuit may be suitable if the load current is fairly constant. R4 is calculated as described in Alternate #2 above. 20216626 FIGURE 7. Reduced Ripple Configuration b) Alternate #2: In Figure 8, R3, C7 and C8 are removed, and R4 is installed, resulting in a higher ripple level than the 13 www.national.com LM34917A 20216628 FIGURE 9. Alternate Minimum Output Ripple Configuration Minimum Load Current The LM34917A requires a minimum load current of 1 mA. If the load current falls below that level, the bootstrap capacitor (C4) may discharge during the long off-time, and the circuit will either shutdown, or cycle on and off at a low frequency. If the load current is expected to drop below 1 mA in the application, R1 and R2 should be chosen low enough in value so they provide the minimum required current at nominal VOUT. PC BOARD LAYOUT Refer to application note AN-1112 for PC board guidelines for the Micro SMD package. The LM34917A regulation, over-voltage, and current limit comparators are very fast, and respond to short duration noise pulses. Layout considerations are therefore critical for optimum performance. The layout must be as neat and com- pact as possible, and all of the components must be as close as possible to their associated pins. The two major current loops have currents which switch very fast, and so the loops should be as small as possible to minimize conducted and radiated EMI. The first loop is that formed by C1, through the VIN to SW pins, L1, C2, and back to C1.The second current loop is formed by D1, L1, C2 and the SGND and ISEN pins. The power dissipation within the LM34917A can be approximated by determining the total conversion loss (PIN - POUT), and then subtracting the power losses in the free-wheeling diode and the inductor. The power loss in the diode is approximately: PD1 = Iout x VF x (1-D) where Iout is the load current, VF is the diode’s forward voltage drop, and D is the on-time duty cycle. The power loss in the inductor is approximately: PL1 = Iout2 x RL x 1.1 where RL is the inductor’s DC resistance, and the 1.1 factor is an approximation for the AC losses. If it is expected that the internal dissipation of the LM34917A will produce excessive junction temperatures during normal operation, good use of the PC board’s ground plane can help to dissipate heat. Additionally the use of wide PC board traces, where possible, can help conduct heat away from the IC. Judicious positioning of the PC board within the end product, along with the use of any available air flow (forced or natural convection) can help reduce the junction temperatures. www.national.com 14 LM34917A Physical Dimensions inches (millimeters) unless otherwise noted Note: X1 = 1.971 mm, ±0.030 mm X2 = 2.301 mm, ±0.030 mm X3 = 0.60 mm, ±0.075 mm 12-Bump micro SMD Package NS Package Number TLA12UNA 15 www.national.com LM34917A Ultra Small 33V, 1.25A Constant On-Time Buck Switching Regulator with Intelligent Current Limit Notes For more National Semiconductor product information and proven design tools, visit the following Web sites at: Products Amplifiers Audio Clock Conditioners Data Converters Displays Ethernet Interface LVDS Power Management Switching Regulators LDOs LED Lighting PowerWise Serial Digital Interface (SDI) Temperature Sensors Wireless (PLL/VCO) www.national.com/amplifiers www.national.com/audio www.national.com/timing www.national.com/adc www.national.com/displays www.national.com/ethernet www.national.com/interface www.national.com/lvds www.national.com/power www.national.com/switchers www.national.com/ldo www.national.com/led www.national.com/powerwise www.national.com/sdi www.national.com/tempsensors www.national.com/wireless WEBENCH Analog University App Notes Distributors Green Compliance Packaging Design Support www.national.com/webench www.national.com/AU www.national.com/appnotes www.national.com/contacts www.national.com/quality/green www.national.com/packaging www.national.com/quality www.national.com/refdesigns www.national.com/feedback Quality and Reliability Reference Designs Feedback THE CONTENTS OF THIS DOCUMENT ARE PROVIDED IN CONNECTION WITH NATIONAL SEMICONDUCTOR CORPORATION (“NATIONAL”) PRODUCTS. NATIONAL MAKES NO REPRESENTATIONS OR WARRANTIES WITH RESPECT TO THE ACCURACY OR COMPLETENESS OF THE CONTENTS OF THIS PUBLICATION AND RESERVES THE RIGHT TO MAKE CHANGES TO SPECIFICATIONS AND PRODUCT DESCRIPTIONS AT ANY TIME WITHOUT NOTICE. NO LICENSE, WHETHER EXPRESS, IMPLIED, ARISING BY ESTOPPEL OR OTHERWISE, TO ANY INTELLECTUAL PROPERTY RIGHTS IS GRANTED BY THIS DOCUMENT. TESTING AND OTHER QUALITY CONTROLS ARE USED TO THE EXTENT NATIONAL DEEMS NECESSARY TO SUPPORT NATIONAL’S PRODUCT WARRANTY. EXCEPT WHERE MANDATED BY GOVERNMENT REQUIREMENTS, TESTING OF ALL PARAMETERS OF EACH PRODUCT IS NOT NECESSARILY PERFORMED. NATIONAL ASSUMES NO LIABILITY FOR APPLICATIONS ASSISTANCE OR BUYER PRODUCT DESIGN. BUYERS ARE RESPONSIBLE FOR THEIR PRODUCTS AND APPLICATIONS USING NATIONAL COMPONENTS. PRIOR TO USING OR DISTRIBUTING ANY PRODUCTS THAT INCLUDE NATIONAL COMPONENTS, BUYERS SHOULD PROVIDE ADEQUATE DESIGN, TESTING AND OPERATING SAFEGUARDS. EXCEPT AS PROVIDED IN NATIONAL’S TERMS AND CONDITIONS OF SALE FOR SUCH PRODUCTS, NATIONAL ASSUMES NO LIABILITY WHATSOEVER, AND NATIONAL DISCLAIMS ANY EXPRESS OR IMPLIED WARRANTY RELATING TO THE SALE AND/OR USE OF NATIONAL PRODUCTS INCLUDING LIABILITY OR WARRANTIES RELATING TO FITNESS FOR A PARTICULAR PURPOSE, MERCHANTABILITY, OR INFRINGEMENT OF ANY PATENT, COPYRIGHT OR OTHER INTELLECTUAL PROPERTY RIGHT. LIFE SUPPORT POLICY NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS PRIOR WRITTEN APPROVAL OF THE CHIEF EXECUTIVE OFFICER AND GENERAL COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein: Life support devices or systems are devices which (a) are intended for surgical implant into the body, or (b) support or sustain life and whose failure to perform when properly used in accordance with instructions for use provided in the labeling can be reasonably expected to result in a significant injury to the user. A critical component is any component in a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system or to affect its safety or effectiveness. National Semiconductor and the National Semiconductor logo are registered trademarks of National Semiconductor Corporation. All other brand or product names may be trademarks or registered trademarks of their respective holders. Copyright© 2008 National Semiconductor Corporation For the most current product information visit us at www.national.com National Semiconductor Americas Technical Support Center Email: support@nsc.com Tel: 1-800-272-9959 www.national.com National Semiconductor Europe Technical Support Center Email: europe.support@nsc.com German Tel: +49 (0) 180 5010 771 English Tel: +44 (0) 870 850 4288 National Semiconductor Asia Pacific Technical Support Center Email: ap.support@nsc.com National Semiconductor Japan Technical Support Center Email: jpn.feedback@nsc.com