LM4805_0508

LM4805 Low Voltage High Power Audio Power Amplifier August 2005 LM4805 Low Voltage High Power Audio Power Amplifier General Description The LM4805 is a boosted audio power amplifier designed for driving 8ohm speakers in portable applications. It delivers at least 1W continuous power to an 8Ω load from any input voltage between 3V and 4.6V with less than 2% THD+N. Boomer audio power amplifiers were designed specifically to provide high quality output power with a minimal amount of external components. The LM4805 does not require bootstrap capacitors, or snubber circuits. Therefore it is ideally suited for portable applications requiring high power and minimal size. The LM4805 features a low-power consumption shutdown mode along with an internal thermal shutdown protection mechanism and short circuit protection. The LM4805 contains advanced pop & click circuitry that eliminates noises which would otherwise occur during turn-on and turn-off transitions. The LM4805 is unity-gain stable and can be configured by external gain-setting resistors. Key Specifications j Quiescent Power Supply Current (VDD = 3V) j Output Power 14mA (typ) 1W (typ) 2µA (max) (VDD = 3.0V, RL = 8Ω, THD+N = 2%) j Shutdown Current Features n Pop & click circuitry eliminates noise during turn-on and turn-off transitions n Low, 2µA (max) shutdown current n Low, 14mA (typ) quiescent current n Unity-gain stable n External gain configuration capability Applications n Cellphone n PTT (Push To Talk) mobile phones Connection Diagrams LM4805LQ (5x5) LQ Marking 20126256 Top View U = Wafer Fab Z = Assembly Plant Code XY = Date Code TT = Die Run Code 20126284 Top View Order Number LM4805LQ See NS Package Number LQA28A Boomer ® is a registered trademark of National Semiconductor Corporation. © 2005 National Semiconductor Corporation DS201262 www.national.com LM4805 Typical Application 20126210 * Cf2 is optional. FIGURE 1. Typical Audio Amplifier Application Circuit www.national.com 2 LM4805 Absolute Maximum Ratings (Notes 1, 2) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. Supply Voltage (VDD) Supply Voltage (V1) Storage Temperature Input Voltage Power Dissipation (Note 3) ESD Susceptibility (Note 4) ESD Susceptibility (Note 5) 6.5V 6.5V −65˚C to +150˚C −0.3V to VDD + 0.3V Internally limited 2000V 200V Junction Temperature Thermal Resistance θJA (LLP) 125˚C 59˚C/W See AN-1187 ’Leadless Leadframe Packaging (LLP).’ Operating Ratings Temperature Range TMIN ≤ TA ≤ TMAX Supply Voltage (VDD) Supply Voltage (V1) −40˚C ≤ TA ≤ +85˚C 2.7V ≤ VDD ≤ 4.6V 2.7V ≤ V1 ≤ 6.1V The following specifications apply for VDD = 4.2V, AV-BTL = 26dB, RL = 8Ω, CB = 1.0µF, R1 = 51.1kΩ, R2 = 15kΩ unless otherwise specified. Limits apply for TA = 25˚C. See Figure 1. Symbol Parameter Conditions LM4805 Typical (Note 6) IDD ISD VSDIH VSDIL TWU VOS TSD POUT THD+N eOS PSRR VFB Quiescent Power Supply Current Shutdown Current Shutdown Voltage Input High Shutdown Voltage Input Low Wake-up Time Output Offset Voltage Thermal Shutdown Temperature Output Power Total Harmomic Distortion + Noise Output Noise Power Supply Rejection Ratio Feedback Pin Reference Voltage THD = 1% (max), f = 1kHz, Mono BTL PO = 500mW, f = 1kHz A-Weighted Filter, VIN = 0V VRIPPLE = 200mVp-p, f = 100Hz, inputs terminated Note 11 1.2 0.2 105 66 1.23 VIN = 0, RLOAD = ∞ VSHUTDOWN = GND (Notes 9, 10) S/D1 and S/D2 S/D1 and S/D2 CB = 1.0µF 80 5 10 0.1 Limit (Notes 7, 8) 23 2 1.5 0.4 110 40 125 0.9 0.5 Units (Limits) mA (max) µA (max) V (min) V (max) msec (max) mV (max) ˚C (min) W (min) % (max) µV dB V Electrical Characteristics VDD = 4.2V (Notes 1, 2) The following specifications apply for VDD = 3.0V, AV-BTL = 26dB, RL = 8Ω, CB = 1.0µF, R1 = 51.1kΩ, R2 = 15kΩ unless otherwise specified. Limits apply for TA = 25˚C. Symbol Parameter Conditions LM4805 Typical (Note 6) IDD ISD VSDIH VSDIL TWU VOS TSD POUT THD+N eOS PSRR Quiescent Power Supply Current Shutdown Current Shutdown Voltage Input High Shutdown Voltage Input Low Wake-up Time Output Offset Voltage Thermal Shutdown Temperature Output Power Total Harmomic Distortion + Noise Output Noise Power Supply Rejection Ratio THD = 2% (max), f = 1kHz, Mono BTL PO = 500mW, fIN = 1kHz A-Weighted Filter, VIN = 0V VRIPPLE = 200mVp-p, f = 100Hz 3 Electrical Characteristics VDD = 3.0V (Notes 1, 2) Limit (Notes 7, 8) 27 2 1.5 0.4 Units (Limits) mA (max) µA (max) V (min) V (max) msec (max) mV (max) ˚C (min)) W (min) % (max) µV dB (min) VDD = 3.2V, VIN = 0, RLOAD = ∞ VSHUTDOWN = GND (Notes 9, 10) S/D1 and S/D2 S/D1 and S/D2 CB = 1.0µF 14 0.1 80 5 110 40 125 1 0.25 105 66 0.85 0.55 www.national.com LM4805 Electrical Characteristics VDD = 3.0V (Notes 1, 2) Symbol Parameter (Continued) The following specifications apply for VDD = 3.0V, AV-BTL = 26dB, RL = 8Ω, CB = 1.0µF, R1 = 51.1kΩ, R2 = 15kΩ unless otherwise specified. Limits apply for TA = 25˚C. Conditions LM4805 Typical (Note 6) Limit (Notes 7, 8) 1.205 1.255 Units (Limits) V (max) V (min) VFB Feedback Pin Reference Voltage (Note 11) 1.23 Note 1: All voltages are measured with respect to the GND pin, unless otherwise specified. Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit is given, however, the typical value is a good indication of device performance. Note 3: The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX, θJA, and the ambient temperature, TA. The maximum allowable power dissipation is PDMAX = (TJMAX − TA) / θJA or the given in Absolute Maximum Ratings, whichever is lower. Note 4: Human body model, 100pF discharged through a 1.5kΩ resistor. Note 5: Machine Model, 220pF–240pF discharged through all pins. Note 6: Typicals are measured at 25˚C and represent the parametric norm. Note 7: Limits are guaranteed to National’s AOQL (Average Outgoing Quality Level). Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis. Note 9: Shutdown current is measured at an ambient temperature of 25˚C. The Shutdown pin should be driven as close as possible to GND for minimum shutdown current. Note 10: Shutdown current is measured with components R1 and R2 removed. Note 11: Feedback pin reference voltage is measured with the Audio Amplifier’s V1 (pin 26) floating and no addition load connected to the cathode of D1 (see Figure 1). www.national.com 4 LM4805 Typical Performance Characteristics THD+N vs Frequency VDD = 3V, AV = 6dB, RL = 8Ω THD+N vs Frequency VDD = 3V, AV = 6dB, RL = 16Ω 20126211 20126212 THD+N vs Frequency VDD = 3V, AV = 26dB, RL = 8Ω THD+N vs Frequency VDD = 3V, AV = 26dB, RL = 16Ω 20126213 20126214 THD+N vs Frequency VDD = 4.2V, AV = 6dB, RL = 8Ω THD+N vs Frequency VDD = 4.2V, AV = 6dB, RL = 16Ω 20126215 20126216 5 www.national.com LM4805 Typical Performance Characteristics THD+N vs Frequency VDD = 4.2V, AV = 26dB, RL = 8Ω (Continued) THD+N vs Frequency VDD = 4.2V, AV = 26dB, RL = 16Ω 20126217 20126218 THD+N vs Output Power VDD = 3V, AV = 6dB, RL = 8Ω THD+N vs Output Power VDD = 3V, AV = 6dB, RL = 16Ω 20126219 20126220 THD+N vs Output Power VDD = 3V, AV = 26dB, RL = 8Ω THD+N vs Output Power VDD = 3V, AV = 26dB, RL = 16Ω 20126221 20126222 www.national.com 6 LM4805 Typical Performance Characteristics THD+N vs Output Power VDD = 4.2V, AV = 6dB, RL = 8Ω (Continued) THD+N vs Output Power VDD = 4.2V, AV = 6dB, RL = 16Ω 20126223 20126224 THD+N vs Output Power VDD = 4.2V, AV = 26dB, RL = 8Ω THD+N vs Output Power VDD = 4.2V, AV = 26dB, RL = 16Ω 20126225 20126226 Supply Current vs Supply Voltage Power Dissipation vs Output Power VDD = 3V, RL = 8Ω, f = 1kHz 20126227 20126228 7 www.national.com LM4805 Typical Performance Characteristics Power Dissipation vs Output Power VDD = 3V, RL = 16Ω, f = 1kHz (Continued) Power Dissipation vs Output Power VDD = 4.2V, RL = 8Ω, f = 1kHz 20126229 20126230 Power Dissipation vs Output Power VDD = 4.2V, RL = 16Ω, f = 1kHz Output Power vs Load Resistance VDD = 3V 20126231 20126232 Output Power vs Load Resistance VDD = 4.2V Output Power vs Supply Voltage RL = 8Ω, f = 1kHz 20126233 20126234 www.national.com 8 LM4805 Typical Performance Characteristics Output Power vs Supply Voltage RL = 16Ω, f = 1kHz (Continued) PSRR vs FREQUENCY VDD = 3V, AV = 6dB Vripple = 200mVP-P 20126235 20126236 PSRR vs FREQUENCY VDD = 3V, AV = 26dB Vripple = 200mVP-P PSRR vs FREQUENCY VDD = 4.2V, AV = 6dB Vripple = 200mVP-P 20126237 20126238 PSRR vs FREQUENCY VDD = 4.2V, AV = 26dB Vripple = 200mVP-P Load Current vs VDD 20126239 20126240 9 www.national.com LM4805 Typical Performance Characteristics Amplifier Frequency Response vs Input Capacitor Size (Continued) Amplifier Open Loop vs Frequency Respose 20126242 20126241 Switch Current Limit vs Duty Cycle - ”X” Oscillator Frequency vs Temperature - ”X” 20126244 20126243 Feedback Voltage vs Temperature Feedback Bias Current vs Temperature 20126246 20126249 www.national.com 10 LM4805 Typical Performance Characteristics Maximum Duty Cycle vs Temperature - ”X” (Continued) RDS (ON) vs Temperature 20126245 20126247 RDS (ON) vs VDD 20126248 Application Information BRIDGE CONFIGURATION EXPLANATION The Audio Amplifier portion of the LM4805 has two internal amplifiers allowing different amplifier configurations. The first amplifier’s gain is externally configurable, whereas the second amplifier is internally fixed in a unity-gain, inverting configuration. The closed-loop gain of the first amplifier is set by selecting the ratio of Rf to Ri while the second amplifier’s gain is fixed by the two internal 20kΩ resistors. Figure 1 shows that the output of amplifier one serves as the input to amplifier two. This results in both amplifiers producing signals identical in magnitude, but out of phase by 180˚. Consequently, the differential gain for the Audio Amplifier is AVD = 2 *(Rf/Ri) By driving the load differentially through outputs VO1 and VO2, an amplifier configuration commonly referred to as “bridged mode” is established. Bridged mode operation is different from the classic single-ended amplifier configuration where one side of the load is connected to ground. A bridge amplifier design has a few distinct advantages over the single-ended configuration. It provides differential drive 11 to the load, thus doubling the output swing for a specified supply voltage. Four times the output power is possible as compared to a single-ended amplifier under the same conditions. This increase in attainable output power assumes that the amplifier is not current limited or clipped. In order to choose an amplifier’s closed-loop gain without causing excessive clipping, please refer to the Audio Power Amplifier Design section. The bridge configuration also creates a second advantage over single-ended amplifiers. Since the differential outputs, VO1 and VO2, are biased at half-supply, no net DC voltage exists across the load. This eliminates the need for an output coupling capacitor which is required in a single supply, single-ended amplifier configuration. Without an output coupling capacitor, the half-supply bias across the load would result in both increased internal IC power dissipation and also possible loudspeaker damage. AMPLIFIER POWER DISSIPATION Power dissipation is a major concern when designing a successful amplifier, whether the amplifier is bridged or single-ended. A direct consequence of the increased power delivered to the load by a bridge amplifier is an increase in internal power dissipation. Since the amplifier portion of the LM4805 has two operational amplifiers, the maximum inter- www.national.com LM4805 Application Information (Continued) nal power dissipation is 4 times that of a single-ended amplifier. The maximum power dissipation for a given BTL application can be derived from Equation 1. (1) PDMAX(AMP) = 4(VDD)2 / (2π2RL) may be connected to a large plane of continuous unbroken copper. This plane forms a thermal mass, heat sink, and radiation area. Further detailed and specific information concerning PCB layout, fabrication, and mounting an LD (LLP) package is found in National Semiconductor’s Package Engineering Group under application note AN1187. SHUTDOWN FUNCTION In many applications, a microcontroller or microprocessor output is used to control the shutdown circuitry to provide a quick, smooth transition into shutdown. Another solution is to use a single-pole, single-throw switch, and a pull-up resistor. One terminal of the switch is connected to GND. The other side is connected to the two shutdown pins and the terminal of the pull-up resistor. The remaining resistance terminal is connected to VDD. If the switch is open, then the external pull-up resistor connected to VDD will enable the LM4805. This scheme guarantees that the shutdown pins will not float thus preventing unwanted state changes. PROPER SELECTION OF EXTERNAL COMPONENTS Proper selection of external components in applications using integrated power amplifiers, and switching boost converters, is critical for optimizing device and system performance. Consideration to component values must be used to maximize overall system quality. The best capacitors for use with the switching converter portion of the LM4805 are multi-layer ceramic capacitors. They have the lowest ESR (equivalent series resistance) and highest resonance frequency, which makes them optimum for high frequency switching converters. When selecting a ceramic capacitor, only X5R and X7R dielectric types should be used. Other types such as Z5U and Y5F have such severe loss of capacitance due to effects of temperature variation and applied voltage, they may provide as little as 20% of rated capacitance in many typical applications. Always consult capacitor manufacturer’s data curves before selecting a capacitor. High-quality ceramic capacitors can be obtained from Taiyo-Yuden, AVX, and Murata. POWER SUPPLY BYPASSING As with any amplifier, proper supply bypassing is critical for low noise performance and high power supply rejection. The capacitor location on both V1 and VDD (Cs2 and Cs1) pins should be as close to the device as possible. SELECTING INPUT CAPACITOR FOR AUDIO AMPLIFIER One of the major considerations is the closedloop bandwidth of the amplifier. To a large extent, the bandwidth is dictated by the choice of external components shown in Figure 1. The input coupling capacitor, Ci, forms a first order high pass filter which limits low frequency response. This value should be chosen based on needed frequency response for a few distinct reasons. High value input capacitors are both expensive and space hungry in portable designs. Clearly, a certain value capacitor is needed to couple in low frequencies without severe attenuation. However, speakers used in portable systems, whether internal or external, have little ability to reproduce signals below 100Hz to 150Hz. Thus, using a high value input capacitor may not increase actual system performance. In addition to system cost and size, click and pop performance is affected by the value of the input coupling capaci12 BOOST CONVERTER POWER DISSIPATION At higher duty cycles, the increased ON-time of the switch FET means the maximum output current will be determined by power dissipation within the LM2731 FET switch. The switch power dissipation from ON-time conduction is calculated by Equation 2. (2) PDMAX(SWITCH) = DC x IIND(AVE)2 x RDS(ON) where DC is the duty cycle. There will be some switching losses as well, so some derating needs to be applied when calculating IC power dissipation. TOTAL POWER DISSIPATION The total power dissipation for the LM4805 can be calculated by adding Equation 1 and Equation 2 together to establish Equation 3: PDMAX(TOTAL) = [4*(VDD)2/2π2RL] + [DC x IIND(AVE)2 x RDS(ON)] (3) The result from Equation 3 must not be greater than the power dissipation that results from Equation 4: PDMAX = (TJMAX - TA) / θJA (4) For the LQA28A, θJA = 59˚C/W. TJMAX = 125˚C for the LM4805. Depending on the ambient temperature, TA, of the system surroundings, Equation 4 can be used to find the maximum internal power dissipation supported by the IC packaging. If the result of Equation 3 is greater than that of Equation 4, then either the supply voltage must be increased, the load impedance increased or TA reduced. For the typical application of a 3V power supply, with V1 set to 5.5V and an 8Ω load, the maximum ambient temperature possible without violating the maximum junction temperature is approximately 111˚C provided that device operation is around the maximum power dissipation point. Thus, for typical applications, power dissipation is not an issue. Power dissipation is a function of output power and thus, if typical operation is not around the maximum power dissipation point, the ambient temperature may be increased accordingly. Refer to the Typical Performance Characteristics curves for power dissipation information for lower output levels. EXPOSED-DAP PACKAGE PCB MOUNTING CONSIDERATIONS The LM4805’s exposed-DAP (die attach paddle) package (LD) provides a low thermal resistance between the die and the PCB to which the part is mounted and soldered. The low thermal resistance allows rapid heat transfer from the die to the surrounding PCB copper traces, ground plane, and surrounding air. The LD package should have its DAP soldered to a copper pad on the PCB. The DAP’s PCB copper pad www.national.com LM4805 Application Information (Continued) SETTING THE OUTPUT VOLTAGE (V1) OF BOOST CONVERTER The output voltage is set using the external resistors R1 and R2 (see Figure 1). A value of approximately 15k is recommended for R2 to establish a divider current of approximately 92µA. R1 is calculated using the formula: R1 = R2 X (V1/1.23 − 1) (5) tor, Ci. A high value input coupling capacitor requires more charge to reach its quiescent DC voltage (nominally 1/2 VDD). This charge comes from the output via the feedback and is apt to create pops upon device enable. Thus, by minimizing the capacitor value based on desired low frequency response, turn-on pops can be minimized. SELECTING BYPASS CAPACITOR FOR AUDIO AMPLIFIER Besides minimizing the input capacitor value, careful consideration should be paid to the bypass capacitor value. Bypass capacitor, CB, is the most critical component to minimize turn-on pops since it determines how fast the amplifier turns on. The slower the amplifier’s outputs ramp to their quiescent DC voltage (nominally 1/2 VDD), the smaller the turn-on pop. Choosing CB equal to 1.0µF along with a small value of Ci (in the range of 0.039µF to 0.39µF), should produce a virtually clickless and popless shutdown function. Although the device will function properly, (no oscillations or motorboating), with CB equal to 0.1µF, the device will be much more susceptible to turn-on clicks and pops. Thus, a value of CB equal to 1.0µF is recommended in all but the most cost sensitive designs. SELECTING FEEDBACK CAPACITOR FOR AUDIO AMPLIFIER The LM4805 is unity-gain stable which gives the designer maximum system flexability. However, a typical application requires a closed-loop differential gain of 10. In this case a feedback capacitor (Cf2) can be used as shown in Figure 2 to bandwidth limit the amplifier. This feedback capacitor creates a low pass filter that eliminates possible high frequency oscillations. Care should be taken when calculating the -3dB frequency because an incorrect combination of Rf and Cf2 will cause rolloff before the desired frequency SELECTING OUTPUT CAPACITOR (CO) FOR BOOST CONVERTER A single 4.7µF to 10µF ceramic capacitor will provide sufficient output capacitance for most applications. If larger amounts of capacitance are desired for improved line support and transient response, tantalum capacitors can be used. Aluminum electrolytics with ultra low ESR such as Sanyo Oscon can be used, but are usually prohibitively expensive. Typical AI electrolytic capacitors are not suitable for switching frequencies above 500 kHz because of significant ringing and temperature rise due to self-heating from ripple current. An output capacitor with excessive ESR can also reduce phase margin and cause instability. In general, if electrolytics are used, we recommended that they be paralleled with ceramic capacitors to reduce ringing, switching losses, and output voltage ripple. SELECTING INPUT CAPACITOR (Cs1) FOR BOOST CONVERTER An input capacitor is required to serve as an energy reservoir for the current which must flow into the coil each time the switch turns ON. This capacitor must have extremely low ESR, so ceramic is the best choice. We recommend a nominal value of 4.7µF, but larger values can be used. Since this capacitor reduces the amount of voltage ripple seen at the input pin, it also reduces the amount of EMI passed back along that line to other circuitry. 13 FEED-FORWARD COMPENSATION FOR BOOST CONVERTER Although the LM4805’s internal Boost converter is internally compensated, the external feed-forward capacitor Cf1 is required for stability (see Figure 1). Adding this capacitor puts a zero in the loop response of the converter. The recommended frequency for the zero fz should be approximately 6kHz. Cf1 can be calculated using the formula: Cf1 = 1 / (2 X R1 X fz) (6) SELECTING DIODES The external diode used in Figure 1 should be a Schottky diode. A 20V diode such as the MBR0520 is recommended. The MBR05XX series of diodes are designed to handle a maximum average current of 0.5A. For applications exceeding 0.5A average but less than 1A, a Microsemi UPS5817 can be used. DUTY CYCLE The maximum duty cycle of the boost converter determines the maximum boost ratio of output-to-input voltage that the converter can attain in continuous mode of operation. The duty cycle for a given boost application is defined as: Duty Cycle = VOUT + VDIODE - VIN / VOUT + VDIODE - VSW This applies for continuous mode operation. INDUCTANCE VALUE The first question we are usually asked is: “How small can I make the inductor.” (because they are the largest sized component and usually the most costly). The answer is not simple and involves trade-offs in performance. Larger inductors mean less inductor ripple current, which typically means less output voltage ripple (for a given size of output capacitor). Larger inductors also mean more load power can be delivered because the energy stored during each switching cycle is: E = L/2 X (lp)2 Where “lp” is the peak inductor current. An important point to observe is that the LM4805 will limit its switch current based on peak current. This means that since lp(max) is fixed, increasing L will increase the maximum amount of power available to the load. Conversely, using too little inductance may limit the amount of load current which can be drawn from the output. Best performance is usually obtained when the converter is operated in “continuous” mode at the load current range of interest, typically giving better load regulation and less outwww.national.com LM4805 Application Information (Continued) ILOAD = IIND(AVG) x (1 - DC) (7) put ripple. Continuous operation is defined as not allowing the inductor current to drop to zero during the cycle. It should be noted that all boost converters shift over to discontinuous operation as the output load is reduced far enough, but a larger inductor stays “continuous” over a wider load current range. To better understand these trade-offs, a typical application circuit (5V to 12V boost with a 10µH inductor) will be analyzed. We will assume: VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V Since the frequency is 1.6MHz (nominal), the period is approximately 0.625µs. The duty cycle will be 62.5%, which means the ON-time of the switch is 0.390µs. It should be noted that when the switch is ON, the voltage across the inductor is approximately 4.5V. Using the equation: V = L (di/dt) We can then calculate the di/dt rate of the inductor which is found to be 0.45 A/µs during the ON-time. Using these facts, we can then show what the inductor current will look like during operation: Where "DC" is the duty cycle of the application. The switch current can be found by: ISW = IIND(AVG) + 1/2 (IRIPPLE) (8) Inductor ripple current is dependent on inductance, duty cycle, input voltage and frequency: IRIPPLE = DC x (VIN-VSW) / (f x L) (9) combining all terms, we can develop an expression which allows the maximum available load current to be calculated: ILOAD(max) = (1–DC)x(ISW(max)–DC(VIN-VSW))/fL (10) The equation shown to calculate maximum load current takes into account the losses in the inductor or turn-OFF switching losses of the FET and diode. DESIGN PARAMETERS VSW AND ISW The value of the FET "ON" voltage (referred to as VSW in equations 7 thru 10) is dependent on load current. A good approximation can be obtained by multiplying the "ON Resistance" of the FET times the average inductor current. FET on resistance increases at VIN values below 5V, since the internal N-FET has less gate voltage in this input voltage range (see Typical Performance Characteristics curves). Above VIN = 5V, the FET gate voltage is internally clamped to 5V. The maximum peak switch current the device can deliver is dependent on duty cycle. For higher duty cycles, see Typical Performance Characteristics curves. 20126255 FIGURE 2. 10µH Inductor Current 5V - 12V Boost (LM4805) During the 0.390µs ON-time, the inductor current ramps up 0.176A and ramps down an equal amount during the OFFtime. This is defined as the inductor “ripple current”. It can also be seen that if the load current drops to about 33mA, the inductor current will begin touching the zero axis which means it will be in discontinuous mode. A similar analysis can be performed on any boost converter, to make sure the ripple current is reasonable and continuous operation will be maintained at the typical load current values. MAXIMUM SWITCH CURRENT The maximum FET switch current available before the current limiter cuts in is dependent on duty cycle of the application. This is illustrated in a graph in the typical performance characterization section which shows typical values of switch current as a function of effective (actual) duty cycle. CALCULATING OUTPUT CURRENT OF BOOST CONVERTER (IAMP) As shown in Figure 2 which depicts inductor current, the load current is related to the average inductor current by the relation: www.national.com 14 INDUCTOR SUPPLIERS Recommended suppliers of inductors for the LM4805 include, but are not limited to Taiyo-Yuden, Sumida, Coilcraft, Panasonic, TDK and Murata. When selecting an inductor, make certain that the continuous current rating is high enough to avoid saturation at peak currents. A suitable core type must be used to minimize core (switching) losses, and wire power losses must be considered when selecting the current rating. PCB LAYOUT GUIDELINES High frequency boost converters require very careful layout of components in order to get stable operation and low noise. All components must be as close as possible to the LM4805 device. It is recommended that a 4-layer PCB be used so that internal ground planes are available. Some additional guidelines to be observed: 1. Keep the path between L1, D1, and Co extremely short. Parasitic trace inductance in series with D1 and Co will increase noise and ringing. 2. The feedback components R1, R2 and Cf 1 must be kept close to the FB pin of U1 to prevent noise injection on the FB pin trace. 3. If internal ground planes are available (recommended) use vias to connect directly to ground at pin 2 of U1, as well as the negative sides of capacitors Cs1 and Co. LM4805 Application Information GENERAL MIXED-SIGNAL LAYOUT RECOMMENDATION (Continued) Single-Point Power / Ground Connection The analog power traces should be connected to the digital traces through a single point (link). A "Pi-filter" can be helpful in minimizing high frequency noise coupling between the analog and digital sections. It is further recommended to place digital and analog power traces over the corresponding digital and analog ground traces to minimize noise coupling. Placement of Digital and Analog Components This section provides practical guidelines for mixed signal PCB layout that involves various digital/analog power and ground traces. Designers should note that these are only "rule-of-thumb" recommendations and the actual results will depend heavily on the final layout. Power and Ground Circuits For 2 layer mixed signal design, it is important to isolate the digital power and ground trace paths from the analog power and ground trace paths. Star trace routing techniques (bringing individual traces back to a central point rather than daisy chaining traces together in a serial manner) can have a major impact on low level signal performance. Star trace routing refers to using individual traces to feed power and ground to each circuit or even device. This technique will take require a greater amount of design time but will not increase the final price of the board. The only extra parts required may be some jumpers. All digital components and high-speed digital signals traces should be located as far away as possible from analog components and circuit traces. Avoiding Typical Design / Layout Problems Avoid ground loops or running digital and analog traces parallel to each other (side-by-side) on the same PCB layer. When traces must cross over each other do it at 90 degrees. Running digital and analog traces at 90 degrees to each other from the top to the bottom side as much as possible will minimize capacitive noise coupling and crosstalk. 20126250 FIGURE 3. Demo Board Reference Schematic 15 www.national.com LM4805 Demonstration Board Layout Composite Layer 20126251 Top Layer 20126253 www.national.com 16 LM4805 Demonstration Board Layout (Continued) Silkscreen 20126252 Bottom Layer 20126254 17 www.national.com LM4805 Revision History Rev 1.0 Date 8/11/05 Description Changed the project title from 3V, 1W Boosted Amplifier into Low Voltage High Power Audio Power Amplifier (per Nisha P.), then re-released D/S to the WEB. www.national.com 18 LM4805 Low Voltage High Power Audio Power Amplifier Physical Dimensions unless otherwise noted inches (millimeters) LQ Package Order Number LM4805LQ NS Package Number LQA28A National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications. For the most current product information visit us at www.national.com. LIFE SUPPORT POLICY NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury to the user. BANNED SUBSTANCE COMPLIANCE National Semiconductor manufactures products and uses packing materials that meet the provisions of the Customer Products Stewardship Specification (CSP-9-111C2) and the Banned Substances and Materials of Interest Specification (CSP-9-111S2) and contain no ‘‘Banned Substances’’ as defined in CSP-9-111S2. Leadfree products are RoHS compliant. National Semiconductor Americas Customer Support Center Email: new.feedback@nsc.com Tel: 1-800-272-9959 www.national.com National Semiconductor Europe Customer Support Center Fax: +49 (0) 180-530 85 86 Email: europe.support@nsc.com Deutsch Tel: +49 (0) 69 9508 6208 English Tel: +44 (0) 870 24 0 2171 Français Tel: +33 (0) 1 41 91 8790 National Semiconductor Asia Pacific Customer Support Center Email: ap.support@nsc.com National Semiconductor Japan Customer Support Center Fax: 81-3-5639-7507 Email: jpn.feedback@nsc.com Tel: 81-3-5639-7560 2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness.