LMV796

LMV796/LMV797 17 MHz, Low Noise, CMOS Input, 1.8V Operational Amplifiers May 2006 LMV796/LMV797 17 MHz, Low Noise, CMOS Input, 1.8V Operational Amplifiers General Description The LMV796 (Single) and the LMV797 (Dual) low noise, CMOS input operational amplifiers offer a low input voltage while consuming only 1.15 mA noise density of 5.8 nV/ (LMV796) of quiescent current. The LMV796 and LMV797 are unity gain stable op amps and have gain bandwidth of 17 MHz. The LMV796/ LMV797 have a supply voltage range of 1.8V to 5.5V and can operate from a single supply. The LMV796/LMV797 each feature a rail-to-rail output stage capable of driving a 600Ω load and sourcing as much as 60 mA of current. The LMV796 family provides optimal performance in low voltage and low noise systems. A CMOS input stage, with typical input bias currents in the range of a few femtoAmperes, and an input common mode voltage range, which includes ground, make the LMV796 and the LMV797 ideal for low power sensor applications. The LMV796/LMV797 are manufactured using National’s advanced VIP50 process and are offered in a 5-pin SOT23 and an 8-pin MSOP package respectively. Features (Typical 5V supply, unless otherwise noted) n Input referred voltage noise 5.8 nV/ n Input bias current 100 fA n Unity gain bandwidth 17 MHz n Supply current per channel — LMV796 1.15 mA — LMV797 1.30 mA n Rail-to-rail output swing — @ 10 kΩ load 25 mV from rail 35 mV from rail — @ 2 kΩ load n Guaranteed 2.5V and 5.0V performance n Total harmonic distortion 0.01% @1 kHz, 600Ω n Temperature range −40˚C to 125˚C Applications n n n n n Photodiode amplifiers Active filters and buffers Low noise signal processing Medical instrumentation Sensor interface applications Typical Application 20183569 20183539 Photodiode Transimpedance Amplifier Input Referred Voltage Noise vs. Frequency © 2006 National Semiconductor Corporation DS201835 www.national.com LMV796/LMV797 Absolute Maximum Ratings (Note 1) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. ESD Tolerance (Note 2) Human Body Model Machine Model VIN Differential Supply Voltage (V+ – V−) Input/Output Pin Voltage Storage Temperature Range Junction Temperature (Note 3) 2000V 200V Soldering Information Infrared or Convection (20 sec) Wave Soldering Lead Temp (10 sec) 235˚C 260˚C Operating Ratings (Note 1) Temperature Range (Note 3) Supply Voltage (V – V ) −40˚C ≤ TA ≤ 125˚C 0˚C ≤ TA ≤ 125˚C Package Thermal Resistance (θJA (Note 3)) 5-Pin SOT23 8-Pin MSOP 180˚C/W 236˚C/W + − ± 0.3V 6.0V V+ +0.3V, V− −0.3V −65˚C to 150˚C +150˚C −40˚C to 125˚C 2.0V to 5.5V 1.8V to 5.5V 2.5V Electrical Characteristics Unless otherwise specified, all limits are guaranteed for TA = 25˚C, V+ = 2.5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at the temperature extremes. Symbol VOS TC VOS IB Parameter Input Offset Voltage Input Offset Average Drift Input Bias Current LMV796 (Note 6) LMV797 (Note 6) VCM = 1.0V (Notes 7, 8) −40˚C ≤ TA ≤ 85˚C −40˚C ≤ TA ≤ 125˚C IOS CMRR PSRR Input Offset Current Common Mode Rejection Ratio Power Supply Rejection Ratio Input Common-Mode Voltage Range Large Signal Voltage Gain (Note 8) 0V ≤ VCM ≤ 1.4V 2.0V ≤ V+ ≤ 5.5V, VCM = 0V 1.8V ≤ V+ ≤ 5.5V, VCM = 0V CMVR AVOL CMRR ≥ 60 dB CMRR ≥ 55 dB VOUT = 0.15V to 2.2V, RLOAD = 2 kΩ to V+/2 LMV796 LMV797 VOUT = 0.15V to 2.2V, RLOAD = 10 kΩ to V+/2 VOUT Output Swing High RLOAD = 2 kΩ to V+/2 RLOAD = 10 kΩ to V+/2 Output Swing Low RLOAD = 2 kΩ to V+/2 RLOAD = 10 kΩ to V+/2 80 75 80 75 80 −0.3 -0.3 85 80 82 78 88 84 98 92 110 25 20 30 15 75 82 65 71 75 78 65 67 mV from rail Conditions Min (Note 5) Typ (Note 4) 0.1 −1.0 −1.8 0.05 0.05 10 94 100 dB 98 1.5 1.5 V 1 25 1 100 Max (Note 5) Units ± 1.35 ± 1.65 mV µV/˚C pA fA dB dB www.national.com 2 LMV796/LMV797 2.5V Electrical Characteristics IOUT Output Short Circuit Current (Continued) 35 28 7 5 47 15 0.95 1.1 8.5 10.5 14 6.2 0.01 0.01 1.30 1.65 1.50 1.85 mA Sourcing to V− VIN = 200 mV (Note 9) Sinking to V+ VIN = –200 mV (Note 9) IS Supply Current per Amplifier LMV796 LMV797 per channel mA SR GBWP en in THD+N Slew Rate Gain Bandwidth Product Input-Referred Voltage Noise Input-Referred Current Noise Total Harmonic Distortion + Noise AV = +1, Rising (10% to 90%) AV = +1, Falling (90% to 10%) f = 1 kHz f = 1 kHz f = 1 kHz, AV = 1, RLOAD = 600Ω V/µs MHz nV/ pA/ % 5V Electrical Characteristics Unless otherwise specified, all limits are guaranteed for TA = 25˚C, V+ = 5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at the temperature extremes. Symbol VOS TC VOS IB Parameter Input Offset Voltage Input Offset Average Drift Input Bias Current LMV796 (Note 6) LMV797 (Note 6) VCM = 2.0V (Notes 7, 8) −40˚C ≤ TA ≤ 85˚C −40˚C ≤ TA ≤ 125˚C IOS CMRR PSRR Input Offset Current Common Mode Rejection Ratio Power Supply Rejection Ratio Input Common-Mode Voltage Range Large Signal Voltage Gain (Note 8) 0V ≤ VCM ≤ 3.7V 2.0V ≤ V+ ≤ 5.5V, VCM = 0V 1.8V ≤ V+ ≤ 5.5V, VCM = 0V CMVR AVOL CMRR ≥ 60 dB CMRR ≥ 55 dB VOUT = 0.3V to 4.7V, RLOAD = 2 kΩ to V+/2 LMV796 LMV797 VOUT = 0.3V to 4.7V, RLOAD = 10 kΩ to V+/2 80 75 80 75 80 −0.3 -0.3 85 80 82 78 88 84 97 89 110 Conditions Min (Note 5) Typ (Note 4) 0.1 −1.0 −1.8 0.1 0.1 10 100 100 dB 98 4 4 V 1 25 1 100 Max (Note 5) Units ± 1.35 ± 1.65 mV µV/˚C pA fA dB dB 3 www.national.com LMV796/LMV797 5V Electrical Characteristics VOUT Output Swing High (Continued) 35 25 LMV796 LMV797 42 50 20 45 37 10 6 60 21 1.15 1.30 6.0 7.5 9.5 11.5 17 5.8 0.01 0.01 1.40 1.75 1.70 2.05 mA 75 82 65 71 75 78 80 83 65 67 mV from rail RLOAD = 2 kΩ to V+/2 RLOAD = 10 kΩ to V+/2 Output Swing Low RLOAD = 2 kΩ to V+/2 RLOAD = 10 kΩ to V+/2 IOUT Output Short Circuit Current Sourcing to V− VIN = 200 mV (Note 9) Sinking to V+ VIN = –200 mV (Note 9) IS Supply Current per Amplifier LMV796 LMV797 per channel SR GBWP en in THD+N Slew Rate Gain Bandwidth Product Input-Referred Voltage Noise Input-Referred Current Noise Total Harmonic Distortion + Noise f = 1 kHz f = 1 kHz f = 1 kHz, AV = 1, RLOAD = 600Ω AV = +1, Rising (10% to 90%) AV = +1, Falling (90% to 10%) mA V/µs MHz nV/ pA/ % Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see the Electrical Characteristics Tables. Note 2: Human Body Model is 1.5 kΩ in series with 100 pF. Machine Model is 0Ω in series with 200 pF Note 3: The maximum power dissipation is a function of TJ(MAX), θJA. The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) - TA)/θJA. All numbers apply for packages soldered directly onto a PC Board. Note 4: Typical values represent the parametric norm at the time of characterization. Note 5: Limits are 100% production tested at 25˚C. Limits over the operating temperature range are guaranteed through correlations using the statistical quality control (SQC) method. Note 6: Offset voltage average drift is determined by dividing the change in VOS by temperature change. Note 7: Positive current corresponds to current flowing into the device. Note 8: Input bias current and input offset current are guaranteed by design Note 9: The short circuit test is a momentary test, the short circuit duration is 1.5 ms. www.national.com 4 LMV796/LMV797 Connection Diagrams 5-Pin SOT23 8-Pin MSOP 20183501 Top View 20183502 Top View Ordering Information Package 5-Pin SOT23 8-Pin MSOP Part Number LMV796MF LMV796MFX LMV797MM LMV797MMX Package Marking AT3A AU3A Transport Media 1k Units Tape and Reel 3k Units Tape and Reel 1k Units Tape and Reel 3.5k Units Tape and Reel NSC Drawing MF05A MUA08A 5 www.national.com LMV796/LMV797 Typical Performance Characteristics Voltage = 5V, VCM = V /2. Supply Current vs. Supply Voltage (LMV796) + Unless otherwise specified, TA = 25˚C, V– = 0, V+ = Supply Supply Current vs. Supply Voltage (LMV797) 20183505 20183581 VOS vs. VCM VOS vs. VCM 20183509 20183551 VOS vs. VCM VOS vs. Supply Voltage 20183511 20183512 www.national.com 6 LMV796/LMV797 Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, V– = 0, V+ = Supply Voltage = 5V, VCM = V+/2. (Continued) Slew Rate vs. Supply Voltage Input Bias Current vs. VCM 20183529 20183562 Input Bias Current vs. VCM Sourcing Current vs. Supply Voltage 20183587 20183520 Sinking Current vs. Supply Voltage Sourcing Current vs. Output Voltage 20183519 20183550 7 www.national.com LMV796/LMV797 Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, V– = 0, V+ = Supply Voltage = 5V, VCM = V+/2. (Continued) Sinking Current vs. Output Voltage Positive Output Swing vs. Supply Voltage 20183554 20183517 Negative Output Swing vs. Supply Voltage Positive Output Swing vs. Supply Voltage 20183515 20183516 Negative Output Swing vs. Supply Voltage Positive Output Swing vs. Supply Voltage 20183514 20183518 www.national.com 8 LMV796/LMV797 Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, V– = 0, V+ = Supply Voltage = 5V, VCM = V+/2. (Continued) Negative Output Swing vs. Supply Voltage Input Referred Voltage Noise vs. Frequency 20183513 20183539 Overshoot and Undershoot vs. CLOAD THD+N vs. Peak-to-Peak Output Voltage (VOUT) 20183530 20183526 THD+N vs. Peak-to-Peak Output Voltage (VOUT) THD+N vs. Frequency 20183504 20183574 9 www.national.com LMV796/LMV797 Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, V– = 0, V+ = Supply Voltage = 5V, VCM = V+/2. (Continued) THD+N vs. Frequency Open Loop Gain and Phase with Capacitive Load 20183575 20183541 Open Loop Gain and Phase with Resistive Load Closed Loop Output Impedance vs. Frequency 20183573 20183532 Crosstalk Rejection Small Signal Transient Response, AV = +1 20183538 20183580 www.national.com 10 LMV796/LMV797 Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, V– = 0, V+ = Supply Voltage = 5V, VCM = V+/2. (Continued) Large Signal Transient Response, AV = +1 Small Signal Transient Response, AV = +1 20183537 20183533 Large Signal Transient Response, AV = +1 Phase Margin vs. Capacitive Load (Stability) 20183534 20183545 Phase Margin vs. Capacitive Load (Stability) Positive PSRR vs. Frequency 20183546 20183527 11 www.national.com LMV796/LMV797 Typical Performance Characteristics Unless otherwise specified, TA = 25˚C, V– = 0, V+ = Supply Voltage = 5V, VCM = V+/2. (Continued) Negative PSRR vs. Frequency CMRR vs. Frequency 20183528 20183556 Input Common Mode Capacitance vs. VCM 20183576 www.national.com 12 LMV796/LMV797 Application Information ADVANTAGES OF THE LMV796/LMV797 Wide Bandwidth at Low Supply Current The LMV796 and LMV797 are high performance op amps that provide a unity gain bandwidth of 17 MHz while drawing a low supply current of 1.15 mA. This makes them ideal for providing wideband amplification in portable applications. Low Input Referred Noise and Low Input Bias Current The LMV796/LMV797 have a very low input referred voltage at 1 kHz). A CMOS input stage noise density (5.8 nV/ ensures a small input bias current (100 fA) and low input ). This is very helpful in referred current noise (0.01 pA/ maintaining signal fidelity, and makes the LMV796 and LMV797 ideal for audio and sensor based applications. Low Supply Voltage The LMV796 and the LMV797 have performance guaranteed at 2.5V and 5V supply. The LMV796 family is guaranteed to be operational at all supply voltages between 2.0V and 5.5V, for ambient temperatures ranging from −40˚C to 125˚C, thus utilizing the entire battery lifetime. The LMV796 and LMV797 are also guaranteed to be operational at 1.8V supply voltage, for temperatures between 0˚C and 125˚C. This makes the LMV796 family ideal for usage in low-voltage commercial applications. RRO and Ground Sensing Rail-to-rail output swing provides maximum possible dynamic range at the output. This is particularly important when operating at low supply voltages. An innovative positive feedback scheme is used to boost the current drive capability of the output stage. This allows the LMV796 and the LMV797 to source more than 40 mA of current at 1.8V supply. This also limits the performance of the LMV796 family as comparators, and hence the usage of the LMV796 and the LMV797 in an open-loop configuration is not recommended. The input common-mode range includes the negative supply rail which allows direct sensing at ground in single supply operation. Small Size The small footprint of the LMV796 and the LMV797 package saves space on printed circuit boards, and enables the design of smaller electronic products, such as cellular phones, pagers, or other portable systems. Long traces between the signal source and the op amp make the signal path susceptible to noise. By using the physically smaller LMV796 or LMV797 package, the op amp can be placed closer to the signal source, reducing noise pickup and increasing signal integrity. CAPACITIVE LOAD TOLERANCE The LMV796 and LMV797 can directly drive 120 pF in unity-gain without oscillation. The unity-gain follower is the most sensitive configuration to capacitive loading. Direct capacitive loading reduces the phase margin of amplifiers. The combination of the amplifier’s output impedance and the capacitive load induces phase lag. This results in either an underdamped pulse response or oscillation. To drive a heavier capacitive load, the circuit in Figure 1 can be used. In Figure 1, the isolation resistor RISO and the load capacitor CL form a pole to increase stability by adding more phase margin to the overall system. The desired performance depends on the value of RISO. The bigger the RISO resistor value, the more stable VOUT will be. Increased RISO would, however, result in a reduced output swing and short circuit current. 20183561 FIGURE 1. Isolation of CL to Improve Stability INPUT CAPACITANCE AND FEEDBACK CIRCUIT ELEMENTS The LMV796 family has a very low input bias current (100 fA) and a low 1/f noise corner frequency (400 Hz), which makes it ideal for sensor applications. However, to obtain this performance a large CMOS input stage is used, which adds to the input capacitance of the op amp, CIN. Though this does not affect the DC and low frequency performance, at higher frequencies the input capacitance interacts with the input and the feedback impedances to create a pole, which results in lower phase margin and gain peaking. This can be controlled by being selective in the use of feedback resistors, as well as, by using a feedback capacitance, CF. For example, in the inverting amplifier shown in Figure 2, if CIN and CF are ignored and the open loop gain of the op amp is considered infinite then the gain of the circuit is −R2/R1. An op amp, however, usually has a dominant pole, which causes its gain to drop with frequency. Hence, this gain is only valid for DC and low frequency. To understand the effect of the input capacitance coupled with the non-ideal gain of the op amp, the circuit needs to be analyzed in the frequency domain using a Laplace transform. 20183564 FIGURE 2. Inverting Amplifier 13 www.national.com LMV796/LMV797 Application Information (Continued) For simplicity, the op amp is modeled as an ideal integrator with a unity gain frequency of A0 . Hence, its transfer function (or gain) in the frequency domain is A0/s. Solving the circuit equations in the frequency domain, ignoring CF for the moment, results in an expression for the gain shown in Equation (1). the circuit. Adding a capacitance of 2 pF removes the peak, while a capacitance of 5 pF creates a much lower pole and reduces the bandwidth excessively. (1) It can be inferred from the denominator of the transfer function that it has two poles, whose expressions can be obtained by solving for the roots of the denominator and are shown in Equation (2). 20183560 FIGURE 4. Gain Peaking Eliminated by CF (2) Equation (2) shows that as the values of R1 and R2 are increased, the magnitude of the poles, and hence the bandwidth of the amplifier, is reduced. This theory is verified by using different values of R1 and R2 in the circuit shown in Figure 1 and by comparing their frequency responses. In Figure 3 the frequency responses for three different values of R1 and R2 are shown. When both R1 and R2 are 1 kΩ, the response is flattest and widest; whereas, it narrows and peaks significantly when both their values are changed to 10 kΩ or 30 kΩ. So it is advisable to use lower values of R1 and R2 to obtain a wider and flatter response. Lower resistances also help in high sensitivity circuits since they add less noise. AUDIO PREAMPLIFIER WITH BAND PASS FILTERING With low input referred voltage noise, low supply voltage and current, and a low harmonic distortion, the LMV796 family is ideal for audio applications. Its wide unity gain bandwidth allows it to provide large gain for a wide range of frequencies and it can be used to design a preamplifier to drive a load of as low as 600Ω with less than 0.01% distortion. Two amplifier circuits are shown in Figure 5 and Figure 6. Figure 5 is an inverting amplifier, with a 10 kΩ feedback resistor, R2, and a 1kΩ input resistor, R1, and hence provides a gain of −10. Figure 6 is a non-inverting amplifier, using the same values of R1and R2, and provides a gain of 11. In either of these circuits, the coupling capacitor CC1 decides the lower frequency at which the circuit starts providing gain, while the feedback capacitor CF decides the frequency at which the gain starts dropping off. Figure 7 shows the frequency response of the inverting amplifier with different values of CF. 20183559 FIGURE 3. Gain Peaking Caused by Large R1, R2 A way of reducing the gain peaking is by adding a feedback capacitance CF in parallel with R2. This introduces another pole in the system and prevents the formation of pairs of complex conjugate poles which cause the gain to peak. Figure 4 shows the effect of CF on the frequency response of 20183565 FIGURE 5. Inverting Audio Preamplifier www.national.com 14 LMV796/LMV797 Application Information (Continued) high gain. A typical transimpedance amplifier is shown in Figure 8. The output voltage of the amplifier is given by the equation VOUT = −IINRF. Since the output swing of the amplifier is limited, RF should be selected such that all possible values of IIN can be detected. The LMV796/LMV797 have a large gain-bandwidth product (17 MHz), which enables high gains at wide bandwidths. A rail-to-rail output swing at 5.5V supply allows detection and amplification of a wide range of input currents. A CMOS input stage with negligible input current noise and low input voltage noise allows the LMV796/LMV797 to provide high fidelity amplification for wide bandwidths. These properties make the LMV796/LMV797 ideal for systems requiring wide-band transimpedance amplification. 20183566 FIGURE 6. Non-inverting Audio Preamplifier 20183569 FIGURE 8. Photodiode Transimpedance Amplifier As mentioned earlier, the following parameters are used to design a transimpedance amplifier: the amplifier gainbandwidth product, A0; the amplifier input capacitance, CCM; the photodiode capacitance, CD; the transimpedance gain required, RF; and the amplifier output swing. Once a feasible RF is selected using the amplifier output swing, these numbers can be used to design an amplifier with the desired transimpedance gain and a maximally flat frequency response. An essential component for obtaining a maximally flat response is the feedback capacitor, CF. The capacitance seen at the input of the amplifier, CIN, combined with the feedback capacitor, RF, generate a phase lag which causes gainpeaking and can destabilize the circuit. CIN is usually just the sum of CD and CCM. The feedback capacitor CF creates a pole, fP in the noise gain of the circuit, which neutralizes the zero in the noise gain, fZ, created by the combination of RF and CIN. If properly positioned, the noise gain pole created by CF can ensure that the slope of the gain remains at 20 dB/decade till the unity gain frequency of the amplifier is reached, thus ensuring stability. As shown in Figure 9, fP is positioned such that it coincides with the point where the noise gain intersects the op amp’s open loop gain. In this case, fP is also the overall −3 dB frequency of the transimpedance amplifier. The value of CF needed to make it so is given by Equation (3). A larger value of CF causes excessive reduction of bandwidth, while a smaller value fails to prevent gain peaking and instability. 20183558 FIGURE 7. Frequency Response of the Inverting Audio Preamplifier TRANSIMPEDANCE AMPLIFIER CMOS input op amps are often used in transimpedance applications as they have an extremely high input impedance. A transimpedance amplifier converts a small input current into a voltage. This current is usually generated by a photodiode. The transimpedance gain, measured as the ratio of the output voltage to the input current, is expected to be large and wide-band. Since the circuit deals with currents in the range of a few nA, low noise performance is essential. The LMV796/LMV797 are CMOS input op amps providing wide bandwidth and low noise performance, and are hence ideal for transimpedance applications. Usually, a transimpedance amplifier is designed on the basis of the current source driving the input. A photodiode is a very common capacitive current source, which requires transimpedance gain for transforming its miniscule current into easily detectable voltages. The photodiode and the amplifier’s gain are selected with respect to the speed and accuracy required of the circuit. A faster circuit would require a photodiode with lesser capacitance and a faster amplifier. A more sensitive circuit would require a sensitive photodiode and a 15 www.national.com LMV796/LMV797 Application Information (Continued) formula in Equation (3), since the parasitic capacitance of the board and the feedback resistor RF had to be accounted for. TABLE 1. (3) Transimpedance, ATI 470000 470000 470000 47000 47000 47000 CIN CF −3 dB Frequency 350 kHz 250 kHz 150 kHz 1.5 MHz 1 MHz 700 kHz 50 pF 1.5 pF 100 pF 2.0 pF 200 pF 3.0 pF 50 pF 4.5 pF 100 pF 6.0 pF 200 pF 9.0 pF 20183584 FIGURE 9. CF Selection for Stability Calculating CF from Equation (3) can sometimes return unreasonably small values ( < 1 pF), especially for high speed applications. In these cases, it is often more practical to use the circuit shown in Figure 10 in order to allow more reasonable values. In this circuit, the capacitance CF' is (1+ RB/RA) times the effective feedback capacitance, CF. A larger capacitor can now be used in this circuit to obtain a smaller effective capacitance. For example, if a CF of 0.5 pF is needed, while only a 5 pF capacitor is available, RB and RA can be selected such that RB/RA = 9. This would convert a CF' of 5 pF into a CF of 0.5 pF. This relationship holds as long as RA