NCP1421LEDEVB

NCP1421 Boost Converter Sync-Rect, PFM, DC-DC, True-Cutoff, Ring-Killer 600 mA NCP1421 is a monolithic micropower high−frequency step−up switching converter IC specially designed for battery−operated hand−held electronic products up to 600 mA loading. It integrates Sync−Rect to improve efficiency and to eliminate the external Schottky Diode. High switching frequency (up to 1.2 MHz) allows for a low profile, small−sized inductor and output capacitor to be used. When the device is disabled, the internal conduction path from LX or BAT to OUT is fully blocked and the OUT pin is isolated from the battery. This True−Cutoff function reduces the shutdown current to typically only 50 nA. Ring−Killer is also integrated to eliminate the high−frequency ringing in discontinuous conduction mode. In addition to the above, Low−Battery Detector, Logic−Controlled Shutdown, Cycle−by−Cycle Current Limit and Thermal Shutdown provide value−added features for various battery−operated applications. With all these functions on, the quiescent supply current is typically only 8.5 A. This device is available in the compact and low profile Micro8t package. http://onsemi.com MARKING DIAGRAM 8 Micro8 DM SUFFIX CASE 846A 8 1 1421 AYWG G 1 1421= Specific Device Code A = Assembly Location Y = Year W = Work Week G = Pb−Free Package (Note: Microdot may be in either location) Features • High Efficiency: 94% for 3.3 V Output at 200 mA from 2.5 V Input • • • • • • • • • • • • • • 88% for 3.3 V Output at 500 mA from 2.5 V Input High Switching Frequency, up to 1.2 MHz (not hitting current limit) Output Current up to 600 mA at VIN = 2.5 V and VOUT = 3.3 V True−Cutoff Function Reduces Device Shutdown Current to typically 50 nA Anti−Ringing Ring−Killer for Discontinuous Conduction Mode High Accuracy Reference Output, 1.20 V $1.5%, can Supply 2.5 mA Loading Current when VOUT > 3.3 V Low Quiescent Current of 8.5 A Integrated Low−Battery Detector Open Drain Low−Battery Detector Output 1.0 V Startup at No Load Guaranteed Output Voltage from 1.5 V to 5.0 V Adjustable 1.5 A Cycle−by−Cycle Current Limit Multi−function Logic−Controlled Shutdown Pin On Chip Thermal Shutdown with Hysteresis Pb−Free Package is Available Typical Applications • • • • • • PIN CONNECTIONS FB 1 8 LBI/EN LBO 2 3 7 6 REF 4 5 OUT LX GND BAT (Top View) ORDERING INFORMATION Device Package Shipping† NCP1421DMR2 Micro8 4000 Tape & Reel Micro8 (Pb−Free) 4000 Tape & Reel NCP1421DMR2G †For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging Specifications Brochure, BRD8011/D. Personal Digital Assistants (PDA) Handheld Digital Audio Products Camcorders and Digital Still Cameras Hand−held Instruments Conversion from one to two Alkaline, NiMH, NiCd Battery Cells to 3.0−5.0 V or one Lithium−ion cells to 5.0 V White LED Flash for Digital Cameras © Semiconductor Components Industries, LLC, 2009 August, 2019 − Rev. 7 1 Publication Order Number: NCP1421/D NCP1421 VBAT M3 ZLC 0.5 V + - Chip Enable + VDD 20 mV _ZCUR _TSDON M2 VDD 8 M1 _MAINSW2ON 6 GND _CEN + - VDD _PFM _MAINSWOFD PFM _SYNSW2ON REF 4 GND _SYNSWOFD Voltage Reference _VREFOK _ILIM + - 1.20 V LBI/EN 2 VOUT SENSEFETt _MSON FB 1 LX 7 OUT TRUE CUTOFF CONTROL CONTROL LOGIC BAT 5 RSENSE + LBO 3 GND + - GND Figure 1. Detailed Block Diagram http://onsemi.com 2 GND NCP1421 PIN FUNCTION DESCRIPTIONS Pin Symbol 1 FB 2 LBI/EN Low−Battery Detector Input and IC Enable. With this pin pulled down below 0.5 V, the device is disabled and enters the shutdown mode. 3 LBO Open−Drain Low−Battery Detector Output. Output is LOW when VLBI is < 1.20 V. LBO is high impedance in shutdown mode. 4 REF 1.20 V Reference Voltage Output, bypass with 1.0 F capacitor. If this pin is not loaded, bypass with 300 nF capacitor; this pin can be loaded up to 2.5 mA @ VOUT = 3.3 V. 5 BAT Battery input connection for internal ring−killer. 6 GND Ground. 7 LX 8 OUT Description Output Voltage Feedback Input. N−Channel and P−Channel Power MOSFET drain connection. Power Output. OUT also provides bootstrap power to the device. MAXIMUM RATINGS (TC = 25°C unless otherwise noted.) Rating Symbol Value Unit VOUT −0.3, 5.5 V VIO −0.3, 5.5 V PD RJA 520 240 mW _C/W Operating Junction Temperature Range TJ −40 to +150 _C Operating Ambient Temperature Range TA −40 to +85 _C Storage Temperature Range Tstg −55 to +150 _C Power Supply (Pin 8) Input/Output Pins (Pin 1−5, Pin 7) Thermal Characteristics Micro8 Plastic Package Thermal Resistance Junction−to−Air Maximum ratings are those values beyond which device damage can occur. Maximum ratings applied to the device are individual stress limit values (not normal operating conditions) and are not valid simultaneously. If these limits are exceeded, device functional operation is not implied, damage may occur and reliability may be affected. 1. This device contains ESD protection and exceeds the following tests: Human Body Model (HBM) ±2.0 kV per JEDEC standard: JESD22−A114. *Except OUT pin, which is 1k V. Machine Model (MM) ±200 V per JEDEC standard: JESD22−A115. *Except OUT pin, which is 100 V. 2. The maximum package power dissipation limit must not be exceeded. TJ(max) * TA PD + RJA 3. Latchup Current Maximum Rating: ±150 mA per JEDEC standard: JESD78. 4. Moisture Sensitivity Level: MSL 1 per IPC/JEDEC standard: J−STD−020A. http://onsemi.com 3 NCP1421 ELECTRICAL CHARACTERISTICS (VOUT = 3.3 V, TA = 25°C for typical value, −40°C v TA v 85°C for min/max values unless otherwise noted.) Symbol Min Typ Max Unit VIN 1.0 − 5.0 V VOUT 1.5 − 5.0 V Reference Voltage (VOUT = 3.3 V, ILOAD = 0 A, CREF = 200 nF, TA = 25°C) VREF_NL 1.183 1.200 1.217 V Reference Voltage (VOUT = 3.3 V, ILOAD = 0 A, CREF = 200 nF, TA = −40°C to 85°C) VREF_NL 1.174 − 1.220 V Reference Voltage Temperature Coefficient TCVREF − 0.03 − mV/°C IREF − 2.5 − mA Reference Voltage Load Regulation (VOUT = 3.3 V, ILOAD = 0 to 100 A, CREF = 1.0 F) VREF_LOAD − 0.05 1.0 mV Reference Voltage Line Regulation (VOUT from 1.5 V to 5.0 V, CREF = 1.0 F) VREF_LINE − 0.05 1.0 mV/V FB Input Threshold (ILOAD = 0 mA, TA = 25°C) VFB 1.192 1.200 1.208 V FB Input Threshold (ILOAD = 0 mA, TA = −40°C to 85°C) VFB 1.184 − 1.210 V LBI Input Threshold (ILOAD = 0 mA, TA= −40_C to 85_C) VLBI 1.162 1.230 V LBI Input Threshold (TA = 25_C) VLBI 1.182 1.200 1.218 V Internal NFET ON−Resistance RDS(ON)_N − 0.3 −  Internal PFET ON−Resistance RDS(ON)_P − 0.3 −  ILIM − 1.5 − A Operating Current into BAT (VBAT = 1.8 V, VFB = 1.8 V, VLX = 1.8 V, VOUT = 3.3 V) IQBAT − 1.3 3 A Operating Current into OUT (VFB = 1.4 V, VOUT = 3.3 V) IQ − 8.5 14 A LX Switch MAX. ON−Time (VFB = 1.0 V, VOUT = 3.3 V, TA = 25_C) tON 0.46 0.72 1.15 s LX Switch MIN. OFF−Time (VFB = 1.0 V, VOUT = 3.3 V, TA = 25_C) tOFF − 0.12 0.22 s FB Input Current IFB − 1.0 50 nA True−Cutoff Current into BAT (LBI/EN = GND, VOUT = 0, VIN = 3.3 V, LX = 3.3 V) IBAT − 50 − nA RBAT_LX − 100 −  ILBI − 1.5 50 nA VLBO_L − − 0.2 V TSS − 1.5 20 ms EN Pin Shutdown Threshold (TA = 25°C) VSHDN 0.35 0.5 0.67 V Thermal Shutdown Temperature (Note 7) TSHDN − − 145 °C Thermal Shutdown Hysteresis (Note 7) TSDHYS − 30 − °C Characteristic Operating Voltage Output Voltage Range Reference Voltage Load Current (VOUT = 3.3 V, VREF = VREF_NL "1.5% CREF = 1.0 F) (Note 5) LX Switch Current Limit (N−FET) (Note 7) BAT−to−LX Resistance (VFB = 1.4 V, VOUT = 3.3 V) (Note 7) LBI/EN Input Current LBO Low Output Voltage (VLBI = 0, ISINK = 1.0 mA) Soft−Start Time (VIN = 2.5 V, VOUT = 5.0 V, CREF = 200 nF) (Note 6) 5. Loading capability increases with VOUT. 6. Design guarantee, value depends on voltage at VOUT. 7. Values are design guaranteed. http://onsemi.com 4 NCP1421 TYPICAL OPERATING CHARACTERISTICS 1.220 1.200 REFERENCE VOLTAGE, VREF/V REFERENCE VOLTAGE, VREF/V 1.210 1.220 VIN = 1.5 V VOUT = 3.3 V L = 10 H CIN = 22 F COUT = 22 F CREF = 1.0 F TA = 25_C VIN = 2.0 V VIN = 2.5 V 1.190 1.180 1 10 100 1000 CREF = 200 nF IREF = 0 mA TA = 25°C 1.210 1.200 1.190 1.180 1.5 2.5 3 3.5 4 4.5 5 OUTPUT CURRENT, ILOAD/mA VOLTAGE AT OUT PIN, VOUT/V Figure 2. Reference Voltage vs. Output Current Figure 3. Reference Voltage vs. Voltage at OUT Pin SWITCH ON RESISTANCE, RDS(ON)/ REFERENCE VOLTAGE, VREF/V 1.205 1.200 1.195 1.190 VOUT = 3.3 V CREF = 200 nF IREF = 0 mA 1.185 1.180 −40 −20 0 20 40 60 80 100 0.6 VOUT = 3.3 V 0.5 0.4 P−FET (M2) 0.3 0.2 N−FET (M1) 0.1 0.0 −40 AMBIENT TEMPERATURE, TA/°C 0 20 40 60 80 100 Figure 5. Switch ON Resistance vs. Temperature 1.0 MINIMUM STARTUP BATTERY VOLTAGE, VBATT/V 1.6 0.9 0.8 0.7 0.6 0.5 −40 −20 AMBIENT TEMPERATURE, TA/°C Figure 4. Reference Voltage vs. Temperature LX SWITCH MAXIMUM, ON TIME, tON/S 2 −20 0 20 40 60 80 1.4 1.1 0.9 TA = 25°C 0.6 0 100 AMBIENT TEMPERATURE, TA/°C 50 100 150 200 250 OUTPUT LOADING CURRENT, ILOAD/mA Figure 6. LX Switch Max. ON Time vs. Temperature Figure 7. Minimum Startup Battery Voltage vs. Loading Current http://onsemi.com 5 NCP1421 100 90 90 80 VIN = 3.3 V VOUT = 5.0 V L = 12 H CIN = 22 F COUT = 22 F TA = 25_C 70 60 10 100 80 50 1 1000 100 OUTPUT LOADING CURRENT, ILOAD/mA Figure 8. Efficiency vs. Load Current Figure 9. Efficiency vs. Load Current 100 90 90 80 VIN = 2.5 V VOUT = 5.0 V L = 6.8 H CIN = 22 F COUT = 22 F TA = 25_C 70 60 10 100 VIN = 2.0 V VOUT = 3.3 V L = 10 H CIN = 22 F COUT = 22 F TA = 25_C 70 60 50 1 1000 10 100 OUTPUT LOADING CURRENT, ILOAD/mA OUTPUT LOADING CURRENT, ILOAD/mA Figure 10. Efficiency vs. Load Current Figure 11. Efficiency vs. Load Current 100 90 90 80 VIN = 1.5 V VOUT = 5.0 V L = 2.2 H CIN = 22 F COUT = 22 F TA = 25_C 70 60 10 VIN = 1.5 V VOUT = 1.8 V L = 2.2 H CIN = 22 F COUT = 22 F TA = 25_C 70 50 1 1000 10 100 OUTPUT LOADING CURRENT, ILOAD/mA OUTPUT LOADING CURRENT, ILOAD/mA Figure 12. Efficiency vs. Load Current Figure 13. Efficiency vs. Load Current http://onsemi.com 6 1000 80 60 100 1000 80 100 50 1 10 OUTPUT LOADING CURRENT, ILOAD/mA 100 50 1 VIN = 2.5 V VOUT = 3.3 V L = 10 H CIN = 22 F COUT = 22 F TA = 25_C 70 60 EFFICIENCY/% EFFICIENCY/% 50 1 EFFICIENCY/% EFFICIENCY/% 100 EFFICIENCY/% EFFICIENCY/% TYPICAL OPERATING CHARACTERISTICS 1000 NCP1421 TYPICAL OPERATING CHARACTERISTICS 10 OUTPUT VOLTAGE CHANGE/% OUTPUT VOLTAGE CHANGE/% 10 5 VIN = 2.5 V 0 VOUT = 3.3 V L = 5.6 H CIN = 22 F COUT = 22 F TA = 25_C −5 VIN = 2.0 V −10 10 100 5 0 VIN = 3.3 V VOUT = 5.0 V L = 5.6 H CIN = 22 F COUT = 22 F TA = 25_C −5 −10 10 1000 OUTPUT LOADING CURRENT, ILOAD/mA VIN = 1.5 V VIN = 2.5 V 100 1000 OUTPUT LOADING CURRENT, ILOAD/mA Figure 15. Output Voltage Change vs. Load Current Figure 14. Output Voltage Change vs. Load Current RIPPLE VOLTAGE, VRIPPLE/mVp−p 50 VIN = 2.5 V VOUT = 3.3 V L = 6.8 H CIN = 22 F COUT = 22 F TA = 25_C 40 30 500 mA 20 300 mA 10 100 mA 0 1.5 1.7 1.9 2.1 2.3 2.5 Upper Trace: Voltage at LBI Pin, 1.0 V/Division Lower Trace: Voltage at LBO Pin, 1.0 V/Division BATTERY INPUT VOLTAGE, VBATT/V NO LOAD OPERATING CURRENT, IBATT/A Figure 16. Battery Input Voltage vs. Output Ripple Voltage Figure 17. Low Battery Detect 15 12.5 10 7.5 VIN = 2.5 V VOUT = 5.0 V ILOAD = 10 mA 5.0 2.5 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 Upper Trace: Input Voltage Waveform, 1.0 V/Division Lower Trace: Output Voltage Waveform, 2.0 V/Division INPUT VOLTAGE AT OUT PIN, VOUT/V Figure 18. No Load Operating Current vs. Input Voltage at OUT Pin Figure 19. Startup Transient Response http://onsemi.com 7 NCP1421 TYPICAL OPERATING CHARACTERISTICS (VIN = 2.5 V, VOUT = 3.3 V, ILOAD = 50 mA; L = 5.6 H, COUT = 22 F) Upper Trace: Output Voltage Ripple, 20 mV/Division Lower Trace: Voltage at Lx pin, 1.0 V/Division (VIN = 2.5 V, VOUT = 3.3 V, ILOAD = 500 mA; L = 5.6 H, COUT = 22F) Upper Trace: Output Voltage Ripple, 20 mV/Division Lower Trace: Voltage at LX pin, 1.0 V/Division Figure 20. Discontinuous Conduction Mode Switching Waveform Figure 21. Continuous Conduction Mode Switching Waveform (VIN = 1.5 V to 2.5 V; L = 5.6 H, COUT = 22F, ILOAD = 100 mA) Upper Trace: Output Voltage Ripple, 100 mV/Division Lower Trace: Battery Voltage, VIN, 1.0 V/Division (VIN = 1.5 V to 2.5 V; L = 5.6 H, COUT = 22F, ILOAD = 100 mA) Upper Trace: Output Voltage Ripple, 100 mV/Division Lower Trace: Battery Voltage, VIN, 1.0 V/Division Figure 22. Line Transient Response for VOUT = 3.3 V Figure 23. Line Transient Response For VOUT = 5.0 V (VOUT = 3.3 V, ILOAD = 50 mA to 500 mA; L = 5.6 H, COUT = 22 F) Upper Trace: Output Voltage Ripple, 50 mV/Division Lower Trace: Load Current, ILOAD, 500 mA/Division (VOUT = 5.0 V, ILOAD = 50 mA to 500 mA; L = 5.6 H, COUT = 22 F) Upper Trace: Output Voltage Ripple, 100 mV/Division Lower Trace: Load Current, ILOAD, 500 mA/Division Figure 24. Load Transient Response For VIN = 2.5 V Figure 25. Load Transient Response For VIN = 3.0 V http://onsemi.com 8 NCP1421 DETAILED OPERATION DESCRIPTION of dead time is introduced to make sure M1 is completely turned OFF before M2 is being turned ON. The previously mentioned situation occurs when the regulator is operating in CCM, M2 is being turned OFF, M1 is just turned ON, and M2 is not being completely turned OFF. A dead time is also needed to make sure M2 is completely turned OFF before M1 is being turned ON. As coil current is dropped to zero when the regulator is operating in DCM, M2 should be OFF. If this does not occur, the reverse current flows from the output bulk capacitor through M2 and the inductor to the battery input, causing damage to the battery. The ZLC comparator comes with fixed offset voltage to switch M2 OFF before any reverse current builds up. However, if M2 is switched OFF too early, large residue coil current flows through the body diode of M2 and increases conduction loss. Therefore, determination of the offset voltage is essential for optimum performance. With the implementation of the synchronous rectification scheme, efficiency can be as high as 94% with this device. NCP1421 is a monolithic micropower high−frequency step−up voltage switching converter IC specially designed for battery operated hand−held electronic products up to 600 mA loading. It integrates a Synchronous Rectifier to improve efficiency as well as to eliminate the external Schottky diode. High switching frequency (up to 1.2 MHz) allows for a low profile inductor and output capacitor to be used. Low−Battery Detector, Logic−Controlled Shutdown, and Cycle−by−Cycle Current Limit provide value−added features for various battery−operated applications. With all these functions ON, the quiescent supply current is typically only 8.5 A. This device is available in a compact Micro8 package. PFM Regulation Scheme From the simplified functional diagram (Figure 1), the output voltage is divided down and fed back to pin 1 (FB). This voltage goes to the non−inverting input of the PFM comparator whereas the comparator’s inverting input is connected to the internal voltage reference, REF. A switching cycle is initiated by the falling edge of the comparator, at the moment the main switch (M1) is turned ON. After the maximum ON−time (typically 0.72 S) elapses or the current limit is reached, M1 is turned OFF and the synchronous switch (M2) is turned ON. The M1 OFF time is not less than the minimum OFF−time (typically 0.12 S), which ensures complete energy transfer from the inductor to the output capacitor. If the regulator is operating in Continuous Conduction Mode (CCM), M2 is turned OFF just before M1 is supposed to be ON again. If the regulator is operating in Discontinuous Conduction Mode (DCM), which means the coil current will decrease to zero before the new cycle starts, M1 is turned OFF as the coil current is almost reaching zero. The comparator (ZLC) with fixed offset is dedicated to sense the voltage drop across M2 as it is conducting; when the voltage drop is below the offset, the ZLC comparator output goes HIGH and M2 is turned OFF. Negative feedback of closed−loop operation regulates voltage at pin 1 (FB) equal to the internal reference voltage (1.20 V). Cycle−by−Cycle Current Limit In Figure 1, a SENSEFET is used to sample the coil current as M1 is ON. With that sample current flowing through a sense resistor, a sense−voltage is developed. The threshold detector (ILIM) detects whether the sense−voltage is higher than the preset level. If the sense voltage is higher than the present level, the detector output notifies the Control Logic to switch OFF M1, and M1 can only be switched ON when the next cycle starts after the minimum OFF−time (typically 0.12 S). With proper sizing of the SENSEFET and sense resistor, the peak coil current limit is typically set at 1.5 A. Voltage Reference The voltage at REF is typically set at 1.20 V and can output up to 2.5 mA with load regulation ±2% at VOUT equal to 3.3 V. If VOUT is increased, the REF load capability can also be increased. A bypass capacitor of 200 nF is required for proper operation when REF is not loaded. If REF is loaded, a 1.0 F capacitor at the REF pin is needed. Synchronous Rectification The Synchronous Rectifier is used to replace the Schottky Diode to reduce the conduction loss contributed by the forward voltage of the Schottky Diode. The Synchronous Rectifier is normally realized by powerFET with gate control circuitry that incorporates relatively complicated timing concerns. As the main switch (M1) is being turned OFF and the synchronous switch M2 is just turned ON with M1 not being completely turned OFF, current is shunt from the output bulk capacitor through M2 and M1 to ground. This power loss lowers overall efficiency and possibly damages the switching FETs. As a general practice, a certain amount True−Cutoff The NCP1421 has a True−Cutoff function controlled by the multi−function pin LBI/EN (pin 2). Internal circuitry can isolate the current through the body diode of switch M2 to load. Thus, it can eliminate leakage current from the battery to load in shutdown mode and significantly reduce battery current consumption during shutdown. The shutdown function is controlled by the voltage at pin 2 (LBI/EN). When pin 2 is pulled to lower than 0.3 V, the controller enters shutdown mode. In shutdown mode, when switches M1 and M2 are both switched OFF, the internal http://onsemi.com 9 NCP1421 reference voltage of the controller is disabled and the controller typically consumes only 50 nA of current. If the pin 2 voltage is raised to higher than 0.5 V (for example, by a resistor connected to VIN), the IC is enabled again, and the internal circuit typically consumes 8.5 A of current from the OUT pin during normal operation. (LBI/EN) is at a voltage (defined by a resistor divider from the battery voltage) lower than the internal reference voltage of 1.20 V, the comparator output turns on a 50  low side switch. It pulls down the voltage at pin 3 (LBO) which requires a hundred to a thousand k of external pull−high resistance. If the pin 2 voltage is higher than 1.20 V + 30 mV, the comparator output turns off the 50 low side switch. When this occurs, pin 3 becomes high impedance and its voltage is pulled high again. Low−Battery Detection A comparator with 30 mV hysteresis is applied to perform the low−battery detection function. When pin 2 APPLICATIONS INFORMATION Output Voltage Setting voltage/current waveforms. The currents flowing into and out of the capacitors multiply with the Equivalent Series Resistance (ESR) of the capacitor to produce ripple voltage at the terminals. During the Syn−Rect switch−off cycle, the charges stored in the output capacitor are used to sustain the output load current. Load current at this period and the ESR combine and reflect as ripple at the output terminals. For all cases, the lower the capacitor ESR, the lower the ripple voltage at output. As a general guideline, low ESR capacitors should be used. Ceramic capacitors have the lowest ESR, but low ESR tantalum capacitors can also be used as an alternative. A typical application circuit is shown in Figure 26. The output voltage of the converter is determined by the external feedback network comprised of R1 and R2. The relationship is given by: VOUT + 1.20 V ǒ1 ) R1 Ǔ R2 where R1 and R2 are the upper and lower feedback resistors, respectively. Low Battery Detect Level Setting The Low Battery Detect Voltage of the converter is determined by the external divider network that is comprised of R3 and R4. The relationship is given by: VLB + 1.20 V PCB Layout Recommendations Good PCB layout plays an important role in switching mode power conversion. Careful PCB layout can help to minimize ground bounce, EMI noise, and unwanted feedback that can affect the performance of the converter. Hints suggested below can be used as a guideline in most situations. ǒ1 ) R3 Ǔ R4 where R3 and R4 are the upper and lower divider resistors respectively. Inductor Selection Grounding The NCP1421 is tested to produce optimum performance with a 5.6 H inductor at VIN = 2.5 V and VOUT = 3.3 V, supplying an output current up to 600 mA. For other input/output requirements, inductance in the range 3 H to 10 H can be used according to end application specifications. Selecting an inductor is a compromise between output current capability, inductor saturation limit, and tolerable output voltage ripple. Low inductance values can supply higher output current but also increase the ripple at output and reduce efficiency. On the other hand, high inductance values can improve output ripple and efficiency; however, it is also limited to the output current capability at the same time. Another parameter of the inductor is its DC resistance. This resistance can introduce unwanted power loss and reduce overall efficiency. The basic rule is to select an inductor with the lowest DC resistance within the board space limitation of the end application. In order to help with the inductor selection, reference charts are shown in Figure 27 and 28. A star−ground connection should be used to connect the output power return ground, the input power return ground, and the device power ground together at one point. All high−current paths must be as short as possible and thick enough to allow current to flow through and produce insignificant voltage drop along the path. The feedback signal path must be separated from the main current path and sense directly at the anode of the output capacitor. Components Placement Power components (i.e., input capacitor, inductor and output capacitor) must be placed as close together as possible. All connecting traces must be short, direct, and thick. High current flowing and switching paths must be kept away from the feedback (FB, pin 1) terminal to avoid unwanted injection of noise into the feedback path. Feedback Network Feedback of the output voltage must be a separate trace detached from the power path. The external feedback network must be placed very close to the feedback (FB, pin 1) pin and sense the output voltage directly at the anode of the output capacitor. Capacitors Selection In all switching mode boost converter applications, both the input and output terminals see impulsive http://onsemi.com 10 NCP1421 TYPICAL APPLICATION CIRCUIT VIN C1 22 F R2 200 k C4 10 p* R3 220 k R1 350 k 1 2 Shutdown Open Drain Input 3 R4 330 k Low Battery Open Drain Output 4 L 6.5 H NCP1421 FB OUT LBI/EN LX LBO GND REF BAT 8 7 6 C2 22 F + VOUT =3.3 V 500 mA 5 C3 200 nF *Optional Figure 26. Typical Application Schematic for 2 Alkaline Cells Supply GENERAL DESIGN PROCEDURES Switching mode converter design is considered a complicated process. Selecting the right inductor and capacitor values can allow the converter to provide optimum performance. The following is a simple method based on the basic first−order equations to estimate the inductor and capacitor values for NCP1421 to operate in Continuous Conduction Mode (CCM). The set component values can be used as a starting point to fine tune the application circuit performance. Detailed bench testing is still necessary to get the best performance out of the circuit. Design Parameters: VIN = 1.8 V to 3.0 V, Typical 2.4 V VOUT = 3.3 V IOUT = 500 mA (600 mA max) VLB = 2.0 V VOUT−RIPPLE = 45 mVp−p at IOUT = 500 mA Determine the Steady State Duty Ratio, D, for typical VIN. The operation is optimized around this point: VOUT + 1 1*D VIN D+1* Determine the average inductor current, ILAVG, at maximum IOUT: 500 mA I ILAVG + OUT + + 688 mA 1 * 0.273 1*D Determine the peak inductor ripple current, IRIPPLE−P, and calculate the inductor value: Assume IRIPPLE−P is 20% of ILAVG. The inductance of the power inductor can be calculated as follows: L+ Calculate the feedback network: Select R2 = 200 k R1 + R2 3.3 V ǒ1.20 * 1Ǔ + 350 k V VOUT−RIPPLE = 40 mVP−P at IOUT = 500 mA IOUT tON COUT u VOUT*RIPPLE * IOUT Calculate the Low Battery Detect divider: VLB = 2.0 V Select R4 = 330 k ǒ VLB R3 + R4 *1 VREF R3 + 300 k 2.4 V 0.75 S VIN tON + + 6.5 H 2 IRIPPLE*P 2 (137.6 mA) A standard value of 6.5 H is selected for initial trial. Determine the output voltage ripple, VOUT−RIPPLE, and calculate the output capacitor value: ǒVVOUT * 1Ǔ REF R1 + 200 k 2.4 V VIN +1* + 0.273 VOUT 3.3 V ESRCOUT where tON = 0.75 uS and ESRCOUT = 0.05 , 500 mA 0.75 S COUT u + 18.75 F 45 mV * 500 mA 0.05  Ǔ 2.0 V ǒ1.20 * 1Ǔ + 220 k V http://onsemi.com 11 NCP1421 is required, the capacitor C4 can be used as external compensation adjustment to improve system dynamics. In order to provide an easy way for customers to select external parts for NCP1421 in different input voltage and output current conditions, values of inductance and capacitance are suggested in Figure 27, 28 and 29. 16 21 14 18 INDUCTOR VALUE (H) INDUCTOR VALUE (H) From the previous calculations, you need at least 18.75 F in order to achieve the specified ripple level at the conditions stated. Practically, a capacitor that is one level larger is used to accommodate factors not taken into account in the calculations. Therefore, a capacitor value of 22 F is selected. The NCP1421 is internally compensated for most applications, but in case additional compensation 12 10 8 IOUT = 500 mA 6 4 15 12 9 6 3 2 0 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 IOUT = 500 mA 0 1.6 3.0 1.9 2.2 2.5 2.8 3.1 3.4 3.7 4.0 INPUT VOLTAGE (V) INPUT VOLTAGE (V) Figure 27. Suggested Inductance of VOUT = 3.3 V Figure 28. Suggested Inductance of VOUT = 5.0 V 25 40 VOUT−RIPPLE = 40 mV 33 30 25 VOUT−RIPPLE = 45 mV 20 50 15 VOUT−RIPPLE = 50 mV 10 100 CAPACITOR ESR (m) CAPACITOR VALUE (F) 35 5 0 200 250 300 350 400 450 500 550 600 OUTPUT CURRENT (mA) Figure 29. Suggested Capacitance for Output Capacitor Table 1. Suggestions for Passive Components Output Current Inductors Capacitors 500 mA Sumida CR43, CR54,CDRH6D28 series Panasonic ECJ series Kemet TL494 series 250 mA Sumida CR32 series Panasonic ECJ series Kemet TL494 series Micro8 is a trademark of International Rectifier. SENSEFET is a trademark of Semiconductor Components Industries, LLC. http://onsemi.com 12 MECHANICAL CASE OUTLINE PACKAGE DIMENSIONS Micro8 CASE 846A−02 ISSUE K DATE 16 JUL 2020 SCALE 2:1 GENERIC MARKING DIAGRAM* 8 XXXX AYWG G 1 XXXX A Y W G = Specific Device Code = Assembly Location = Year = Work Week = Pb−Free Package (Note: Microdot may be in either location) *This information is generic. Please refer to device data sheet for actual part marking. Pb−Free indicator, “G” or microdot “G”, may or may not be present. Some products may not follow the Generic Marking. DOCUMENT NUMBER: DESCRIPTION: 98ASB14087C MICRO8 STYLE 1: PIN 1. 2. 3. 4. 5. 6. 7. 8. SOURCE SOURCE SOURCE GATE DRAIN DRAIN DRAIN DRAIN STYLE 2: PIN 1. 2. 3. 4. 5. 6. 7. 8. SOURCE 1 GATE 1 SOURCE 2 GATE 2 DRAIN 2 DRAIN 2 DRAIN 1 DRAIN 1 STYLE 3: PIN 1. 2. 3. 4. 5. 6. 7. 8. N-SOURCE N-GATE P-SOURCE P-GATE P-DRAIN P-DRAIN N-DRAIN N-DRAIN Electronic versions are uncontrolled except when accessed directly from the Document Repository. Printed versions are uncontrolled except when stamped “CONTROLLED COPY” in red. PAGE 1 OF 1 ON Semiconductor and are trademarks of Semiconductor Components Industries, LLC dba ON Semiconductor or its subsidiaries in the United States and/or other countries. ON Semiconductor reserves the right to make changes without further notice to any products herein. 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