DATA SHEET
www.onsemi.com
NCP1937
This combination IC integrates power factor correction (PFC) and
quasi−resonant flyback functionality necessary to implement a
compact and highly efficient Switched Mode Power Supply for an
adapter application.
The PFC stage exhibits near−unity power factor while operating in a
Critical Conduction Mode (CrM) with a maximum frequency clamp.
The circuit incorporates all the features necessary for building a robust
and compact PFC stage while minimizing the number of external
components.
The quasi−resonant current−mode flyback stage features a
proprietary valley−lockout circuitry, ensuring stable valley switching.
This system works down to the 4th valley and toggles to a frequency
foldback mode with a minimum frequency clamp beyond the 4th
valley to eliminate audible noise. Skip mode operation allows
excellent efficiency in light load conditions while consuming very low
standby power consumption.
Common General Features
• Wide VCC Range from 9 V to 30 V with Built−in Overvoltage
•
•
•
•
•
•
•
Protection
High−Voltage Startup Circuit and Active Input Filter Capacitor
Discharge Circuitry for Reduced Standby Power
Integrated High−Voltage Brown−Out Detector
Integrated High−Voltage Switch Disconnects PFC Feedback Resistor
Divider to Reduce Standby Power
Fault Input for Severe Fault Conditions, NTC Compatible (Latch and
Auto−Recovery Options)
0.5 A / 0.8 A Source / Sink Gate Drivers
Internal Temperature Shutdown
Power Savings Mode Reduces Supply Current Consumption to
70 mA Enabling Very Low Input Power Applications
MARKING DIAGRAM
HV/X2
BO/X2
PControl
SOIC−20 PONOFF
Narrow Body
QCT
CASE 751BS
Fault
PSTimer
QFB
1
20
NCP1937xxG
AWLYWW
Combination Power Factor
Correction and QuasiResonant Flyback
Controllers for Adapters
PFBHV
PFBLV
GND
PCS/PZCD
PDRV
QDRV
QCS
VCC
QZCD
NCP1937 = Specific Device Code
xx
= A1, A2, A3, B1, B2, B3, B51,
= C1, C4 or C61
A
= Assembly Location
WL
= Wafer Lot
YY
= Year
WW
= Work Week
G
= Pb−Free Package
ORDERING INFORMATION
See detailed ordering and shipping information on page 5 of
this data sheet.
QR Flyback Controller Features
• Valley Switching Operation with Valley−
•
•
•
•
•
•
Lockout for Noise−Free Operation
Frequency Foldback with Minimum Frequency Clamp for Highest Performance in
Standby Mode
Minimum Frequency Clamp Eliminates
Audible Noise
Timer−Based Overload Protection
(Latched or Auto−Recovery options)
Adjustable Overpower Protection
Winding and Output Diode Short−Circuit
Protection
4 ms Soft−Start Timer
PFC Controller Features
• Critical Conduction Mode with Constant On Time Control (Voltage
•
•
•
•
•
•
Mode) and Maximum Frequency Clamp
Accurate Overvoltage Protection
Bi−Level Line−Dependent Output Voltage
Fast Line / Load Transient Compensation
Boost Diode Short−Circuit Protection
Feed−Forward for Improved Operation across Line and Load
Adjustable PFC Disable Threshold Based on Output Power
© Semiconductor Components Industries, LLC, 2015
June, 2022 − Rev. 8
1
Publication Order Number:
NCP1937/D
N
L
PCS/PZCD
N L
PFBHV
PFBLV
BO/X2
GND
PControl PCS/PZCD
PONOFF
PDRV
QCT
QDRV
Fault
QCS
PSTimer
VCC
QZCD
QFB
NCP1937
HV/X2
PDRV
VZCD
PCS/PZCD
PDRV
QCS
VCC
VPSTimer
VCC
QCS
(Aux)
VZCD
PSM Control
NCP1937
Figure 1. Typical Application Circuit
www.onsemi.com
2
NCP1937
Enable
PFC
PUVP
PFC
OVP
Detection
KPOVP(xL)
DPOVP(xL)
IPControl(boost)
KLOW(HYS)
KLOW
Low/High Line
IEA
VPREF(xL)
ON Time
Ramp
+
−
In Regulation
5
PFCDRV
Disable PFC
PUVP
Low
Clamp
tPisable
PSKIP
PDRV
PFCDRV
15
tPFC(off)
Timer
PFCDRV
Frequency
Clamp
Minimum
Frequency
Oscillator
8
IPCS/PZCD
+
−
IPSTimer1/2
PSTIMER
9
In_PSM
11
VQZCD(hys)
QSkip
VQZCD(th)
CT
VQFB
VCO Setpoint
QRDRV
IQCT
QCT
7
nPILIM2
S
S
S
S
S
S
S
Line Removal
Brownout
VCC(reset)
R
R
R
OTP
PSM
Detection
Initial Discharge
Valley
QSkip
VCO
tonQR(MAX)
TSD
QOVLD
nQILIM2
nPILIM2
OVP
OTP
VCCOVP
OVP
RQFB
QSkip
PILIM2
Counter
VFault(OVP)
VFault(OTP_in)
QZCD
ZCD
Detect
VCO
QRDRV
VPILIM1
−
+
Fault
QDRV
14
Soft−start
VQZCD
IQFB
+
− VPILIM2
IOTP
PONOFF
6
+
V
− POFF
tQ(toutx)
Valley
PILIM1
+
−
CCC
tP(tout)
16
LEB2
PFCDRV
PZCD
+
V
− PZCD
PZCD
LEB1
VCC
12
QRDRV
R
Q
Dominant
Reset
S Latch Q
tdelay(QSKIP)
PILIM2
PFCDRV
ZCD
Detect
PCS/PZCD
R
Q
Dominant
Reset Latch
Q
S
POVP
PUVP
PSKIP
PILIM1
PILIM2
HV/X2
1
IPONOFF
VPONHYS
Disable PFC
+
DV
− PSKIP
In Regulation
BO/X2
3
VQFB
QZCD
Soft−Start
Level
Shift
VPCONTROL(MAX)
PControl
QR_EN
Valley
Select
Logic
VQFB
+
QRDRV
QFB
In_PSM
10
/KQFB
VQZCD
Latch
Fault
Logic
QILIM1
LEB1
Auto−recovery
QOVLD
tQOVLD
+
VQZCD
Temperature
TSD
In PSM
QILIM2
nQILIM2
VCC_OK
Figure 2. Functional Block Diagram
www.onsemi.com
3
GND
17
Counter
VQILIM1
LEB2
IQCS
QCS
13
−
+
PFBLV
18
Latch
Auto−recovery
VCC(reset)
Line Removal
Brownout
POVP
Low/High Line
VCC_OK
In PSM
VPFB(HYS)
V
− PFB(disable)
+
Low/High Line
High Voltage
Brownout
Startups,
Line Removal
Detection, and
VCC_OK
Logic
QR_EN
Central
In PSM
Logic
Reset
Istart
VCCOVP
VCC
VCC_OK
Management
VCC(reset)
VDD
Line Removal
ICC(discharge)
Soft−start
VQILIM1
−
+
20
+
−
PFBHV
VQILIM2
NCP1937
Table 1. PIN FUNCTION DESCRIPTION
Pin Out
Name
1
HV/X2
2
3
Function
High voltage startup circuit input. It is also used to discharge the input filter capacitors.
Removed for creepage distance.
BO/X2
4
Performs brown−out detection for the whole IC and it is also used to discharge the input filter capacitors
and detect the line voltage range.
Removed for creepage distance.
5
PControl
Output of the PFC transconductance error amplifier. A compensation network is connected between this
pin and ground to set the loop bandwidth.
6
PONOFF
A resistor between this pin and ground sets the PFC turn off threshold. The voltage on this pin is compared to an internal voltage signal proportional to the output power. The PFC disable threshold is determined by the resistor on this pin and the internal pull–up current source, IPONOFF.
7
QCT
An external capacitor sets the frequency in VCO mode for the QR flyback controller.
8
Fault
The controller enters fault mode if the voltage of this pin is pulled above or below the fault thresholds. A
precise pull up current source allows direct interface with an NTC thermistor. Fault detection triggers a
latch or auto−recovery depending on device option.
9
PSTimer
10
QFB
11
QZCD
12
VCC
Supply input.
13
QCS
Input to the cycle−by−cycle current limit comparator for the QR Flyback section.
14
QDRV
QR flyback controller switch driver.
15
PDRV
PFC controller switch driver.
16
PCS/PZCD
17
GND
18
PFBLV
19
20
Power savings mode (PSM) control and timer adjust. Compatible with an optocoupler for secondary control of PSM. The device enters PSM if the voltage on this pin exceeds the PSM threshold, VPS_in. A capacitor between this pin and GND sets the delay time before the controller enters power savings mode.
Once the controller enters power savings mode the IC is disabled and the current consumption is reduced to a maximum of 70 mA. The input filter capacitor discharge function is available while in power
savings mode. The controller is enabled once VPSTimer drops below VPS_out.
Feedback input for the QR Flyback controller. Allows direct connection to an optocoupler.
Input to the demagnetization detection comparator for the QR Flyback controller. Also used to set the
overpower compensation.
Input to the cycle−by−cycle current limit comparator for the PFC section. Also used to perform the demagnetization detection for the PFC controller.
Ground reference.
Low voltage PFC feedback input. An external resistor divider is used to sense the PFC bulk voltage. The
divider low side resistor connects to this pin. This voltage is compared to an internal reference. The reference voltage is 2.5 V at low line and 4 V at high line. An internal high−voltage switch disconnects the low
side resistor from the high side resistor chain when the PFC is disabled in order to reduce input power.
Removed for creepage distance.
PFBHV
High voltage PFC feedback input. An external resistor divider is used to sense the PFC bulk voltage. The
divider high side resistor chain from the PFC bulk voltage connects to this pin. An internal high−voltage
switch disconnects the high side resistor chain from the low side resistor when the PFC is disabled in
order to reduce input power.
www.onsemi.com
4
NCP1937
Table 2. NCP1937 DEVICE OPTIONS
Fault OTP
VBO(start)
Typ
VBO(stop)
Typ
PFC
Disable
Time
PFC
Frequency
Clamp
Auto−Recovery
Latch
111 V
101 V
0.5 s
250 kHz
NCP1937A2DR2G
Auto−Recovery
Latch
111 V
101 V
0.5 s
131 kHz
NCP1937A3DR2G
Auto−Recovery
Latch
111 V
101 V
4s
131 kHz
NCP1937B1DR2G
Auto−Recovery
Auto−Recovery
111 V
101 V
0.5 s
250 kHz
NCP1937B2DR2G
Auto−Recovery
Auto−Recovery
111 V
101 V
0.5 s
131 kHz
NCP1937B3DR2G
Auto−Recovery
Auto−Recovery
111 V
101 V
4s
131 kHz
NCP1937B51DR2G
Auto−Recovery
Auto−Recovery
97 V
87 V
0.5 s
131 kHz
NCP1937C1DR2G
Latch
Latch
111 V
101 V
0.5 s
250 kHz
NCP1937C4DR2G
Latch
Latch
111 V
101 V
13 s
131 kHz
NCP1937C61DR2G
Latch
Latch
97 V
87 V
4s
131 kHz
Device
Overload
Protection
NCP1937A1DR2G
Package
Shipping†
SOIC−20
(Pb−Free)
2500 / Tape
& Reel
†For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging
Specification Brochure, BRD8011/D.
Table 3. MAXIMUM RATINGS (Notes 1 − 6)
Rating
Pin
Symbol
Value
Unit
High Voltage Startup Circuit Input Voltage
1
VHV/X2
−0.3 to 700
V
High Voltage Startup Circuit Input Current
1
IHV/X2
20
mA
High Voltage Brownout Detector Input Voltage
3
VBO/X2
−0.3 to 700
V
High Voltage Brownout Detector Input Current
3
IBO/X2
20
mA
PFC High Voltage Feedback Input Voltage
20
VPFBHV
−0.3 to 700
V
PFC High Voltage Feedback Input Current
20
IPFBHV
0.5
mA
PFC Low Voltage Feedback Input Voltage
18
VPFBLV
−0.3 to 9
V
PFC Low Voltage Feedback Input Current
18
IPFBLV
0.5
mA
PFC Zero Current Detection and Current Sense Input Voltage (Note 1)
16
VPCS/PZCD
−0.3 to VPCS/PZCD(MAX)
V
PFC Zero Current Detection and Current Sense Input Current
16
IPCS/PZCD
−2/+5
mA
PFC Control Input Voltage
5
VPControl
−0.3 to 5
V
PFC Control Input Current
5
IPControl
10
mA
Supply Input Voltage
12
VCC(MAX)
−0.3 to 30
V
Supply Input Current
12
ICC(MAX)
30
mA
Supply Input Voltage Slew Rate
12
dVCC/dt
1
V/ms
Fault Input Voltage
8
VFault
−0.3 to (VCC + 1.25)
V
Fault Input Current
8
IFault
10
mA
QR Flyback Zero Current Detection Input Voltage
11
VQZCD
−0.9 to (VCC + 1.25)
V
QR Flyback Zero Current Detection Input Current
11
IQZCD
−2/+5
mA
Stresses exceeding those listed in the Maximum Ratings table may damage the device. If any of these limits are exceeded, device functionality
should not be assumed, damage may occur and reliability may be affected.
1. VPCS/PZCD(MAX) is the maximum voltage of the pin shown in the electrical table. When the voltage on this pin exceeds 5 V, the pin sinks
a current equal to (VPCS/PZCD − 5 V) / (2 kW). A VPSC/PZCD of 7 V generates a sink current of approximately 1 mA.
2. Maximum driver voltage is limited by the driver clamp voltage, VXDRV(high), when VCC exceeds the driver clamp voltage. Otherwise, the
maximum driver voltage is VCC.
3. Maximum Ratings are those values beyond which damage to the device may occur. Exposure to these conditions or conditions beyond
those indicated may adversely affect device reliability. Functional operation under absolute maximum–rated conditions is not implied.
Functional operation should be restricted to the Recommended Operating Conditions.
4. This device contains Latch−Up protection and exceeds ± 100 mA per JEDEC Standard JESD78.
5. Low Conductivity Board. As mounted on 80 x 100 x 1.5 mm FR4 substrate with a single layer of 50 mm2 of 2 oz copper traces and heat
spreading area. As specified for a JEDEC51−1 conductivity test PCB. Test conditions were under natural convection of zero air flow.
6. Pins 1, 3, and 20 are rated to the maximum voltage of the part, or 700 V.
www.onsemi.com
5
NCP1937
Table 3. MAXIMUM RATINGS (Notes 1 − 6)
Rating
Pin
Symbol
Value
Unit
QR Feedback Input Voltage
7
VQCT
−0.3 to 10
V
QR Feedback Input Current
7
IQCT
10
mA
QR Flyback Current Sense Input Voltage
13
VQCS
−0.3 to 10
V
QR Flyback Current Sense Input Current
13
IQCS
10
mA
QR Flyback Feedback Input Voltage
10
VQFB
−0.3 to 10
V
QR Flyback Feedback Input Current
10
IQFB
10
mA
PSTimer Input Voltage
9
VPSTimer
−0.3 to 10
V
PSTimer Input Current
9
IPSTimer
10
mA
PFC Driver Maximum Voltage (Note 2)
15
VPDRV
−0.3 to VPDRV(high)
V
PFC Driver Maximum Current
15
IPDRV(SRC)
IPDRV(SNK)
500
800
mA
Flyback Driver Maximum Voltage (Note 2)
14
VQDRV
−0.3 to VQDRV(high)
V
Flyback Driver Maximum Current
14
IQDRV(SRC)
IQDRV(SNK)
500
800
mA
PFC ON/OFF Threshold Adjust Input Voltage
6
VPONOFF
−0.3 to 10
V
PFC ON/OFF Threshold Adjust Input Current
6
IPONOFF
10
mA
Operating Junction Temperature Range
N/A
TJ
−40 to 125
_C
Maximum Junction Temperature
N/A
TJ(MAX)
150
_C
Storage Temperature Range
N/A
TSTG
–60 to 150
_C
Power Dissipation (TA = 75_C, 1 Oz Cu, 0.155 Sq Inch Printed Circuit
Copper Clad)
Plastic Package SOIC−20NB
PD
W
0.62
Thermal Resistance, Junction−to−Ambient
(1 oz. Cu Printed Circuit Copper Clad)
Plastic Package SOIC−20NB
RθJA
Thermal Resistance, Junction−to−Case
RθJC
77
ESD Capability (Note 6)
Human Body Model per JEDEC Standard JESD22−A114F.
Machine Model per JEDEC Standard JESD22−A115−A.
Charge Device Model per JEDEC Standard JESD22−C101E.
HBM
MM
CDM
3000
200
750
_C/W
121
_C/W
V
Stresses exceeding those listed in the Maximum Ratings table may damage the device. If any of these limits are exceeded, device functionality
should not be assumed, damage may occur and reliability may be affected.
1. VPCS/PZCD(MAX) is the maximum voltage of the pin shown in the electrical table. When the voltage on this pin exceeds 5 V, the pin sinks
a current equal to (VPCS/PZCD − 5 V) / (2 kW). A VPSC/PZCD of 7 V generates a sink current of approximately 1 mA.
2. Maximum driver voltage is limited by the driver clamp voltage, VXDRV(high), when VCC exceeds the driver clamp voltage. Otherwise, the
maximum driver voltage is VCC.
3. Maximum Ratings are those values beyond which damage to the device may occur. Exposure to these conditions or conditions beyond
those indicated may adversely affect device reliability. Functional operation under absolute maximum–rated conditions is not implied.
Functional operation should be restricted to the Recommended Operating Conditions.
4. This device contains Latch−Up protection and exceeds ± 100 mA per JEDEC Standard JESD78.
5. Low Conductivity Board. As mounted on 80 x 100 x 1.5 mm FR4 substrate with a single layer of 50 mm2 of 2 oz copper traces and heat
spreading area. As specified for a JEDEC51−1 conductivity test PCB. Test conditions were under natural convection of zero air flow.
6. Pins 1, 3, and 20 are rated to the maximum voltage of the part, or 700 V.
www.onsemi.com
6
NCP1937
Table 4. ELECTRICAL CHARACTERISTICS: (VCC = 12 V, VBO/X2 = 120 V, VHV/X2 = 120 V, VFault = open, VRPFBHV = 20 V,
VPFBLV = 2.4 V, VPControl = 4 V, VPCS/PZCD = 0 V, VQFB = 3 V, VPONOFF = 4 V, VQCS = 0 V, VQZCD = 0 V, VPSTimer = 0 V, RPFBHV = 200 kW,
CVCC = 100 nF , CQCT = 220 pF, CPDRV = 1 nF, CQDRV = 1 nF, for typical values TJ = 25_C, for min/max values, TJ is – 40_C to 125_C,
unless otherwise noted)
Characteristics
Conditions
Pin
Symbol
Min
Typ
Max
VCC increasing
VQFB = 0, VPSTimer = 3 V
VCC decreasing
VCC(on) − VCC(off)
VCC(PS_on) − VCC(off)
VCC decreasing
VCC increasing,
IHV/X2 = 650 mA
12
VCC(on)
VCC(PS_on)
VCC(off)
VCC(HYS)
VCC(DPS_off)
VCC(reset)
VCC(inhibit)
16
–
8.2
7.7
1.65
4.5
0.3
17
11
8.8
–
2.20
5.5
0.7
18
–
9.4
–
2.75
7.5
0.95
VCC = 0 V, VBO/X2 = 0 V
VCC = 0 V, VHV/X2 = 0 V
12
12
Istart1A
Istart1B
0.20
0.20
0.50
0.50
0.65
0.65
12
Istart2A
2.5
5
Istart2B
2.5
5
Istart2A_PSM
9
15
20
Istart2B_PSM
9
15
20
Unit
STARTUP AND SUPPLY CIRCUITS
V
Supply Voltage
Startup Threshold
Regulation Level in PSM
Minimum Operating Voltage
Operating Hysteresis
Delta Between PSM and VCC(off) Levels
Internal Latch / Logic Reset Level
Transition from Istart1 to Istart2
Startup Current in Inhibit Mode
Startup Current
Operating Mode
PSM Mode
Startup Circuit Off−State Leakage Current
Minimum Startup Voltage
Minimum Startup Voltage in PSM
VCC = VCC(on) – 0.5 V
VHV/X2 = 100 V,
VBO/X2 = VCC
VBO/X2 = 100 V,
VHV/X2 = VCC
VHV/X2 = 100 V,
VBO/X2 = 0 V
VBO/X2 = 100 V,
VHV/X2 = 0 V
mA
12
V HV/X2 = 500 V
1
IHV/X2 (off)
–
–
3
mA
Istart2A = 1 mA, VCC =
VCC(on) – 0.5 V
Istart2B = 1 mA, VCC =
VCC(on) – 0.5 V
1
VHV/X2(MIN)
–
–
40
V
3
VBO/X2(MIN)
–
–
40
Istart = 9 mA, VCC =
VCC(PS_on) – 0.5 V
Istart = 9 mA, VCC =
VCC(PS_on) – 0.5 V
1
VHV/X2(MIN)
–
–
60
3
VBO/X2(MIN)
–
–
60
27
28
29
VCC Overvoltage Protection Threshold
12
VCC(OVP)
VCC Overvoltage Protection Delay
12
tdelay(VCC_OVP)
Supply Current
12
In Power Savings Mode
Before Startup, Fault or Latch
Flyback in Skip, PFC Disabled
Flyback in Skip, PFC in Skip
Flyback Enabled, QDRV Low, PFC Disabled
Flyback Enabled, QDRV Low, PFC in Skip
PFC and Flyback switching at 70 kHz
PFC and Flyback switching at 70 kHz
mA
V
ms
mA
ICC1a
ICC2
ICC3a
ICC3b
VCC = VCC(on) – 0.5 V
VQFB = 0.35 V
VQFB = 0.35 V,
VPControl < VPSKIP
VQZCD = 1 V,
VQZCD = 1 V,
VPControl < VPSKIP
CQDRV = CPDRV = open
30.0
V
–
–
0.15
0.3
0.5
0.07
0.25
0.4
1.0
ICC4
ICC5
0.85
1.1
1.35
1.8
ICC6
ICC7
1.5
2.8
4.0
5.2
INPUT FILTER DISCHARGE
VCC = VCC(off) + 200 mV
12
ICC(discharge)
8.0
11.5
15.0
mA
Line Voltage Removal Detection Threshold
VBO/X2 decreasing
3
Vlineremoval
20
30
40
V
Line Voltage Removal Detection Delay
VBO/X2 stays above
Vlineremoval
3
tlineremoval
130
200
270
ms
Current Consumption in Discharge Mode
www.onsemi.com
7
NCP1937
Table 4. ELECTRICAL CHARACTERISTICS: (VCC = 12 V, VBO/X2 = 120 V, VHV/X2 = 120 V, VFault = open, VRPFBHV = 20 V,
VPFBLV = 2.4 V, VPControl = 4 V, VPCS/PZCD = 0 V, VQFB = 3 V, VPONOFF = 4 V, VQCS = 0 V, VQZCD = 0 V, VPSTimer = 0 V, RPFBHV = 200 kW,
CVCC = 100 nF , CQCT = 220 pF, CPDRV = 1 nF, CQDRV = 1 nF, for typical values TJ = 25_C, for min/max values, TJ is – 40_C to 125_C,
unless otherwise noted)
Characteristics
Conditions
Pin
Symbol
Min
Typ
Max
Unit
System Brown−out Thresholds
(See Table 2 for device options)
VBO/X2 increasing
VBO/X2 decreasing
3
VBO(start)
VBO(stop)
102
86
111
101
120
116
V
System Brown−out Thresholds
(See Table 2 for device options) (B51, C61)
VBO/X2 increasing
VBO/X2 decreasing
3
VBO(start)
VBO(stop)
83
79
97
87
111
95
V
Brown−out Hysteresis
VBO/X2 increasing
3
VBO(hys)
4
16
V
Brown−out Detection Blanking Time
VBO/X2 decreasing,
duration below VBO(stop)
for a Brown−out fault
3
tBO(stop)
43
54
65
ms
Brown−out Drive Disable Threshold
VBO/X2 decreasing,
threshold to disable
switching
3
VBO(DRV_disable)
20
30
40
V
Line Level Detection Threshold
Line Level Detection Threshold (B51, C61)
VBO/X2 increasing
3
VBO(lineselect)
216
199
240
221
264
243
V
High to Low Line Mode Selector Timer
VBO/X2 decreasing
3
thigh to low line
43
54
65
ms
Low to High Line Mode Selector Timer
VBO/X2 increasing
3
tlow to high line
200
350
450
ms
V BO/X2 = 500 V
3
IBO/X2(off)
–
–
42
mA
15
tPFC(off1)
tPFC(off2)
100
700
200
1000
300
1300
ms
Cycle by Cycle Current Sense Threshold
16
VPILIM1
0.45
0.50
0.55
V
Cycle by Cycle Leading Edge
Blanking Duration
16
tPCS(LEB1)
250
325
400
ns
Cycle by Cycle Current Sense
Propagation Delay
16
tPCS(delay1)
100
200
ns
Abnormal Overcurrent Fault Threshold
16
VPILIM2
1.12
1.25
1.38
V
Abnormal Overcurrent Fault Leading Edge
Blanking Duration
16
tPCS(LEB2)
100
175
250
ns
Abnormal Overcurrent Fault Propagation Delay
16
tPCS(delay2)
100
200
ns
Number of Consecutive Abnormal Overcurrent
Faults to Enter Latch Mode
15
nPILIM2
–
4
–
VPCS/PZCD = 1.5 V
16
IPCS/PZCD
0.7
1.0
1.3
mA
VBO/X2 > VBO(lineselect)
VBO/X2 < VBO(lineselect)
18
VPREF(HL)
VPREF(LL)
3.92
2.45
4.00
2.50
4.08
2.55
V
5
IEA(SRCHL)
IEA(SNKHL)
IEA(SRCLL)
IEA(SNKLL)
16
16
10
10
32
32
20
20
48
48
30
30
VPFBLV = VPREF(LL) ± 4%
VPFBLV = VPREF(HL) ± 4%
5
gm
gm_HL
100
100
200
200
300
300
mS
VPFBLV * KLOW(PFCxL),
CPControl = 10 nF
5
VPControl(MAX)
–
4.5
–
V
VPFBLV * KPOVP(xL),
CPControl = 10 nF
5
VPControl(MIN)
–
0.5
–
V
BROWN−OUT DETECTION
Brownout Pin Off State Leakage Current
PFC MAXIMUM OFF TIME TIMER
Maximum Off Time
VPCS/PZCD > VPILIM2
PFC CURRENT SENSE
Pull−up Current Source
PFC REGULATION BLOCK
Reference Voltage
Error Amplifier Current
Source
Sink
Source
Sink
Open Loop Error Amplifier Transconductance
Maximum Control Voltage
Minimum Control Voltage (PWM Offset)
PFC Enabled
VPFBLV = 0.96 x VPREF(HL)
VPFBLV = 1.04 x VPREF(HL)
VPFBLV = 0.96 x VPREF(LL)
VPFBLV = 1.04 x VPREF(LL)
www.onsemi.com
8
mA
NCP1937
Table 4. ELECTRICAL CHARACTERISTICS: (VCC = 12 V, VBO/X2 = 120 V, VHV/X2 = 120 V, VFault = open, VRPFBHV = 20 V,
VPFBLV = 2.4 V, VPControl = 4 V, VPCS/PZCD = 0 V, VQFB = 3 V, VPONOFF = 4 V, VQCS = 0 V, VQZCD = 0 V, VPSTimer = 0 V, RPFBHV = 200 kW,
CVCC = 100 nF , CQCT = 220 pF, CPDRV = 1 nF, CQDRV = 1 nF, for typical values TJ = 25_C, for min/max values, TJ is – 40_C to 125_C,
unless otherwise noted)
Characteristics
Conditions
Pin
Symbol
Min
Typ
Max
Unit
EA Output Control Voltage Range
VPControl(MAX) VPControl(MIN)
5
DVPControl
3.8
4.0
4.2
V
Delta Between Minimum Control Voltage and
Lower Clamp PControl Voltages
VPControl(MIN) −
VPClamp(lower)
5
DVPClamp(lower)
−125
−100
−75
mV
Ratio between the Vout Low Detect Threshold
and the Regulation Level
VPFBLV decreasing,
VBOOST / VPREF(HL)
VPFBLV decreasing,
VBOOST / VPREF(LL)
18
KLOW(PFCHL)
0.940
0.945
0.950
KLOW(PFCLL)
0.940
0.945
0.950
VPFBLV increasing
18
KLOW(HYSHL)
KLOW(HYSLL)
0.950
0.950
0.960
0.960
0.965
0.965
5
IPControl(boost)
190
240
290
mA
VPControl increasing
5
IIn_Regulation
−6.5
–
0
mA
IPControl = 5 mA
5
RPControl
4
25
50
W
VPControl decreasing,
measured from
VPClamp(lower)
5
DVPSKIP
5
25
50
mV
PFC Skip Hysteresis
VPControl increasing
5
VPSKIP(HYS)
25
50
75
mV
Delay Exiting Skip Mode
Apply 1 V step from
VPClamp(lower)
5
tdelay(PSKIP)
–
50
60
ms
18
KPOVP(LL)
1.06
1.08
1.10
KPOVP(HL)
1.05
1.06
1.08
PFC REGULATION BLOCK
Ratio between the Vout Low Exit Threshold
and the Regulation Level
Source Current During Vout Low Detect
PFC In Regulation Threshold
Resistance of Internal Pull Down Switch
PFC SKIP MODE
Delta Between Skip Level and Lower Clamp
PControl Voltages
PFC FAULT PROTECTION
Ratio between the Hard Overvoltage Protection Threshold and Regulation Level
Soft Overvoltage Protection Threshold
VPFBLV increasing
KPOVP(LL) =
VPFBLV/VPREF(LL)
KPOVP(HL) =
VPFBLV/VPREF(HL)
VPSOVP(LL) = soft
overvoltage level
DPOVP(LL) = KPOVP *
VPREF(LL) − VPSOVP(LL)
DPOVP(HL) = KPOVP *
VPREF(HL) − VPSOVP(HL)
mV
18
DPOVP(LL)
20
–
55
DPOVP(HL)
20
–
55
PFC Feedback Pin Disable Threshold
VPFBLV decreasing
18
VPFB(disable)
0.225
0.30
0.35
V
PFC Feedback Pin Enable Threshold
VPFBLV increasing
18
VPFB(enable)
0.275
0.35
0.40
V
PFC Feedback Pin Hysteresis
VPFBLV increasing
18
VPFB(HYS)
25
50
mV
18
tdelay(PFB)
30
ms
PFC Feedback Disable Delay
PFC ON TIME CONTROL
PFC Maximum On Time
VPControl = VPControl(MAX),
VBO/X2 = 163 V
VBO/X2 = 325 V
15
Minimum On−Time
VPControl = VPControl(MIN)
PFC Frequency Clamp
(See Table 2 for device options)
www.onsemi.com
9
ms
ton1a
ton1b
12.5
4.25
15
5.00
17.5
5.75
15
tP(on−time)
–
–
200
ns
15
fclamp(PFC)
112
215
131
250
150
285
kHz
NCP1937
Table 4. ELECTRICAL CHARACTERISTICS: (VCC = 12 V, VBO/X2 = 120 V, VHV/X2 = 120 V, VFault = open, VRPFBHV = 20 V,
VPFBLV = 2.4 V, VPControl = 4 V, VPCS/PZCD = 0 V, VQFB = 3 V, VPONOFF = 4 V, VQCS = 0 V, VQZCD = 0 V, VPSTimer = 0 V, RPFBHV = 200 kW,
CVCC = 100 nF , CQCT = 220 pF, CPDRV = 1 nF, CQDRV = 1 nF, for typical values TJ = 25_C, for min/max values, TJ is – 40_C to 125_C,
unless otherwise noted)
Characteristics
Conditions
Pin
Symbol
Min
Typ
Max
Unit
Voltage to Current Conversion Ratio
VQFB = 3 V, Low Line
VQFB = 3 V, High Line
6
Iratio1(QFB/PON)
Iratio2(QFB/PON)
14
14
15
15
16
16
mA
PFC Disable Threshold
VPONOFF decreasing
6
VPOFF
1.9
2.0
2.1
V
PFC Enable Hysteresis
VPONOFF = increasing
6
VPONHYS
0.135
0.160
0.185
V
PFC DISABLE
PONOFF Operating Mode Voltage
tdemag/T = 70%,
RPONOFF = 191 kW,
CPONOFF = 1 nF
VQFB = 1.8 V (decreasing)
VQFB = 3 V (decreasing)
V
6
VPONOFF1
VPONOFF2
1.08
1.8
1.20
2.0
1.32
2.2
6
tPdisable
0.45
3.6
11.7
0.50
4
13
0.55
4.4
14.3
s
6
tPenable(filter)
50
100
150
ms
PONOFF Increasing
6
tPenable
200
–
500
ms
PFC Off−State Leakage Current
VPONOFF = 1 V,
VPFBHV = 500 V
20
IPFBHV(off)
–
0.1
3
mA
PFC Feedback Switch On Resistance
VPFBHV = 4.25 V,
IPFBHV = 100 mA
20
RPFBswitch(on)
–
–
10
kW
Rise Time (10−90%)
VPDRV from 10 to 90%
of VCC
15
tPDRV(rise)
–
40
80
ns
Fall Time (90−10%)
90 to 10% of VPDRV
15
tPDRV(fall)
–
20
40
ns
15
RPDRV(SRC)
RPDRV(SNK)
VPDRV = 2 V
VPDRV = 10 V
15
IPDRV(SRC)
IPDRV(SNK)
–
–
500
800
–
–
mA
VCC = VCC(off) + 0.2 V,
RPDRV = 10 kW
VCC = 26 V,
RPDRV = 10 kW
15
VPDRV(high)
8
–
–
V
10
12
14
VFault = 4 V
15
VPDRV(low)
–
–
0.25
V
Zero Current Detection Threshold
VPCS/PZCD rising
VPCS/PZCD falling
16
VPZCD(rising)
VPZCD(falling)
675
200
750
250
825
300
mV
Hysteresis on Voltage Threshold
VPZCD(rising) – VPZCD(falling)
16
VPZCD(HYS)
375
500
625
mV
16
tPZCD
50
100
170
ns
16
VPCS/PZCD(MAX)
VPCS/PZCD(MIN)
6.5
−0.9
7
−0.7
7.5
0
V
16
tSYNC
–
70
200
ns
16
tP(tout)
8
10
12
ms
PFC Disable Timer
(See Table 2 for device options)
Disable Timer
PFC Enable Filter Delay
PFC Enable Timer
PFC GATE DRIVE
Driver Resistance
Source
Sink
Current Capability
Source
Sink
High State Voltage
Low Stage Voltage
13
7
W
PFC ZERO CURRENT DETECTION
Propagation Delay
Input Voltage Excursion
Upper Clamp
Negative Clamp
IPCS/PZCD = 1 mA
IPCS/PZCD = −2 mA
Minimum detectable ZCD Pulse Width
Missing Valley Timeout Timer
Measured after last ZCD
transition
www.onsemi.com
10
NCP1937
Table 4. ELECTRICAL CHARACTERISTICS: (VCC = 12 V, VBO/X2 = 120 V, VHV/X2 = 120 V, VFault = open, VRPFBHV = 20 V,
VPFBLV = 2.4 V, VPControl = 4 V, VPCS/PZCD = 0 V, VQFB = 3 V, VPONOFF = 4 V, VQCS = 0 V, VQZCD = 0 V, VPSTimer = 0 V, RPFBHV = 200 kW,
CVCC = 100 nF , CQCT = 220 pF, CPDRV = 1 nF, CQDRV = 1 nF, for typical values TJ = 25_C, for min/max values, TJ is – 40_C to 125_C,
unless otherwise noted)
Characteristics
Conditions
Pin
Symbol
Min
Typ
Max
Unit
VQDRV from 10 to 90%
14
tQDRV(rise)
–
40
80
ns
90 to 10% of VQDRV
14
tQDRV(fall)
–
20
40
ns
14
RQDRV(SRC)
RQDRV(SNK)
VQDRV = 2 V
VQDRV = 10 V
14
IQDRV(SRC)
IQDRV(SNK)
–
–
500
800
–
–
mA
VCC = VCC(off) + 0.2 V,
RQDRV = 10 kW
VCC = 26 V,
RQDRV = 10 kW
14
VQDRV(high)
8
–
–
V
10
12
14
VFault = 4 V
14
VQDRV(low)
–
–
0.25
V
Internal Pull−Up Current Source
10
IQFB
48
50
52
mA
Feedback Input Open Voltage
10
VQFB(open)
4.8
5.0
5.2
V
VQFB to Internal Current Setpoint Division Ratio
10
KQFB
3.95
4.0
4.15
–
10
RQFB
365
400
435
kW
VH2D
VH3D
VH4D
VHVCOD
VHVCOI
VH4I
VH3I
VH2I
1.316
1.128
0.846
0.752
1.316
1.504
1.692
1.880
1.400
1.200
0.900
0.800
1.400
1.600
1.800
2.000
1.484
1.272
0.954
0.848
1.484
1.696
1.908
2.120
QR FLYBACK GATE DRIVE
Rise Time (10−90%)
Fall Time (90−10%)
Driver Resistance
Source
Sink
Current Capability
Source
Sink
High State Voltage
Low Stage Voltage
13
7
W
QR FLYBACK FEEDBACK
QFB Pull Up Resistor
VPSTimer = 3 V;
VQFB = 0.4 V
Valley Thresholds
Transition from 1st to 2nd valley
Transition from 2nd to 3rd valley
Transition from 3rd to 4th valley
Transition from 4th valley to VCO
Transition from VCO to 4th valley
Transition from 4th to 3rd valley
Transition from 3rd to 2nd valley
Transition from 2nd to 1st valley
VQFB decreasing
VQFB decreasing
VQFB decreasing
VQFB decreasing
VQFB increasing
VQFB increasing
VQFB increasing
VQFB increasing
Skip Threshold
VQFB decreasing
10
VQSKIP
0.35
0.40
0.45
V
VQFB increasing
10
VQSKIP(HYS)
25
50
75
mV
Apply 1 V step from VQSKIP
10
tdelay(QSKIP)
–
–
10
ms
14
tonQR(MAX)
26
32
38
ms
VQFB = 0.5 V
7
VQCT(peak)
3.815
4.000
4.185
V
VQCT = 0 V
7
IQCT
18
20
22
mA
7
VQCT(min)
–
–
90
mV
7
fVCO(MIN)
23.5
27
30.5
kHz
Skip Hysteresis
Delay Exiting Skip Mode to
1st
QDRV Pulse
10
Maximum On Time
V
QR FLYBACK TIMING CAPACITOR
QCT Operating Voltage Range
On Time Control Source Current
Minimum voltage on QCT Input
Minimum Operating Frequency in VCO Mode
VQCT = VQCT(peak) +
100 mV
www.onsemi.com
11
NCP1937
Table 4. ELECTRICAL CHARACTERISTICS: (VCC = 12 V, VBO/X2 = 120 V, VHV/X2 = 120 V, VFault = open, VRPFBHV = 20 V,
VPFBLV = 2.4 V, VPControl = 4 V, VPCS/PZCD = 0 V, VQFB = 3 V, VPONOFF = 4 V, VQCS = 0 V, VQZCD = 0 V, VPSTimer = 0 V, RPFBHV = 200 kW,
CVCC = 100 nF , CQCT = 220 pF, CPDRV = 1 nF, CQDRV = 1 nF, for typical values TJ = 25_C, for min/max values, TJ is – 40_C to 125_C,
unless otherwise noted)
Characteristics
Conditions
Pin
Symbol
Min
Typ
Max
Unit
QZCD threshold voltage
VQZCD decreasing
11
VQZCD(th)
35
55
90
mV
QZCD hysteresis
VQZCD increasing
11
VQZCD(HYS)
15
35
55
mV
Demagnetization Propagation Delay
VQZCD step from
4.0 V to −0.3 V
11
tDEM
–
150
250
ns
Input Voltage Excursion
IQZCD = 5.0 mA
IQZCD = −2.0 mA
11
VQZCD(MAX)
VQZCD(MIN)
12.4
−0.9
12.7
−0.7
13.25
0
V
11
tZCD(blank)
2
3
4
ms
During soft−start
After soft−start
14
tQ(tout1)
tQ(tout2)
80
5.1
100
6
120
6.9
ms
VQCS increasing
VQCS increasing,
VQZCD = 1 V
13
VQILIM1a
VQILIM1b
0.760
0.760
0.800
0.800
0.840
0.840
V
Cycle by Cycle Leading Edge Blanking
Duration
13
tQCS(LEB1)
220
275
350
ns
Cycle by Cycle Current Sense Propagation
Delay
13
tQCS(delay1)
–
125
175
ns
13
VQILIM2
1.125
1.200
1.275
V
Abnormal Overcurrent Fault Leading Edge
Blanking Duration
13
tQCS(LEB2)
90
120
150
ns
Abnormal Overcurrent Fault Propagation Delay
13
tQCS(delay2)
–
125
175
ns
Number of Consecutive Abnormal Overcurrent
Faults to Enter Latch Mode
13
nQILIM2
–
4
–
QR FLYBACK DEMAGNETIZATION INPUT
Upper Clamp
Negative Clamp
Blanking Delay After Turn−Off
Timeout After Last Demagnetization Detection
QR FLYBACK CURRENT SENSE
Current Sense Voltage Threshold
Immediate Fault Protection Threshold
VQCS increasing,
VQFB = 4 V
Minimum Peak Current Level in VCO Mode
VQFB = 0.4 V,
VQCS increasing
13
Ipeak(VCO)
11
12.5
14
%
Set point decrease for VQZCD = − 250 mV
VQCS Increasing,
VQFB = 4 V
13
VOPP(MAX)
28
31.25
33
%
11
tQOPP(delay)
–
125
175
ns
13
IQCS
0.7
1.0
1.3
mA
13
tSSTART
2.8
4.0
5.0
ms
13
tQOVLD
60
80
100
ms
Overpower Protection Delay
Pull−up Current Source
VQCS = 1.5 V
QR FLYBACK FAULT PROTECTION
Soft−Start Period
Flyback Overload Fault Timer
VQCS = VQILIM1
www.onsemi.com
12
NCP1937
Table 4. ELECTRICAL CHARACTERISTICS: (VCC = 12 V, VBO/X2 = 120 V, VHV/X2 = 120 V, VFault = open, VRPFBHV = 20 V,
VPFBLV = 2.4 V, VPControl = 4 V, VPCS/PZCD = 0 V, VQFB = 3 V, VPONOFF = 4 V, VQCS = 0 V, VQZCD = 0 V, VPSTimer = 0 V, RPFBHV = 200 kW,
CVCC = 100 nF , CQCT = 220 pF, CPDRV = 1 nF, CQDRV = 1 nF, for typical values TJ = 25_C, for min/max values, TJ is – 40_C to 125_C,
unless otherwise noted)
Characteristics
Conditions
Pin
Symbol
Min
Typ
Max
Unit
Overvoltage Protection (OVP) Threshold
VFault increasing
8
VFault(OVP)
2.79
3.00
3.21
V
Delay Before Fault Confirmation
Used for OVP Detection
Used for OTP Detection
VFault increasing
VFault decreasing
tdelay(Fault_OVP)
tdelay(Fault_OTP)
22.5
22.5
30.0
30.0
37.5
37.5
COMMON FAULT PROTECTION
8
ms
Overtemperature Protection (OTP) Threshold
(Note 7)
VFault decreasing
8
VFault(OTP_in)
0.38
0.40
0.42
V
Overtemperature Protection (OTP) Exiting
Threshold (Note 7)
VFault increasing,
Options B and D
8
VFault(OTP_out)
0.874
0.920
0.966
V
VFault =
VFault(OTP_in) + 0.2 V
TJ = 110_C
8
IFault(OTP)
IFault(OTP_110)
42.5
–
45.5
45.5
48.5
–
VFault = open
8
VFault(clamp)
1.5
1.75
2.0
V
RFault(clamp)
1.32
1.55
1.82
kW
OTP Pull−up Current Source (Note 7)
Fault Input Clamp Voltage
Fault Input Clamp Series Resistor
mA
POWER SAVINGS MODE
PSM Enable Threshold
VPSTimer increasing
9
VPS_in
3.325
3.500
3.675
V
PSM Disable Threshold
VPSTimer decreasing
9
VPS_out
0.45
0.50
0.55
V
VPSTimer = 0.9 V
VPSTimer = 3.4 V
9
IPSTimer1
IPSTimer2
9
800
10
1000
11
1200
mA
IPSTimer2 Enable Threshold
9
VPSTimer2
0.95
1.0
1.05
V
Filter Delay Before Entering PSM
9
tdelay(PS_in)
VHV_X2 increasing
VBO_X2 increasing
1
3
VHV_X2(PS)
VBO_X2(PS)
20
20
30
30
40
40
V
VPSTimer = VPSTimer(off)
+ 10 mV
9
IPSTimer(DIS)
200
–
–
mA
VPSTimer decreasing
9
VPSTimer(off)
50
100
150
mV
Thermal Shutdown
Temperature increasing
N/A
TSHDN
150
_C
Thermal Shutdown Hysteresis
Temperature decreasing
N/A
TSHDN(HYS)
40
_C
PSTimer Pull Up Current Sources
Startup Circuits Turn−on Thresholds in PSM
PSTimer Discharge Current
PSTimer Discharge Turn Off Threshold
40
ms
THERMAL PROTECTION
7. NTC with R110 = 8.8 kW (TTC03−474)
www.onsemi.com
13
NCP1937
DETAILED OPERATING DESCRIPTION
Introduction
A dedicated comparator monitors VCC when the QR stage
is enabled and latches off the controller if VCC exceeds
VCC(OVP), typically 28 V.
The controller is disabled once a fault is detected. The
controller will restart the next time VCC reaches VCC(on) and
all non−latching faults have been removed.
The supply capacitor provides power to the controller
during power up. The capacitor must be sized such that a
VCC voltage greater than VCC(off) is maintained while the
auxiliary supply voltage is building up. Otherwise, VCC will
collapse and the controller will turn off. The operating IC
bias current, ICC4, and gate charge load at the drive outputs
must be considered to correctly size CCC. The increase in
current consumption due to external gate charge is
calculated using Equation 1.
The NCP1937 is a combination critical mode (CrM)
power factor correction (PFC) and quasi−resonant (QR)
flyback controller optimized for off−line adapter
applications. This device includes all the features needed to
implement a highly efficient adapter with extremely low
input power in no−load conditions.
This device reduces standby input power by integrating an
active input filter capacitor discharge circuit and
disconnecting the PFC feedback resistor divider when the
PFC is disabled.
High Voltage Startup Circuit
The NCP1937 integrates two high voltage startup circuits
accessible by the HV_X2 and BO_X2 pins. The startup
circuits are also used for input filter capacitor discharge. The
BO_X2 input is also used for monitoring the ac line voltage
and detecting brown−out faults. The startup circuits are
rated at a maximum voltage of 700 V.
A startup regulator consists of a constant current source
that supplies current from the ac input terminals (Vin) to the
supply capacitor on the VCC pin (CCC). The startup circuit
currents (Istart2A/B) are typically 3.75 mA. Istart2A/B are
disabled if the VCC pin is below VCC(inhibit). In this
condition the startup current is reduced to Istart1A/B,
typically 0.5 mA. The internal high voltage startup circuits
eliminate the need for external startup components. In
addition, these regulators reduce no load power and increase
the system efficiency as they use negligible power in the
normal operation mode.
Once CCC is charged to the startup threshold, VCC(on),
typically 17 V, the startup regulators are disabled and the
controller is enabled. The startup regulators remain disabled
until VCC falls below the minimum operating voltage
threshold, VCC(off), typically 8.8 V. Once reached, the PFC
and flyback controllers are disabled reducing the bias
current consumption of the IC. Both startup circuits are then
enabled allowing VCC to charge back up.
In power savings mode VCC is regulated by enabling the
startup circuits once the supply voltage decays below
VCC(PS_on), typically 11 V. The startup circuit is disabled
once VCC exceeds VCC(PS_on). This provides enough
headroom from VCC(off) to maintain a supply voltage and
allow the controller to detect the line voltage removal in
order to discharge the input filter capacitor(s). In this mode,
the supply capacitor is charged by the startup circuit on the
HV_X2 and BO_X2 pins once the voltage on these pin
exceeds 30 V, typically. This reduces the average voltage
during which the startup circuit is enabled reducing power
consumption. Both startup circuits are enabled once the
controller exits power savings mode in order to quickly
charge VCC. A new startup sequence commences once VCC
reaches VCC(on).
I CC(gate charge) + f @ Q G
(eq. 1)
where f is the operating frequency and QG is the gate charge
of the external MOSFETs.
Line Voltage Sense
The BO/X2 pin provides access to the brown−out and line
voltage detectors. It also provides access to the input filter
capacitor discharge circuit. The brown−out detector detects
mains interruptions and the line voltage detector determines
the presence of either 110 V or 220 V ac mains. Depending
on the detected input voltage range device parameters are
internally adjusted to optimize the system performance.
This pin connects to either line or neutral to achieve
half−wave rectification as shown in Figure 3. A diode is used
to prevent the pin from going below ground. A resistor in
series with the BO/X2 pin can be used for protection, but a
low value (< 3 kW) resistor should be used to reduce the
voltage offset while sensing the line voltage.
Figure 3. Brown−out and Line Voltage Detectors
Configuration
The flyback stage is enabled once VBO_X2 is above the
brown−out threshold, VBO(start), and VCC reaches VCC(on).
The high voltage startups are immediately enabled when the
voltage on VBO_X2 crosses over the brown−out start
threshold, VBO(start), to ensure that device is enabled quickly
upon exiting a brown−out state. Figure 4 shows typical
power up waveforms.
www.onsemi.com
14
NCP1937
VBO_X2
VBO(start)
Startup turns on when
device exits a brown−out
VBO/X2(MIN)
time
V CC
VCC(on)
VCC(off)
Startup
current = Istart2
Startup
current = Istart1
VCC(inhibit)
time
QDRV
Figure 4. Startup Timing Diagram
time
Figure 5 shows typical operation for the line voltage
detector. The default power−up mode of the controller is low
line. The controller switches to “high line” mode if VBO_X2
exceeds the line select threshold for longer than the low to
high line timer, t(low to high line), typically 300 ms, as long as
it was not previously in high line mode. If the controller has
switched from “high line” to “low line” mode, the low to
high line timer, t(low to high line), is inhibited until VBO/X2 falls
below VBO(stop). This prevents the controller from toggling
back to “high line” until at least one VBO(stop) transition has
occurred. The timer and logic is included to prevent
unwanted noise from toggling the operating line level.
In “high line” mode the high to low line timer, t(high to low
line), (typically 54 ms) is enabled once VBO_X2 falls below
VBO(lineselect). It is reset once VBO_X2 exceeds
VBO(lineselect). The controller switches back to “low line”
mode if the high to low line timer expires.
A timer is enabled once VBO_X2 drops below its stop
threshold, VBO(stop). If the timer, tBO, expires the device will
begin monitoring the voltage on VBO_X2 and disable the
PFC and flyback stages when that voltage is below the
Brown−out Drive Disable threshold, VBO(DRV_disable),
typically 30 V. This ensures that device switching is stopped
in a low energy state which minimizes inductive voltage
kick from the EMI components and ac mains. The timer, tBO,
typically 54 ms, is set long enough to ignore a single cycle
drop−out.
Line Voltage Detector
The input voltage range is detected based on the peak
voltage measured at the BO_X2 pin. Discrete values are
selected for the PFC stage gain (feedforward) depending on
the input voltage range. The controller compares VBO_X2 to
an internal line select threshold, VBO(lineselect). Once
VBO_X2 exceeds VBO(lineselect), the PFC stage operates in
“high line” (Europe/Asia) or “220 Vac” mode. In high line
mode the maximum on time is reduced by a factor of 3,
resulting in a maximum output power independent of input
voltage.
www.onsemi.com
15
NCP1937
HL
transition
blanked by
t low to high line
VBO/X2
HL transition
blanked by
VBO(stop)
counter
VBO(lineselect)
VBO(stop)
High
Line
Entered
Line
Timer
Starts
Line
Timer
Reset
Line
Timer
Starts
t(low to high line)
Low Line
Select
Timer
Line
Timer
Expires
Low Line
Entered
t (high to low line)
time
Operating
Mode
Transition to
High Line
Allowed?
Low Line
High Line
Low Line
time
Yes
No
Yes
time
Figure 5. Line Detector Waveforms
Input Filter Capacitor Discharge
both startup circuits are enabled. The startup circuits will
then source current from the BO_X2 and HV_X2 inputs to
the VCC pin and discharge the input filter capacitors by
transferring its charge to the VCC capacitor(s). The input
filter capacitor(s) are typically discharged once the startup
circuit turns on the 1st time because the energy stored in the
input filter capacitor(s) is significantly lower than the energy
needed to charge the VCC capacitor from VCC(off) to
VCC(on). After the initial discharge the controller enters a
low current mode (ICC2) once VCC drops to VCC(off).
In the event that the input filter capacitor is not fully
discharged, a larger VCC capacitor should be used. But, this
is not a concern for most applications because the supply
capacitor value will be large enough to maintain VCC during
skip operation. Figure 6 shows typical behavior of the filter
capacitor discharge when the ac line is removed.
Safety agency standards require the input filter capacitors
to be discharged once the ac line voltage is removed. A
resistor network is the most common method to meet this
requirement. Unfortunately, the resistor network consumes
power across all operating modes and is a major contributor
to the total input power dissipation during light−load and
no−load conditions.
The NCP1937 eliminates the need for external discharge
resistors by integrating active input filter capacitor
discharge circuitry. A novel approach is used to reconfigure
the high voltage startup circuits to discharge the input filter
capacitors upon removal of the ac line voltage.
Once the controller detects the absence of the ac line
voltage, the controller is disabled and VCC is discharged by
a current source, ICC(discharge), typically 11.5 mA. This will
cause VCC to fall down to VCC(off). Upon reaching VCC(off),
www.onsemi.com
16
NCP1937
AC Mains Unplugged
HV Startups Turn On
VAC
Input Filter Cap
Discharged
0V
tlineremoval
VCC
ICC(discharge) Begins
VCC(off)
QDRV
Figure 6. Input Filter Capacitor Discharge Waveforms
The diode connecting the AC line to the BO_X2 pin should
be placed after the system fuse. A resistor in series with the
BO_X2 pin is recommended to limit the current during
transient events. A low value resistor (< 3 kW) should be used
to reduce the voltage drop when the startup circuit is enabled.
once VPSTimer falls below its minimum operating level,
VPSTimer(MIN) (maximum of 50 mV). The time to enter PSM
mode is calculated using Equations 2 through 4. The time to
exit PSM mode is calculated using Equation 5.
t PSM(in) + t PSM(in1) ) t PSM(in2)
ǒ
ǒ
Power Savings Mode
t PSM(in1) + −R PSMC PSM @ In 1−
The NCP1937 has a low current consumption mode
known as power savings mode (PSM). The supply current
consumption in this mode is below 70 mA. PSM operation
is controlled by an external control signal. This signal is
typically generated on the secondary side of the power
supply and fed via an optocoupler.
The NCP1937 is configured as active on logic, that is it
enters PSM in the absence of the control signal. The control
signal is applied to the PSTimer pin. The block diagram for
NCP1937 PSTimer pin is shown in Figure 7. Power savings
mode operating waveforms for the NCP1937 are shown in
Figure 8.
The NCP1937 controller starts once VCC reaches VCC(on)
and no faults are present. At this time the current source on
the PSTimer pin, IPSTimer1, is enabled. IPSTimer1 is typically
10 mA. The current source charges the capacitor connected
from this pin to ground. Once VPSTimer reaches VPSTimer2 a
2nd current source, IPSTimer2, is enabled to speed up the
charge of CPSM. VPSTimer2 and IPSTimer2 are typically 1 V
and 1 mA, respectively. The controller enters PSM if the
voltage on VPSTimer exceeds VPS_in, typically 3.5 V. An
external optocoupler or switch needs to pull down on this pin
before its voltage reaches VPS_in to prevent entering PSM.
Once the controller enters PSM, IPSTimer1/2 is disabled. A
resistor between this pin and ground discharges the PSTimer
capacitor. The controller exits PSM once VPSTimer drops
below VPS_out, typically 0.5 V. Once the QR stage is
enabled, the capacitor on the PSTimer pin is discharged with
an internal pull down transistor. The transistor is disabled
t PSM(in2) [ −R PSMC PSM @ In 1−
ǒ
t PSM(out) + −R PSMC PSM @ In
Ǔ
Ǔ
V PSTimer2
I PSTimer1 @ R PSM
V PS_in−V PSTimer2
I PSTimer2 @ R PSM
Ǔ
V PS_out
V PS_in
(eq. 2)
(eq. 3)
(eq. 4)
(eq. 5)
In PSM the startup circuits on the HV_X2 and BO_X2
pins work to maintain VCC above VCC(off). The input filter
capacitor discharge circuitry continues operation in PSM.
The supply voltage is maintained in PSM by enabling one of
the startup circuits once VCC falls below VCC(PS_on)
(typically 11 V) and either VHV_X2 exceeds VHV_X2(PS) or
VBO_X2 exceeds VBO_X2(PS) (typically 30 V). The startup
circuit is disabled once VCC exceeds VCC(PS_on). A voltage
offset is observed on VCC while the startup circuit is enabled
due to the capacitor ESR. This will cause the startup circuit
to turn off because VCC exceeds VCC(PS_on). Internal
circuitry prevents the startup circuit from turning on
multiple times during the same ac line half−cycle. The
complementary startup circuit will then turn on during the
next half−cycle. Eventually, VCC will be regulated several
millivolts below VCC(PS_on). The offset is dependent on the
capacitor ESR.
This architecture enables the startup circuit for the exact
amount of time needed to regulate VCC. This results in a
significant reduction in power dissipation because the
average input voltage is greatly reduced.
www.onsemi.com
17
NCP1937
Figure 7. NCP1937 Power Savings Mode Control Block Diagram
Figure 8. NCP1937 Power Savings Mode Operating Waveforms
www.onsemi.com
18
NCP1937
Fault Input
the upper threshold, the external pull−up current has to be
higher than the pull−down capability of the clamp (set by
RFault(clamp) at VFault(clamp)). The upper fault threshold is
intended to be used for an overvoltage fault using a Zener
diode and a resistor in series from the auxiliary winding
voltage, VAUX. The controller is latched once VFault exceeds
VFault(OVP).
The Fault input signal is filtered to prevent noise from
triggering the fault detectors. Upper and lower fault detector
blanking delays, tdelay(Fault_OVP) and tdelay(Fault_OTP) are
both typically 30 ms. A fault is detected if the fault condition
is asserted for a period longer than the blanking delay.
A bypass capacitor is usually connected between the Fault
and GND pins and it will take some time for VFault to reach
its steady state value once IFault(OTP) is enabled. Therefore,
a lower fault (i.e. overtemperature) is ignored during
soft−start. In Options B and D, IFault(OTP) remains enabled
while the lower fault is present independent of VCC in order
to provide temperature hysteresis. The controller can detect
an upper OVP fault once VCC exceeds VCC(reset). The OVP
fault detection remains active provided the device is not in
PSM.
Once the controller is latched, it is reset if a brown−out
condition is detected or if VCC is cycled down to its reset
level, VCC(reset). In the typical application these conditions
occur only if the ac voltage is removed from the system.
Prior to reaching VCC(reset), Vfault(clamp) is set at 0 V.
The NCP1937 includes a dedicated fault input accessible
via the Fault pin. The controller can be latched by pulling the
pin above the upper fault threshold, VFault(OVP), typically
3.0 V. The controller is disabled if the Fault pin voltage,
VFault, is pulled below the lower fault threshold,
VFault(OTP_in), typically 0.4 V. The lower threshold is
normally used for detecting an overtemperature fault. The
controller operates normally while the Fault pin voltage is
maintained within the upper and lower fault thresholds.
Figure 9 shows the architecture of the Fault input.
The lower fault threshold is intended to be used to detect
an overtemperature fault using an NTC thermistor. A pull up
current source IFault(OTP), (typically 45.5 mA) generates a
voltage drop across the thermistor. The resistance of the
NTC thermistor decreases at higher temperatures resulting
in a lower voltage across the thermistor. The controller
detects a fault once the thermistor voltage drops below
VFault(OTP_in). Options A and C latch−off the controller after
an overtemperature fault is detected. In Options B and D the
controller is re−enabled once the fault is removed such that
VFault increases above VFault(OTP_out) and VCC reaches
VCC(on). Figure 10 shows typical waveforms related to the
latch option whereas Figure 11 shows waveforms of the
auto−recovery option.
An active clamp prevents the Fault pin voltage from
reaching the upper latch threshold if the pin is open. To reach
VAUX
blanking
tdelay(Fault_OVP)
+
−
VDD
S
Q
Latch
VFault(OVP)
R
IFault(OTP)
Fault
−
+
blanking
NTC
Thermistor
RFault(clamp)
VFault(clamp)
Soft−start
end
Hysteresis
Control
Option
Figure 9. Fault Detection Schematic
www.onsemi.com
19
BO_OK
Line Removal
Reset
tdelay(Fault_OTP)
VFault(OTP)
Auto−restart
Control
Auto−restart
NCP1937
VCC
VCC(on)
VCC(off)
VFault
VFault(clamp)
VFault(OTP)
tdelay(Fault_OTP)
OTP
Fault
Flag
OTP Fault Detected
QDRV
Figure 10. Latch−off Function Timing Diagram
time
VCC
VCC(on)
VCC(off)
VFault
VFault(clamp)
VFault(OTP_OUT)
VFault(OTP_IN)
VFault(OTP)
OTP
Fault
Flag
tdelay(Fault_OTP)
OTP Fault Detected
OTP Fault is Cleared
tSSTART
(OTP Fault
Ignored)
QDRV
Figure 11. OTP Auto−recovery Timing Diagram
www.onsemi.com
20
time
NCP1937
QR Flyback Valley Lockout
patent pending valley lockout circuitry to eliminate valley
jumping. Once a valley is selected, the controller stays
locked in this valley until the output power changes
significantly. Like a traditional QR flyback controller, the
frequency increases when the load decreases. Once a higher
valley is selected the frequency decreases very rapidly. It
will continue to increase if the load is further reduced. This
technique extends QR operation over a wider output power
range while maintaining good efficiency and limiting the
maximum operating frequency. Figure 12 shows a
qualitative frequency vs output power relationship.
Figure 13 shows the internal arrangement of the valley
lockout circuitry. The decimal counter increases each time
a valley is detected. The operating valley (1st, 2nd, 3rd or 4th)
is determined by the QFB voltage. As VQFB decreases or
increases, the valley comparators toggle one after another to
select the proper valley. The activation of an “n” valley
comparator blanks the “n−1” or “n+1” valley comparator
output depending if VQFB decreases or increases,
respectively.
A valley is detected once VQZCD falls below the QR
flyback demagnetization threshold, VQZCD(th), typically 55
mV. The controller will switch once the valley is detected or
increment the valley counter depending on QFB voltage.
The NCP1937 integrates a quasi−resonant (QR) flyback
controller. The power switch turn−off of a QR converter is
determined by the peak current set by the feedback loop. The
switch turn−on is determined by the transformer
demagnetization. The demagnetization is detected by
monitoring the transformer auxiliary winding voltage.
Turning on the power switch once the transformer is
demagnetized or reset reduces switching losses. Once the
transformer is demagnetized, the drain voltage starts ringing
at a frequency determined by the transformer magnetizing
inductance and the drain lump capacitance eventually
settling at the input voltage. A QR controller takes
advantage of the drain voltage ringing and turns on the
power switch at the drain voltage minimum or “valley” to
reduce switching losses and electromagnetic interference
(EMI).
The operating frequency of a traditional QR flyback
controller is inversely proportional to the system load. That
is, a load reduction increases the operating frequency. This
tradionally requires a maximum frequency clamp to limit
the operating frequency. This causes the controller to
become unstable and jump (or hesitate) between two valleys
generating audible noise. The NCP1937 incorporates a
Figure 12. Valley Lockout Frequency vs. Output Power Relationship
www.onsemi.com
21
NCP1937
Figure 13. Valley Lockout Detection Circuitry Internal Schematic
VQFB falls below VHVCOD. In VCO mode the peak current
is set as shown in Figure 15. The operating frequency in
VCO mode is adjusted to deliver the required output power.
A hysteresis between valleys provides noise immunity
and helps stabilize the valley selection in case of small
perturbations on VQFB.
Figure 14 shows the operating valley versus VQFB. Once
a valley is asserted by the valley selection circuitry, the
controller is locked in this valley until VQFB decreases or
increases such that VQFB reaches the next valley threshold.
A decrease in output power causes the controller to switch
from “n” to “n+1” valley until reaching the 4th valley.
A further reduction of output power causes the controller
to enter the voltage control oscillator (VCO) mode once
Valley
VCO
4th
3rd
2nd
1st
V HVCOD
VH4D
V H3D
V H2D
V HVCOI
V QFB
V H4I
VH3I
V H2I
Figure 14. Selected Operating Valley vs. VQFB
www.onsemi.com
22
V QILIM1*KQFB
NCP1937
Peak current
Setpoint
Skip
VCO
Mode
QR Mode
VQILIM1
ÎÎÎÎÎÎ
ÎÎÎÎÎÎ
ÎÎÎÎÎÎ
ÎÎÎÎÎÎ
ÎÎÎÎÎÎ
ÎÎÎÎÎÎ
ÎÎÎÎÎÎ
ÎÎÎÎÎÎ
ÎÎÎÎÎÎ
ÎÎÎÎÎÎ
1st Valley
2nd
3rd
Fault
4th
Ipeak(VCO)*VQILIM1
VQFB(TH)
VQSKIP VHVCOD
VH4D
VH3D
VH2D
VQILIM1*KQFB
VQFB
Figure 15. Operating Valley vs. VQFB
Figure 16 through Figure 19 show drain voltage, VQFB
and VQCT simulation waveforms for a reduction in output
power. The transitions between 2nd to 3rd, 3rd to 4th and 4th
valley to VCO mode are observed without any instabilities
or valley jumping.
Plot1
vdrain in volts
700
500
300
1
100
Plot2
feedback in volts
−100
2.00
1.60
1.20
800m
400m
2
Plot3
vct in volts
7.00
3
5.00
3.00
1.00
−1.00
3.64m
4.91m
6.18m
time in seconds
7.45m
8.72m
Figure 16. Operating Mode Transitions between 2nd to 3rd, 3rd to 4th and 4th Valley to VCO Mode
www.onsemi.com
23
NCP1937
1 vdrain
2 feedback
3 vct
Plot1
vdrain in volts
700
500
1
300
100
Plot2
feedback in volts
−100
2.15
2.05
2
1.95
1.85
1.75
Plot3
vct in volts
6.00
4.00
2.00
3
0
−2.00
3.70m
3.78m
3.86m
time in seconds
3.94m
4.02m
Figure 17. Zoom 1: 2nd to 3rd Valley Transition
1 vdrain
2 feedback
3 vct
Plot1
vdrain in volts
700
500
300
1
100
−100
Plot2
feedback in volts
1.38
2
1.34
1.30
1.26
1.22
4.00
vct i n volts
3.00
2.00
1.00
3
0
5.90m
5.95m
6.00m
time in seconds
6.05m
6.11m
Figure 18. Zoom 2: 3rd to 4th Valley Transition
1 vdrain
2 feedback
3 vct
Plot1
vdrain in volts
700
500
300
1
100
Plot2
feedback in volts
−100
1.12
1.02
2
919m
819m
719m
Plot3
vct in volts
8.00
6.00
4.00
3
2.00
0
7.10m
7.21m
7.32m
time in seconds
7.43m
7.55m
Figure 19. Zoom 3: 4th Valley to VCO Mode Transition
www.onsemi.com
24
NCP1937
VCO Mode
capacitor is charged with a constant current source, IQCT,
typically 20 mA.
The capacitor voltage, VQCT, is compared to an internal
voltage level, Vf(QFB), inversely proportional to VQFB The
relationship between and Vf(QFB) and VQFB is given by
Equation 6.
The controller enters VCO mode once VQFB falls below
VHVCOD and remains in VCO until VQFB exceeds VHVCOI.
In VCO mode the peak current is set to VQILIM1*Ipeak(VCO)
and the operating frequency is linearly dependent on VQFB.
The product of VQILIM1*Ipeak(VCO) is typically 12.5%. A
minimum frequency clamp, fVCO(MIN), typically 27 kHz,
prevents operation in the audible range. Further reduction in
output power causes the controller to enter skip operation.
The minimum frequency clamp is only enabled when
operating in VCO mode.
The VCO mode operating frequency is set by the timing
capacitor connected between the QCT and GND pins. This
V f (QFB) + 5 * 2 @ V QFB
A drive pulse is generated once VQCT exceeds Vf(QFB)
followed by the immediate discharge of the timing capacitor.
The timing capacitor is also discharged once the minimum
frequency clamp is reached.
Figure 20 shows simulation waveforms of Vf(QFB),
VQDRV and output current while operating in VCO mode.
800m
600m
IOUT
400m
200m
1
0
7.00
2
5.00
3
3.00
VQCT
Vf(QFB)
1.00
−1.00
30.0
VQDRV
20.0
10.0
0
5
−10.0
7.57m
7.78m
(eq. 6)
7.99m
time in seconds
8.20m
Figure 20. VCO Mode Operating Waveforms
www.onsemi.com
25
8.40m
NCP1937
Flyback Timeout
During startup, the voltage offset added by the overpower
compensation diode, DOPP, prevents the QZCD Comparator
from accurately detecting the valleys. In this condition, the
steady state timeout period will be shorter than the inductor
demagnetization period causing continuous current mode
(CCM) operation. CCM operation lasts for a few cycles until
the voltage on the QZCD pin is high enough to detect the
valleys. A longer timeout period, tQ(tout1), (typically 100 ms)
is set during soft−start to limit CCM operation. Figure 22
and Figure 23 show the timeout period generator related
waveforms.
In case of extremely damped oscillations, the QZCD
comparator may be unable to detect the valleys. In this
condition, drive pulses will stop waiting for the next valley
or ZCD event. The NCP1937 ensures continued operation
by incorporating a maximum timeout period after the last
demagnetization detection. The timeout signal is a substitute
for the ZCD signal for the valley counter. Figure 21 shows
the timeout period generator circuit schematic. The steady
state timeout period, tQ(tout2), is set at 6 ms to limit the
frequency step.
Figure 21. Timeout Period Generator Circuit Schematic
www.onsemi.com
26
NCP1937
3
4
high
The 3rd valley is
validated
VQZCD(th)
VQZCD
14
2nd, 3rd
low
high
The 3rd valley is not detected
by the QZCD Comparator
The 2nd valley is detected
by the QZCD Comparator
low
12
15
QZCD
Comparator
Output
16
Timeout
17
Clk
high
low
Timeout adds a pulse to account for
the missing 3rd valley
high
low
Figure 22. Timeout Operation with a Missing 3rd Valley
3
4
high
The 4th valley is
validated
VQZCD(th)
VQZCD
18
3rd, 4th
low
14
high
QZCD
low
15 Comparator
Output
high
16 Timeout
low
high
Timeout adds 2 pulses to account
for the missing 3rd and 4th valleys
low
17 Clk
Figure 23. Timeout Operation with Missing 3rd and 4th Valleys
www.onsemi.com
27
NCP1937
QR Flyback Current Sense and Overload
The Maximum Peak Current Comparator compares the
current sense signal to a reference voltage to limit the
maximum peak current of the system. The maximum peak
current reference voltage, VQILIM1, is typically 0.8 V. The
maximum peak current setpoint is reduced by the overpower
compensation circuitry. An overload condition causes the
output of the Maximum Peak Current Comparator to
transition high and enable the overload timer. Figure 24
shows the implementation of the current sensing circuitry.
The power switch on time is modulated by comparing a
ramp proportional to the switch current to VQFB/KQFB using
the PWM Comparator. The switch current is sensed across
a current sense resistor, RSENSE, and the resulting voltage is
applied to the QCS pin. The current signal is blanked by a
leading edge blanking (LEB) circuit. The blanking period
eliminates the leading edge spike and high frequency noise
during the switch turn−on event. The LEB period,
tQCS(LEB1), is typically 275 ns. The drive pulse terminates
once the current sense signal exceeds VQFB/KQFB.
VDD
VQFB
RQFB
In_PSM
IQFB
Skip
QFB
+
VQSKIP
Ipeak(VCO) =
KQCS(VCO)
−
VQZCD
PWM
Comparator
Overload Timer
tQOVLD
LEB
Peak Current
Comparator
Count Down
Count Up
/KQFB
+
tQCS(LEB1)
+
VQZCD
Fault
Over Current
Comparator
LEB
VDD
IQCS
QCS
tQCS(LEB2)
Count
Reset
+
−
CSStop
−
Counter
+
Disable QDRV
VQILIM1
QDRV
VQILIM2
Figure 24. Current Sensing Circuitry Schematic
and the overload timer counts down. The controller can latch
(options C and D) or allow for auto−recovery (options A and
B) once the overload timer expires. Auto−recovery requires
a VCC triple hiccup before the controller restarts. Figure 25
and Figure 26 show operating waveforms for latched and
auto−recovery overload conditions.
The overload timer integrates the duration of the overload
fault. That is, the timer count increases while the fault is
present and reduces its count once it is removed. The
overload timer duration, tQOVLD, is typically 80 ms. If both
the PWM and Maximum Peak Current Comparators toggle
at the same time, the PWM Comparator takes precedence
www.onsemi.com
28
NCP1937
Figure 25. Latched Overload Operation
Figure 26. Auto−recovery Overload Operation
www.onsemi.com
29
NCP1937
A severe overload fault like a secondary side winding
short−circuit causes the switch current to increase very
rapidly during the on−time. The current sense signal
significantly exceeds VQILIM1. But, because the current sense
signal is blanked by the LEB circuit during the switch turn
on, the system current can get extremely high causing
system damage.
The NCP1937 protects against this fault by adding an
additional comparator, Fault Overcurrent Comparator. The
current sense signal is blanked with a shorter LEB duration,
tQCS(LEB2), typically 120 ns, before applying it to the Fault
Overcurrent Comparator. The voltage threshold of the
comparator, VQILIM2, typically 1.2 V, is set 50% higher than
VQILIM1, to avoid interference with normal operation. Four
consecutive faults detected by the Fault Overcurrent
Comparator causes the controller to enter latch mode. The
count to 4 provides noise immunity during surge testing. The
counter is reset each time a QDRV pulse occurs without
activating the Fault Overcurrent Comparator. A 1 mA
(typically) pull−up current source, IQCS, pulls up the QCS
pin to disable the controller if the pin is left open.
ramps up from 0 V once the controller powers up. The
soft−start duration, tSSTART, is typically 4 ms.
During soft−start the timeout duration is extended and the
lower latch or OTP Comparator signal (typically for
overtemperature) is blanked. Soft−start ends once VSSTART
exceeds the peak current sense signal threshold.
QR Flyback Overpower Compensation
The input voltage of the QR flyback stage varies with the
line voltage and operating mode of the PFC converter. At
low line the PFC bulk voltage is 250 V and at high line it is
400 V. In addition, the PFC can be disabled at light loads to
reduce input power at which point the PFC bulk voltage is
set by the rectified peak line voltage.
An integrated overpower circuit provides a relative
constant output power across PFC bulk voltage, Vbulk. It
also reduces the variation on VQFB during the PFC stage
enable or disable transitions. Figure 27 shows the circuit
schematic for the overpower detector.
The auxiliary winding voltage during the power switch on
time is a reflection of the input voltage scaled by the primary
to auxiliary winding turns ratio, NP,AUX, as shown in
Figure 28.
QR Flyback Soft−Start
Soft−start is achieved by ramping up an internal reference,
VSSTART, and comparing it to current sense signal. VSSTART
QZCD
DOPP
RQCZD
LAUX
ROPPU
QZCD Comparator
+
−
ROPPL
QFB
VQZCD(th)
Peak Current
Comparator
+
/4
+
V
− QILIM1
+
Other Faults
KQCS(VCO)
QCS
Disable
QDRV
+
−
PWM
Comparator
LEB
tQCS(LEB1)
Figure 27. Overpower Compensation Circuit Schematic
Figure 28. Auxiliary Winding Voltage Waveform
www.onsemi.com
30
NCP1937
Overpower compensation is achieved by scaling down the
on−time reflected voltage and applying it to the QZCD pin.
The voltage is scaled down using ROPPU and ROPPL. The
negative voltage applied to the pin is referred to as VOPP.
The internal current setpoint is the sum of VOPP and peak
current sense threshold, VQILIM1. VOPP is also subtracted
from VQFB to compensate for the PWM Comparator delay
and improve the PFC on/off accuracy.
The current setpoint is calculated using Equation 7. For
example, a VOPP of −0.15 V results in a current setpoint of
0.65 V.
Current setpoint + V QILIM1 ) V OPP
Substituting values in Equation 8 and solving for ROPPU
we obtain,
R QZCD ) R OPPU
0.18 @ 370 * (−0.25)
+*
+ 271
(−0.25)
R OPPL
R OPPU + 271 @ R OPPL * R QZCD
R OPPU + 271 @ 1 k * 1 k + 270 k
Power Factor Correction
The PFC stage operates in critical conduction mode
(CrM). In CrM, the PFC inductor current, IL(t) reaches zero
at the end of each switch cycle. Figure 29 shows the PFC
inductor current while operating in CrM. The average input
current, Iin(t), is in phase with the ac line voltage, Vin(t), to
achieve power factor correction.
(eq. 7)
To ensure optimal zero−crossing detection, a diode is
needed to bypass ROPPU during the off−time. Equation 8 is
used to calculate ROPPU and ROPPL.
N P,AUX @ V bulk * V OPP
R QZCD ) R OPPU
+*
(eq. 8)
R OPPL
V OPP
ROPPU is selected once a value is chosen for ROPPL.
ROPPL is selected large enough such that enough voltage is
available for the zero crossing detection during the off−time.
It is recommended to have at least 8 V applied on the QZCD
pin for good detection. The maximum voltage is internally
clamped to VCC. The off−time voltage on the QZCD is given
by Equation 9.
V QZCD +
R OPPL
@ ǒV AUX * V FǓ
R QZCD ) R OPPL
(eq. 9)
Figure 29. Inductor Current in CrM
Where VAUX is the voltage across the auxiliary winding and
VF is the DOPP forward voltage drop.
The ratio between RQZCD and ROPPL is given by
Equation 10. It is obtained combining Equations 8 and 9.
V AUX * V F * V QZCD
R QZCD
+
R OPPL
V QZD
High power factor is achieved in CrM by maintaining a
constant on time (ton) for a given RMS input voltage
(Vac(RMS)) and load condition. Equation 11 shows the
relationship between on time and system operating conditions.
(eq. 10)
t on +
A design example is shown below:
System Parameters:
VAUX = 18 V
VF = 0.6 V
NP,AUX = 0.18
The ratio between RQZCD and ROPPL is calculated using
Equation 10 for a minimum VQZCD of 8 V.
2 @ P out @ L
h @ V ac(RMS) 2
(eq. 11)
where Pout is the output power, L is the PFC choke
inductance and η is the system efficiency.
PFC Feedback
The PFC feedback circuitry is shown in Figure 30. A
transconductance error amplifier regulates the PFC output
voltage, Vbulk, by comparing the PFC feedback signal to an
internal reference voltage, VPREF. The feedback signal is
applied to the inverting input and the reference is connected
to the non−inverting input of the error amplifier. A resistor
divider consisting of R1 and R2 scales down Vbulk to
generate the PFC feedback signal. VPREF is trimmed during
manufacturing to achieve an accuracy of ± 2%.
The PFC stage is disabled at light loads to reduce input power.
The NCP1937 integrates a 700 V switch, PFC FB Switch,
between the PFBHV and PFBLV pins. The PFC FB Switch
is in series between R1 and R2 to disconnect the resistors and
reduce input power when the PFC stage is disabled.
R QZCD
+ 18 * 0.6 * 8 [ 1.2
8
R OPPL
RQZCD is arbitrarily set to 1 kW. ROPPL is also set to 1 kW
because the ratio between the resistors is close to 1.
The NCP1937 maximum overpower compensation or
peak current setpoint reduction is 31.25% for a VOPP of
−250 mV. We will use this value for the following example:
www.onsemi.com
31
NCP1937
Vbulk
R1
PFBHV
PFC FB
Switch
PFC UVP
Comparator
PFBLV
PUVP
+
R2
−
VPREF
+
−
Delay
tPenable
VPFB(disable)
Error
Amplifier
Enable PFC
Latchoff
PILIM1, PILIM2
Stop
Q
R
Dominant
Reset
Latch
Q
S
PFC Faults
and
Disable Signals
PFCDRV
VDD
High
Clamp
PControl
Disable PFC
PUVP
Low
Clamp
Figure 30. PFC Regulation Circuit Schematic
The maximum on resistance of the PFC FB Switch,
RPFBswitch(on), is 10 kW. Because the PFC FB Switch is in
series with R1 and R1’s value is several orders of
magnitudes larger, the switch introduces minimum error on
the regulation level. The off state leakage current of the PFC
FB Switch, IPFBSwitch(off), is less than 3 mA.
The NCP1937 safely disables the controller if the PFBLV
pin is grounded. A short pin detector disables the controller
if VPFBLV is below the disable threshold, VPFB(disable),
typically 0.3 V. If the PFBLV pin is open, the PFC FB Switch
will raise VPFBLV above the overvoltage threshold and
disable the controller. Equation 12 shows the relationship
between the PFC output voltage, the PFC reference
threshold, R1 and R2.
V PFC + V PREF(xL) @ R1 ) R2
R2
fp +
gm
2pC PControl
(eq. 13)
where, CPControl is the capacitor on the PControl pin to
ground.
The output of the error amplifier is held low when the PFC
is disabled by means of an internal pull−down transistor. The
pull down transistor is disabled once the PFC stage is
enabled. An internal voltage clamp is then enabled to
quickly raise VPControl to its minimum voltage,
VPClamp(lower).
PFC On−Time
The PFC on time is controlled by VPControl. The PFC
On−Time Comparator terminates the drive pulse once the
PFC on−time ramp voltage plus offset exceeds VPControl.
The ramp is generated by charging an internal timing
capacitor, CPCT, with a constant current source, IPCT. The
capacitor ramp is level shifted to achieve 0 duty ratio (stop
drive pulses) at the minimum VPControl. VPControl is
proportional to the output power and it is fixed for a given
RMS line voltage and output load, satisfying Equation 11.
Lower and upper voltage clamps limit the excursion of
VPControl. The maximum on−time, ton(MAX), occurs when
VPControl is at its maximum value, VPControl(MAX). The PFC
drive pulses are inhibited once VPControl is below its
minimum value, VPControl(MIN). The maximum PFC
on−time in the NCP1937 is set internally. The maximum on
time at low line is typically 15 ms.
(eq. 12)
PFC Error Amplifier
A transconductance amplifier has a voltage−to−current
gain, gm. That is, the amplifier’s output current is controlled
by the differential input voltage. The NCP1937 amplifier
has a typical gm of 200 mS. The PControl pin provides access
to the amplifier output for compensation. The compensation
network is ground referenced allowing the PFC feedback
signal to be used to detect an overvoltage condition.
The compensation network on the PControl pin is selected
to filter the bulk voltage ripple such that a constant control
voltage is maintained across the ac line cycle. A capacitor
between the PControl pin and ground sets a pole. A pole at
or below 20 Hz is enough to filter the ripple voltage for a 50
and 60 Hz system. The low frequency pole, fp, of the system
is calculated using Equation 13.
www.onsemi.com
32
NCP1937
PFC Transient Response
The boost current source becomes active as soon as the
PFC is enabled. Coupled with the lower control clamp, the
current provided by the boost current source assists in
rapidly bringing VPControl to its set point to allow the bulk
voltage to quickly reach regulation. Achieving regulation is
detected by monitoring the error amplifier output current.
The error amplifier output current drops to zero once the
PFC output voltage reaches the target regulation level.
The maximum PFC output voltage is limited by the
overvoltage protection circuitry. The NCP1937
incorporates both soft and hard overvoltage protection. The
hard overvoltage protection function immediately
terminates and prevents further PFC drive pulses when
VPFBLV exceeds the hard−OVP level, VPREF(xL) *
KPOVP(xL). Soft−OVP reduces the on−time proportional to
the delta between VPFBLV and the hard−OVP level.
Soft−OVP is enabled once the delta, DPOVP(xL), between
VPFBLV and the hard−OVP level, is between 20 and 55 mV.
Figure 31 shows the circuit schematic of the boost and
Soft−OVP circuits.
The PFC bandwidth is set low enough to achieve good
power factor. A low bandwidth system is slow and fast load
transients can result in large output voltage excursions. The
NCP1937 incorporates dedicated circuitry to help mantain
regulation of the output voltage independent of load
transients.
An undervoltage detector monitors Vbulk and prevents it
from dropping below from its regulation level. Once the
ratio between VPFBLV and VPREF(xL) exceeds KLOW(PFCxL),
typically 5.5%, a pull−up current source on the PControl pin,
IPControl(boost), is enabled to speed up the charge of the
compensation capacitor(s). This results in an increased
on−time and thus output power. IPControl(boost) is typically
240 mA. The boost current source is disabled once the ratio
between VPFBLV and VPREF(xL) drops below KLOW(PFCxL),
typically 4%.
Figure 31. Boost and Soft−OVP Circuit Schematics
PFC Current Sense and Zero Current Detection
During power up, VPControl exceeds the regulation level
due to the system’s inherently low bandwidth. This causes
the bulk voltage to rapidly increase and exceed its
regulation. The on time starts to decrease when soft−OVP is
activated. Once the bulk voltage decreases to its regulation
level the PFC on time is no longer controlled by the
soft−OVP circuitry.
The NCP1937 uses a novel architecture combining the
PFC current sense and zero current detectors (ZCD) in a
single input terminal. Figure 32 shows the circuit schematic
of the current sense and ZCD detectors.
www.onsemi.com
33
NCP1937
tPFC(off) Timer
PDRV
ZCD
Comparator
PFC Boost
Diode
PFC Inductor
+
−
Reset
PILIM2
VDD
Q
ZCD_LAT
CLK
RPsense
R
PDRV
PCS/PZCD
D
Q
CLK Q
PFC Timeout
tPFC(out)
Reset
PFC Switch
RPZCD
D
VZCD
PDRV
RPCS
To PDRV Set
Frequency
Clamp
Timer
LEB1
tPCS(LEB1)
VPILIM1
Q
Current Limit
Comparator
+
PILIM1
PDRV
−
PDRV
LEB2
PILIM2
tPCS(LEB2)
Short Circuit
Detector
VPILIM2
Other Latching Faults
PDRVRST
Boost Diode
Counter
Count
Reset
Q
S
Dominant
Reset
Latch
Q
R
Latchoff
Figure 32. PFC Current Sense and ZCD Detectors Schematic
PFC Current Sense
than VPILIM1, to avoid interference with normal operation.
Four consecutive faults detected by the Short Circuit
Comparator causes the controller to enter latch mode. The
count to 4 provides noise immunity during surge testing. The
counter is reset each time a PDRV pulse occurs without
activating the Short Circuit Comparator.
The PFC watchdog timer duration increases to tPFC(off2)
(typically 1 ms) when a VPILIM2 fault is detected
independent of the PFC ZCD state.
The PFC Switch current is sensed across a sense resistor,
RPsense, and the resulting voltage ramp is applied to the
PCS/PZCD pin. The current signal is blanked by a leading
edge blanking (LEB) circuit. The blanking period eliminates
the leading edge spike and high frequency noise during the
switch turn−on event. The LEB period, tPCS(LEB1), is
typically 325 ns. The Current Limit Comparator disables the
PFC driver once the current sense signal exceeds the PFC
current sense reference, VPILIM1, typically 0.5 V.
A severe overload fault like a PFC boost diode short
circuit causes the switch current to increase very rapidly
during the on−time. The current sense signal significantly
exceeds VPILIM1. But, because the current sense signal is
blanked by the LEB circuit during the switch turn on, the
system current can get extremely high causing system
damage.
The NCP1937 protects against this fault by adding an
additional comparator, PFC Short Circuit Comparator. The
current sense signal is blanked with a shorter LEB duration,
tPCS(LEB2), typically 175 ns, before applying it to the PFC
Short Circuit Comparator. The voltage threshold of the
comparator, VPILIM2, typically 1.25 V, is set 250% higher
PFC Zero Current Detection
The off−time in a CrM PFC topology varies with the
instantaneous line voltage and is adjusted every switching
cycle to allow the inductor current to reach zero before the
next switching cycle begins. The inductor is demagnetized
once its current reaches zero. Once the inductor is
demagnetized the drain voltage of the PFC switch begins to
drop. The inductor demagnetization is detected by sensing
the voltage across the inductor using an auxiliary winding.
This winding is commonly known as a zero crossing
detector (ZCD) winding. This winding provides a scaled
version of the inductor voltage. Figure 33 shows the ZCD
winding arrangement.
www.onsemi.com
34
NCP1937
threshold. The watchdog timer is reset at the beginning of a
PFC drive pulse. It is disabled during a PFC hard
overvoltage and feedback input short circuit condition.
Figure 33. ZCD Winding Implementation
The ZCD voltage, VZCD, is positive while the PFC Switch
is off and current flows through the PFC inductor. VZCD
drops to and rings around zero volts once the inductor is
demagnetized. The next switching cycle begins once a
negative transition is detected on the PCS/PZCD pin. A
positive transition (corresponding to the PFC switch turn
off) arms the ZCD detector to prevent false triggering. The
arming of the ZCD detector, VPZCD(rising), is typically
0.75 V (VPCS/PZCD increasing). The trigger threshold,
VPZCD(falling), is typically 0.25 V (VPCS/PZCD decreasing).
The PCS/PZCD pin is internally clamped to 5 V with a
Zener diode and a 2 kW resistor. A resistor in series with the
PCS/PZCD pin is required to limit the current into pin. The
Zener diode also prevents the voltage from going below
ground. Figure 34 shows typical ZCD waveforms.
During startup there are no ZCD transitions to set the PFC
PWM Latch and generate a PDRV pulse. A watchdog timer,
tPFC(off1), starts the drive pulses in the absence of ZCD
transitions. Its duration is typically 200 ms. The timer is also
useful if the line voltage transitions from low line to high line
and while operating at light load because the amplitude of
the ZCD signal may be too small to cross the ZCD arming
Figure 34. ZCD Winding Waveforms
The watchdog timer duration increases to tPFC(off2),
typically 1 ms, when a VPILIM2 fault is detected.
PFC Frequency Clamp
The PFC operating frequency naturally increases when
the line voltage gets near to zero due to the reduced
demagnetization time or when the PFC is operating at light
loads. A maximum frequency clamp, fclamp(PFC), limits the
PFC frequency to improve efficiency and facilitate
compliance with EMI requirements. The NCP1937 has
options for PFC frequency clamp values of 131 kHz or
250 kHz.
The PDRV pulse is blanked until the frequency clamp
timer expires. Once expired, the controller waits for the next
ZCD transition to initiate PDRV. This ensures valley
switching to reduce switching losses. A timeout timer,
tP(tout), starts the next PDRV pulse in the absence of a ZCD
transition. The timeout timer duration is typically 10 ms. The
timer is reset at every ZCD event. Figure 35 shows the block
diagram of the PFC frequency clamp.
tPFC(off) Timer
PDRV
PFC Boost
Diode
PFC Inductor
PFC Switch
ZCD
Comparator
To PDRV Set
Frequency
Clamp
Timer
+
D
V
− ZCD
PDRV
Q
CLK Q
PFC Timeout
tPFC(out)
Reset
RPsense
RPCS
RPZCD
Reset
PILIM2
VDD
D
Q
CLK
PCS/PZCD
R
PDRV
Q
Figure 35. PFC Frequency Clamp Schematic
www.onsemi.com
35
ZCD_LAT
NCP1937
PFC Enable & Disable
The PONOFF pin voltage, VPONOFF, is compared to an
internal reference, VPOFF (typically 2 V) to disable the PFC
stage. The PFC disable point is typically set between 25 and
50% or between 50 and 75% of the maximum system load.
These setpoints provide the best system efficiency across
low line and high line.
Once VPONOFF decreases below VPOFF, the PFC disable
timer, tPdisable, is enabled. The NCP1937 has options for
500 ms, 4 s, or 13 s PFC disable timer. The PFC stage is
disabled once the timer expires. The PFC stage is enabled
once VPONOFF exceeds VPOFF by VPONHYS for a period
longer than the PFC enable filter, tPenable(filter), typically
100 ms. A shorter delay for the PFC enable threshold is used
to reduce the bulk capacitor requirements during a step load
response. Figure 36 shows the block diagram of the PFC
disable circuit.
In some applications it is desired to disable the PFC at
lighter loads to increase the overall system efficiency. The
NCP1937 integrates a novel architecture that allows the user
to program the PFC disable threshold based on the percentage
of QR output power. The PFC enable circuitry is inactive
until the QR flyback soft start period has ended. A voltage
to current (V−I) converter generates a current proportional
to VQFB. This current is pulse width modulated by the
demagnetization time of the flyback controller to generate
a current, IPONOFF, proportional to the output power. An
external resistor, RPONOFF, between the PONOFF and GND
pins generate a voltage proportional to the output power.
This resistor is used to scale the output power signal. A
capacitor, CPONOFF, in parallel with RPONOFF is required to
average the signal on this pin. A good compromise between
voltage ripple and speed is achieved by setting the time
constant of CPONOFF and RPONOFF to 160Ăms.
QFB
V to I
Converter
Demag
Time
Calculator
PFC Disable
Timer
Enable
QZCD
QDRV
tPdisable
Reset
IPONOFF
PONOFF
VPOFF
−
RCPONOFF
+
CPONOFF
PONOFF
Comparator
−
+
Soft−Start Complete
Q
S
Dominant
Reset
Latch
Q
R
Filter Delay
Release
PControl
Disable PFC
PFC Enable
Timer
tPenable
tPenable(filter)
Turn on
PFC FB Switch
Hysteresis Control
Figure 36. PFC On/Off Control Circuitry
PFC Skip
Auto Recovery
The PFC stage incorporates skip cycle operation at light
loads to reduce input power. Skip operation disables the PFC
stage if the PControl voltage decreases below the skip
threshold. The skip threshold voltage is typically 25 mV
(DVPSKIP) above the PControl lower voltage clamp,
Vclamp(lower). The PFC stage is enabled once VPControl
increases above the skip threshold by the skip hysteresis,
VPSKIP(HYS). PFC skip is disabled during any initial PFC
startup and when the PFC is in a UVP. Skip operation will
become active after the PFC has reached regulation.
The controller is disabled and enters “triple−hiccup”
mode if VCC drops below VCC(off). The controller will also
enter “triple−hiccup” mode if an overload fault is detected
on the non−latching version. A hiccup consists of VCC
falling down to VCC(off) and charging up to VCC(on). The
controller needs to complete 3 hiccups before restarting.
Temperature Shutdown
An internal thermal shutdown circuit monitors the
junction temperature of the IC. The controller is disabled if
the junction temperature exceeds the thermal shutdown
threshold, TSHDN, typically 150_C. A continuous VCC
hiccup is initiated after a thermal shutdown fault is detected.
The controller restarts at the next VCC(on) once the IC
temperature drops below below TSHDN by the thermal
shutdown hysteresis, TSHDN(HYS), typically 40_C.
The thermal shutdown fault is also cleared if VCC drops
below VCC(reset), a brown−out fault is detected or if the line
voltage is removed. A new power up sequences commences
at the next VCC(on) once all the faults are removed.
PFC and Flyback Drivers
The NCP1937 maximum supply voltage, VCC(MAX), is
30 V. Typical high voltage MOSFETs have a maximum gate
voltage rating of 20 V. Both the PFC and flyback drivers
incorporate an active voltage clamp to limit the gate voltage
on the external MOSFETs. The PFC and flyback voltage
clamps, VPDRV(high) and VQDRV(high), are typically 12 V
with a maximum limit of 14 V.
www.onsemi.com
36
NCP1937
PCB Layout Recommendations
therefore necessary to pay particular attention to the current
paths and grounding patterns to avoid interactions between
the two converters.
Before laying out a PCB for the NCP1937 it is
recommended to identify and annotate the various grounds
as shown in Figure 37. Table 5 below includes a description
of the different grounds. The grounds are divided into power
grounds, denoted as PGNDx, and analog or signal grounds,
denoted as AGNDx.
In any power converter, the PCB layout and routing
require consideration to minimize noise generation and
ensure stable operation. As a combo device, the NCP1937
controls two variable switching frequency converters that
operate independently of each other and can therefore
switch asynchronously. A turn−on or turn−off event of one
converter can occur at any point in the other converter’s
switching cycle possibly disrupting its operation. It is
Figure 37. Typical Application with Annotated Grounds
For the NCP1937 the following routing requirements are
recommended for the primary side power grounds:
• The current path from PGND3 to PGND4 and the
current path from PGND4 to PGND5 are isolated to the
greatest extent possible to provide separate paths for the
switching currents of the PFC and flyback converter.
This will avoid the switching currents and the gate
drive currents from the two converters overlapping
• Path between PGND6 and PGND1 is key for improved
surge performance. It is necessary to use a separate,
isolated trace to connect PGND6 back to PGND1.
Make this trace as wide as possible.
• The connection between PGND4 and PGND5 should
be as short as possible with as wide of a trace as
possible.
Table 5. DESCRIPTION OF ANNOTATED GROUND
NODES IN NCP1937 APPLICATION
Label
Location
PGND1
Diode bridge and input bulk ground
PGND2
PFC pi filter ground
PGND3
PFC current sense resistor ground
PGND4
Negative of PFC bulk cap ground
PGND5
current sense resistor ground of flyback
PGND6
Primary Y−cap and Aux winding ground
AGND1
All programming components of NCP1937
AGND2
Ground of VCC bulk capacitor
AGND3
PFC ZCD auxiliary winding
AGND
GND at the IC
www.onsemi.com
37
NCP1937
• PGND4 will be the center point of a star connection for
•
recommendations are not followed, it is possible that the
switching events from one converter can interrupt operation
of the other converter. One particular sensitivity that may
occur is that the PFC switching and gate drive currents can
interfere with QR current sense signal resulting in erratic
drive pulses. The QR current sense signal is particularly
critical for stable operation of the QR and RC filtering
decoupled to the appropriate ground should be employed.
When assessing noise on the QR current sense signal,
QCS, it is important to determine the noise source and take
appropriate measures.
• Self−generated noise, i.e., noise spikes generated by a
QR switching event, should be maintained to a duration
< 100 ns for proper QR operation.
• Coupled noise spikes generated by a PFC switching
event can generally lead to more disruptive behavior
and should be maintained to a duration < 20 ns for
proper QR operation.
• If the noise spikes on the QCS signal do not meet either
of the above requirements, or if improper switching
behavior of the QR is still observed, then it is
recommended to increase the time constant of the RC
filter going to the QCS pin.
the analog signal grounds. The trace connection
between PGND4 and AGND should be as short and
wide as possible.
The path between PGND1 and PGND2 and PGND3
can be sequential paths, i.e., it is not necessary to
isolate these paths.
Routing requirements for the primary side analog grounds:
• AGND is the NCP1937 IC ground and will be the
•
•
•
center point of the analog star configuration. The three
other analog grounds should intersect at this point. The
trace between AGND and PGND4 (bulk capacitor
ground) should be as short and wide as possible.
AGND3 which originates from the PFC choke auxiliary
winding should have its own isolated trace back to
AGND. To the greatest extent possible AGND1 &
AGND3 should not overlap except for their intersection
point at the AGND star.
AGND2 should its own trace back to the AGND star
intersection point and try to avoid overlap with the
other analog grounds.
Star ground connections are well known in the industry
and a good practice for optimal layouts. Figure 38 is an
example star grounding configuration for the primary
grounds
In summary:
• Layout is a critical consideration for power converter
operation. This is especially true with a combination
controller operating two power converters
asynchronously
• Follow the suggested grounding recommendation
outlined above. These recommendations are intended to
mitigate noise from one converter coupling onto the
sensitive control signals of the other converter
• An RC filter with a time constant of ≥ 100 ns should be
placed close to the QCS pin of the IC, with the
capacitor decoupled to AGND1 as shown above
• If any erratic drive operation of the QR is observed, it is
recommended to increase the time constant of the RC
filter. Time constants up to 250 – 300 ns are reasonable
Figure 38. Example Star Ground Configuration for
NCP1937
The above recommendations are meant to serve as a
general guideline for most applications. If the above
www.onsemi.com
38
MECHANICAL CASE OUTLINE
PACKAGE DIMENSIONS
SOIC20 NB LESS PINS 2, 4 & 19
CASE 751BS−01
ISSUE O
DATE 28 APR 2011
SCALE 1:1
0.10 C A-B
NOTE 4
D
NOTE 5
20
NOTE 4
E1
0.10 C A-B
D
A
11
L2
ÇÇÇÇ
ÇÇÇÇ
1
NOTE 5
NOTES:
1. DIMENSIONING AND TOLERANCING PER ASME Y14.5M, 1994.
2. CONTROLLING DIMENSION: MILLIMETERS.
3. DIMENSION b DOES NOT INCLUDE DAMBAR PROTRUSION.
ALLOWABLE PROTRUSION SHALL BE 0.10 TOTAL IN EXCESS
OF THE b DIMENSION. DIMENSION b APPLIES TO THE FLAT
PORTION OF THE LEAD AND SHALL BE MEASURED BETWEEN
0.13 AND 0.25 FROM THE TIP.
4. DIMENSIONS D AND E1 DO NOT INCLUDE MOLD FLASH,
PROTRUSIONS, OR GATE BURRS BUT DO INCLUDE MOLD
MISMATCH. MOLD FLASH, PROTRUSIONS, OR GATE BURRS
SHALL NOT EXCEED 0.15mm PER SIDE. DIMENSIONS D AND
E1 ARE DETERMINED AT DATUM H.
5. DATUMS A AND B ARE TO BE DETERMINED AT DATUM H.
6. A1 IS DEFINED AS THE VERTICAL DISTANCE FROM THE SEAT
ING PLANE TO THE LOWEST POINT ON THE PACKAGE BODY.
7. CHAMFER FEATURE IS OPTIONAL. IF NOT PRESENT, THEN A
PIN ONE IDENTIFIER MUST BE LOCATED IN THIS AREA.
L
E
SEATING
PLANE
C
M
DETAIL A
10
e
17X
B
b
0.25
0.20 C
C A-B D
M
NOTE 3
NOTE 7
h
0.10 C
h
DIM
A
A1
b
c
D
E
E1
e
h
L
L2
M
H
0.10 C
A
A1
C
RECOMMENDED
SOLDERING FOOTPRINT*
SEATING
PLANE
c
GENERIC
MARKING DIAGRAM*
17X
0.85
20
XXXXXXXXXXXXG
XXXXXXXXXXXXX
AWLYWW
11
20
1
6.30
1
XXXXX
A
WL
YY
WW
G
10
17X
0.53
1.00
PITCH
DIMENSIONS: MILLIMETERS
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
DOCUMENT NUMBER:
DESCRIPTION:
DETAIL A
98AON56699E
MILLIMETERS
MIN
MAX
--1.70
0.00
0.20
0.31
0.51
0.10
0.25
9.90 BSC
6.00 BSC
3.90 BSC
1.00 BSC
0.25
0.50
0.40
0.85
0.25 BSC
0_
8_
= Specific Device Code
= Assembly Location
= Wafer Lot
= Year
= Work Week
= Pb−Free Package
*This information is generic. Please refer to
device data sheet for actual part marking.
Pb−Free indicator, “G” or microdot “ G”,
may or may not be present.
Electronic versions are uncontrolled except when accessed directly from the Document Repository.
Printed versions are uncontrolled except when stamped “CONTROLLED COPY” in red.
SOIC20 NB LESS PINS 2, 4 & 19
PAGE 1 OF 1
ON Semiconductor and
are trademarks of Semiconductor Components Industries, LLC dba ON Semiconductor or its subsidiaries in the United States and/or other countries.
ON Semiconductor reserves the right to make changes without further notice to any products herein. ON Semiconductor makes no warranty, representation or guarantee regarding
the suitability of its products for any particular purpose, nor does ON Semiconductor assume any liability arising out of the application or use of any product or circuit, and specifically
disclaims any and all liability, including without limitation special, consequential or incidental damages. ON Semiconductor does not convey any license under its patent rights nor the
rights of others.
© Semiconductor Components Industries, LLC, 2019
www.onsemi.com
onsemi,
, and other names, marks, and brands are registered and/or common law trademarks of Semiconductor Components Industries, LLC dba “onsemi” or its affiliates
and/or subsidiaries in the United States and/or other countries. onsemi owns the rights to a number of patents, trademarks, copyrights, trade secrets, and other intellectual property.
A listing of onsemi’s product/patent coverage may be accessed at www.onsemi.com/site/pdf/Patent−Marking.pdf. onsemi reserves the right to make changes at any time to any
products or information herein, without notice. The information herein is provided “as−is” and onsemi makes no warranty, representation or guarantee regarding the accuracy of the
information, product features, availability, functionality, or suitability of its products for any particular purpose, nor does onsemi assume any liability arising out of the application or use
of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. Buyer is responsible for its products
and applications using onsemi products, including compliance with all laws, regulations and safety requirements or standards, regardless of any support or applications information
provided by onsemi. “Typical” parameters which may be provided in onsemi data sheets and/or specifications can and do vary in different applications and actual performance may
vary over time. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. onsemi does not convey any license
under any of its intellectual property rights nor the rights of others. onsemi products are not designed, intended, or authorized for use as a critical component in life support systems
or any FDA Class 3 medical devices or medical devices with a same or similar classification in a foreign jurisdiction or any devices intended for implantation in the human body. Should
Buyer purchase or use onsemi products for any such unintended or unauthorized application, Buyer shall indemnify and hold onsemi and its officers, employees, subsidiaries, affiliates,
and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death
associated with such unintended or unauthorized use, even if such claim alleges that onsemi was negligent regarding the design or manufacture of the part. onsemi is an Equal
Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
PUBLICATION ORDERING INFORMATION
LITERATURE FULFILLMENT:
Email Requests to: orderlit@onsemi.com
onsemi Website: www.onsemi.com
◊
TECHNICAL SUPPORT
North American Technical Support:
Voice Mail: 1 800−282−9855 Toll Free USA/Canada
Phone: 011 421 33 790 2910
Europe, Middle East and Africa Technical Support:
Phone: 00421 33 790 2910
For additional information, please contact your local Sales Representative
工商网监
湘ICP备2023018690号
