NE5204AN

INTEGRATED CIRCUITS NE/SA5204A Wide-band high-frequency amplifier Product specification RF Communications Handbook 1992 Feb 25 Philips Semiconductors Philips Semiconductors Product specification Wide-band high-frequency amplifier NE/SA5204A DESCRIPTION The NE/SA5204A family of wideband amplifiers replaces the NE/SA5204 family. The ‘A’ parts are fabricated on a rugged 2µm bipolar process featuring excellent statistical process control. Electrical performance is nomically identical to the original parts. The NE/SA5204A is a high-frequency amplifier with a fixed insertion gain of 20dB. The gain is flat to ±0.5dB from DC to 200MHz. The -3dB bandwidth is greater than 350MHz. This performance makes the amplifier ideal for cable TV applications. The NE/SA5204A operates with a single supply of 6V, and only draws 25mA of supply current, which is much less than comparable hybrid parts. The noise figure is 4.8dB in a 75Ω system and 6dB in a 50Ω system. The NE/SA5204A is a relaxed version of the NE5205. Minimum guaranteed bandwidth is relaxed to 350MHz and the “S” parameter Min/Max limits are specified as typicals only. Until now, most RF or high-frequency designers had to settle for discrete or hybrid solutions to their amplification problems. Most of these solutions required trade-offs that the designer had to accept in order to use high-frequency gain stages. These include high power consumption, large component count, transformers, large packages with heat sinks, and high part cost. The NE/SA5204A solves these problems by incorporating a wideband amplifier on a single monolithic chip. The part is well matched to 50 or 75Ω input and output impedances. The standing wave ratios in 50 and 75Ω systems do not exceed 1.5 on either the input or output over the entire DC to 350MHz operating range. Since the part is a small, monolithic IC die, problems such as stray capacitance are minimized. The die size is small enough to fit into a very cost-effective 8-pin small-outline (SO) package to further reduce parasitic effects. No external components are needed other than AC-coupling capacitors because the NE/SA5204A is internally compensated and matched to 50 and 75Ω. The amplifier has very good distortion specifications, with second and third-order intermodulation intercepts of +24dBm and +17dBm, respectively, at 100MHz. The part is well matched for 50Ω test equipment such as signal generators, oscilloscopes, frequency counters, and all kinds of signal analyzers. Other applications at 50Ω include mobile radio, CB radio, and data/video transmission in fiber optics, as well as broadband LANs and telecom systems. A gain greater than 20dB can be achieved by cascading additional NE/SA5204As in series as required, without any degradation in amplifier stability. PIN CONFIGURATION N, D Packages VCC VIN GND GND 1 20dB 2 3 4 TOP VIEW 8 7 6 5 VCC VOUT GND GND SR00193 Figure 1. Pin Configuration FEATURES • Bandwidth (min.) 200 MHz, ±0.5dB 350 MHz, -3dB • 20dB insertion gain • 4.8dB (6dB) noise figure ZO=75Ω (ZO=50Ω) • No external components required • Input and output impedances matched to 50/75Ω systems • Surface-mount package available • Cascadable • 2000V ESD protection APPLICATIONS • Antenna amplifiers • Amplified splitters • Signal generators • Frequency counters • Oscilloscopes • Signal analyzers • Broadband LANs • Networks • Modems • Mobile radio • Security systems • Telecommunications ORDERING INFORMATION DESCRIPTION 8-Pin Plastic Dual In-Line Package (DIP) 8-Pin Plastic Small Outline (SO) package 8-Pin Plastic Dual In-Line Package (DIP) 8-Pin Plastic Small Outline (SO) package TEMPERATURE RANGE 0 to +70°C 0 to +70°C –40 to +85°C –40 to +85°C ORDER CODE NE5204AN NE5204AD SA5204AN SA5204AD DWG # SOT97-1 SOT96-1 SOT97-1 SOT96-1 1992 Feb 25 2 853-1599 05790 Philips Semiconductors Product specification Wide-band high-frequency amplifier NE/SA5204A ABSOLUTE MAXIMUM RATINGS SYMBOL VCC VIN TA Supply voltage AC input voltage Operating ambient temperature range NE grade SA grade PDMAX Maximum power dissipation1, 2 TA=25°C(still–air) N package D package TJ TSTG TSOLD Junction temperature Storage temperature range Lead temperature (soldering 60s) 1160 780 150 –55 to +150 300 mW mW °C °C °C 0 to +70 –40 to +85 °C °C PARAMETER RATING 9 5 UNIT V VP–P NOTES: 1. Derate above 25°C, at the following rates N package at 9.3mW/°C D package at 6.2mW/°C 2. See “Power Dissipation Considerations” section. EQUIVALENT SCHEMATIC VCC R1 Q3 Q6 R2 R0 VOUT Q2 VIN Q1 RF1 RE2 RE1 Q4 R3 Q5 RF2 SR00194 Figure 2. Equivalent Schematic 1992 Feb 25 3 Philips Semiconductors Product specification Wide-band high-frequency amplifier NE/SA5204A DC ELECTRICAL CHARACTERISTICS VCC=6V, ZS=ZL=ZO=50Ω and TA=25°C, in all packages, unless otherwise specified. SYMBOL VCC ICC S21 S11 S22 S12 BW BW PARAMETER Operating supply voltage range Supply current Insertion gain Input return loss Input return loss Output return loss Output return loss Isolation Isolation Bandwidth Bandwidth Noise figure (75Ω) Noise figure (50Ω) Saturated output power 1dB gain compression Third–order intermodulation intercept (output) Second–order intermodulation intercept (output) tR tP Rise time Propagation delay TEST CONDITIONS CONDITIONS Over temperature Over temperature f=100MHz, over temperature f=100MHz DC –550MHz f=100MHz DC –550MHz f=100MHz DC –550MHz ±0.5dB –3dB f=100MHz f=100MHz f=100MHz f=100MHz f=100MHz f=100MHz 200 350 LIMITS Min 5 19 16 25 19 25 12 27 12 –25 –18 350 550 4.8 6.0 +7.0 +4.0 +17 +24 500 500 Typ Max 8 33 22 UNIT V mA dB dB dB dB MHz MHz dB dB dBm dBm dBm dBm ps ps 35 34 SUPPLY CURRENT—mA NOISE FIGURE—dBm 32 30 28 26 24 22 20 18 16 5 5.5 6 6.5 7 7.5 8 SUPPLY VOLTAGE—V TA = 25oC 9 8 vcc = 8v vcc = 7v vcc = 6v vcc = 5v ZO = 50Ω TA = 25oC 7 6 5 101 2 4 SR00195 6 8 102 2 FREQUENCY—MHz 4 6 8 103 SR00196 Figure 3. Supply Current vs Supply Voltage Figure 4. Noise Figure vs Frequency 1992 Feb 25 4 Philips Semiconductors Product specification Wide-band high-frequency amplifier NE/SA5204A 25 vcc = 8v INSERTION GAIN—dB 20 OUTPUT LEVEL—dBm vcc = 7v vcc = 6v 15 ZO = 50Ω TA = 25oC 10 101 vcc = 5v 10 9 8 7 6 5 4 3 2 1 0 –1 –2 –3 –4 –5 –6 101 VCC = 8V VCC = 6V VCC = 7V VCC = 5V ZO = 50Ω TA = 25oC 2 4 6 8 102 2 FREQUENCY—MHz 4 6 8 103 SR00197 2 4 6 8 102 2 FREQUENCY—MHz 4 6 8 103 SR00198 Figure 5. Insertion Gain vs Frequency (S21) Figure 8. 1dB Gain Compression vs Frequency SECOND–ORDER INTERCEPT—dBm 25 TA = 55oC TA = 25oC 20 40 35 30 25 20 15 10 4 5 6 7 8 9 10 POWER SUPPLY VOLTAGE—V ZO = 50Ω TA = 25oC INSERTION GAIN—dB TA = 85oC 15 VCC = 8V ZO = 50Ω 10 101 TA = 125oC 2 4 6 8 102 2 4 6 8 103 FREQUENCY—MHz SR00199 SR00200 Figure 6. Insertion Gain vs Frequency (S21) Figure 9. Second-Order Output Intercept vs Supply Voltage 11 10 9 8 7 6 5 4 3 2 1 0 –1 –2 –3 –4 –5 –6 101 30 THIRD–ORDER INTERCEPT—dBm 25 OUTPUT LEVEL—dBm VCC = 7V VCC = 6V VCC = 5V VCC = 8V 20 ZO = 50Ω TA = 25oC 15 10 ZO = 50Ω TA = 25oC 5 2 4 6 8 102 2 4 6 8 103 4 5 6 7 8 9 10 FREQUENCY—MHz POWER SUPPLY VOLTAGE—V SR00201 SR00202 Figure 7. Saturated Output Power vs Frequency Figure 10. Third-Order Intercept vs Supply Voltage 1992 Feb 25 5 Philips Semiconductors Product specification Wide-band high-frequency amplifier NE/SA5204A 2.0 1.9 1.8 INPUT VSWR 1.7 1.6 1.5 1.4 1.3 1.2 1.1 1.0 101 ZO = 75Ω ZO = 50Ω TA = 25oC VCC = 6V 10 . ISOLATION—dB –15 ZO = 50Ω TA = 25oC VCC = 6V –20 –25 –30 2 4 6 8 102 2 4 6 8 103 SR00203 101 2 4 6 8 102 2 4 6 8 103 FREQUENCY—MHz FREQUENCY—MHz SR00204 Figure 11. Input VSWR vs Frequency 25 2.0 1.9 1.8 INPUT VSWR 1.7 1.6 1.5 1.4 1.3 1.2 1.1 1.0 101 ZO = 50Ω 10 ZO = 75Ω Tamb = 25oC VCC = 6V ISOLATION GAIN—dB 20 Figure 14. Isolation vs Frequency (S12) vcc = 8v vcc = 7v vcc = 6v 15 ZO = 75Ω TA = 25oC 101 vcc = 5v 2 4 2 4 6 8 102 2 4 6 8103 SR00205 6 8 102 2 FREQUENCY—MHz 4 6 8 103 FREQUENCY—MHz SR00206 Figure 12. Output VSWR vs Frequency Figure 15. Insertion Gain vs Frequency (S21) 40 OUTPUT RETURN LOSS—dB INPUT RETURN LOSS—dB INSERTION GAIN—dB 35 30 OUTPUT 25 20 15 10 101 VCC = 6V ZO = 50Ω TA = 25oC 25 TA = –55oC TA = 25oC 20 TA = 85oC TA = 125oC ZO = 75Ω VCC = 6V INPUT 15 2 4 6 8 102 2 4 6 8103 10 101 2 4 6 8 102 2 4 6 8 103 FREQUENCY—MHz FREQUENCY—MHz SR00207 SR00208 Figure 13. Input (S11) and Output (S22) Return Loss vs Frequency Figure 16. Insertion Gain vs Frequency (S21) 1992 Feb 25 6 Philips Semiconductors Product specification Wide-band high-frequency amplifier NE/SA5204A THEORY OF OPERATION The design is based on the use of multiple feedback loops to provide wide-band gain together with good noise figure and terminal impedance matches. Referring to the circuit schematic in Figure 17, the gain is set primarily by the equation: V OUT V IN + (R F1 ) R E1) eliminates problems of shunt-feedback loading on the output. The value of RF1=140Ω is chosen to give the desired nominal gain. The DC output voltage VOUT can be determined by: VOUT=VCC–(IC2+IC6)R2,(4) where VCC=6V, R2=225Ω, IC2=8mA and IC6=5mA. From here, it can be seen that the output voltage is approximately 3.1V to give relatively equal positive and negative output swings. Diode Q5 is included for bias purposes to allow direct coupling of RF2 to the base of Q1. The dual feedback loops stabilize the DC operating point of the amplifier. The output stage is a Darlington pair (Q6 and Q2) which increases the DC bias voltage on the input stage (Q1) to a more desirable value, and also increases the feedback loop gain. Resistor R0 optimizes the output VSWR (Voltage Standing Wave Ratio). Inductors L1 and L2 are bondwire and lead inductances which are roughly 3nH. These improve the high-frequency impedance matches at input and output by partially resonating with 0.5pF of pad and package capacitance. R E1 (1) which is series-shunt feedback. There is also shunt-series feedback due to RF2 and RE2 which aids in producing wide-band terminal impedances without the need for low value input shunting resistors that would degrade the noise figure. For optimum noise performance, RE1 and the base resistance of Q1 are kept as low as possible, while RF2 is maximized. The noise figure is given by the following equation: KT 2ql C1 r b ) R E1 ) NF + 10Log 1) RO dB (2) POWER DISSIPATION CONSIDERATIONS where IC1=5.5mA, RE1=12Ω, rb=130Ω, KT/q=26mV at 25°C and R0=50 for a 50Ω system and 75 for a 75Ω system. The DC input voltage level VIN can be determined by the equation: VIN=VBE1+(IC1+IC3) RE1(3) where RE1=12Ω, VBE=0.8V, IC1=5mA and IC3=7mA (currents rated at VCC=6V). Under the above conditions, VIN is approximately equal to 1V. Level shifting is achieved by emitter-follower Q3 and diode Q4, which provide shunt feedback to the emitter of Q1 via RF1. The use of an emitter-follower buffer in this feedback loop essentially VCC When using the part at elevated temperature, the engineer should consider the power dissipation capabilities of each package. At the nominal supply voltage of 6V, the typical supply current is 25mA (32mA max). For operation at supply voltages other than 6V, see Figure 3 for ICC versus VCC curves. The supply current is inversely proportional to temperature and varies no more than 1mA between 25°C and either temperature extreme. The change is 0.1% per °C over the range. The recommended operating temperature ranges are air-mount specifications. Better heat-sinking benefits can be realized by mounting the SO and N package bodies against the PC board plane. R1 650 Q3 R2 225 R0 10 L2 3nH VOUT Q6 VIN L1 Q1 3nH Q4 R3 140 RF1 140 RE1 12 RE2 12 Q2 Q5 RF2 200 SR00209 Figure 17. Schematic Diagram 1992 Feb 25 7 Philips Semiconductors Product specification Wide-band high-frequency amplifier NE/SA5204A PC BOARD MOUNTING In order to realize satisfactory mounting of the NE5204A to a PC board, certain techniques need to be utilized. The board must be double-sided with copper and all pins must be soldered to their respective areas (i.e., all GND and VCC pins on the package). The power supply should be decoupled with a capacitor as close to the VCC pins as possible, and an RF choke should be inserted between the supply and the device. Caution should be exercised in the connection of input and output pins. Standard microstrip should be observed wherever possible. There should be no solder bumps or burrs or any obstructions in the signal path to cause launching problems. The path should be as straight as possible and lead lengths as short as possible from the part to the cable connection. Another important consideration is that the input and output should be AC-coupled. This is because at VCC=6V, the input is approximately at 1V while the output is at 3.1V. The output must be decoupled into a low-impedance system, or the DC bias on the output of the amplifier will be loaded down, causing loss of output power. The easiest way to decouple the entire amplifier is by soldering a high-frequency chip capacitor directly to the input and output pins of the device. This circuit is shown in Figure 18. Follow these recommendations to get the best frequency response and noise immunity. The board design is as important as the integrated circuit design itself. Actual S-parameter measurements using an HP network analyzer (model 8505A) and an HP S-parameter tester (models 8503A/B) are shown in Figure 20. Values for the figures below are measured and specified in the data sheet to ease adaptation and comparison of the NE/SA/SE5204A to other high-frequency amplifiers. The most important parameter is S21. It is defined as the square root of the power gain, and, in decibels, is equal to voltage gain as shown below: ZD=ZIN=ZOUT for the NE/SA/SE5204A V IN ) ZD 2 NE5204A ZD V OUT ZD V IN ZD 2 2 P IN P OUT ) V OUT ZD 2 N P OUT P IN + + V OUT V IN 2 2 + PI PI=VI 2 PI=Insertion Power Gain SCATTERING PARAMETERS The primary specifications for the NE5204A are listed as S-parameters. S-parameters are measurements of incident and reflected currents and voltages between the source, amplifier, and load as well as transmission losses. The parameters for a two-port network are defined in Figure 19. VI=Insertion Voltage Gain Measured value for the NE/SA/SE5204A = |S21 | 2 = 100 + | S 21 | 2 + 100 P IN V OUT + P I + S 21 + 10 and V I + V IN In decibels: PI(dB) =10 Log | S21 | 2 = 20dB NP I + P OUT VCC RF CHOKE DECOUPLING CAPACITOR VIN AC COUPLING CAPACITOR NE5204A AC COUPLING CAPACITOR VOUT VI(dB) = 20 Log S21 = 20dB ∴ PI(dB) = VI(dB) = S21(dB) = 20dB Also measured on the same system are the respective voltage standing wave ratios. These are shown in Figure 21. The VSWR can be seen to be below 1.5 across the entire operational frequency range. SR00210 Figure 18. Circuit Schematic for Coupling and Power Supply Decoupling Relationships exist between the input and output return losses and the voltage standing wave ratios. These relationships are as follows: 1992 Feb 25 8 Philips Semiconductors Product specification Wide-band high-frequency amplifier NE/SA5204A POWER REFLECTED FROM INPUT PORT S11 — INPUT RETURN LOSS S21 S11 = POWER AVAILABLE FROM GENERATOR AT INPUT PORT S11 S22 S12 — REVERSE TRANSMISSION LOSS OSOLATION S12 = REVERSE TRANSDUCER POWER GAIN S21 — FORWARD TRANSMISSION LOSS OR INSERTION GAIN S12 S22 — OUTPUT RETURN LOSS S21 = TRANSDUCER POWER GAIN S22 = POWER REFLECTED FROM OUTPUT PORT POWER AVAILABLE FROM GENERATOR AT OUTPUT PORT a. Two-Port Network Defined Figure 19. b. SR00211 1992 Feb 25 9 Philips Semiconductors Product specification Wide-band high-frequency amplifier NE/SA5204A 50Ω System 25 vcc = 8v INSERTION GAIN—dB vcc = 7v 20 ISOLATION GAIN—dB 20 25 75Ω System vcc = 8v vcc = 7v vcc = 6v 15 ZO = 50Ω TA = 25oC 10 101 vcc = 5v vcc = 6v 15 ZO = 75Ω TA = 25oC 10 101 vcc = 5v 2 4 6 8 102 2 4 6 8 103 2 4 6 8 102 2 4 6 8 103 FREQUENCY—MHz FREQUENCY—MHz a. Insertion Gain vs Frequency (S21) 10 10 b. Insertion Gain vs Frequency (S21) ISOLATION—dB ISOLATION—dB –15 ZO = 50Ω TA = 25oC VCC = 6V –15 –20 –20 ZO = 75Ω TA = 25oC VCC = 6V –25 –25 –30 –30 101 2 4 6 8 102 2 4 6 8 103 101 2 4 6 8 102 2 4 6 8 103 FREQUENCY—MHz FREQUENCY—MHz c. Isolation vs Frequency (S12) 40 INPUT RETURN LOSS—dB OUTPUT RETURN LOSS—dB INPUT RETURN LOSS—dB OUTPUT RETURN LOSS—dB 35 30 OUTPUT 25 20 15 10 101 VCC = 6V ZO = 50Ω TA = 25oC 40 35 30 25 20 d. S12 Isolation vs Frequency OUTPUT INPUT INPUT 15 10 VCC = 6V ZO = 75Ω TA = 25oC 2 4 6 8 102 2 4 6 8 103 101 2 4 6 8 102 2 4 6 8 103 FREQUENCY—MHz FREQUENCY—MHz e. Input (S11) and Output (S22) Return Loss vs Frequency f. Input (S11) and Output (S22) Return Loss vs Frequency SR00212 INPUT RETURN LOSS=S11dB S11dB=20 Log | S11 | OUTPUT RETURN LOSS=S22dB S22dB=20 Log | S22 | INPUT VSWR=≤1.5 OUTPUT VSWR=≤1.5 Figure 20. 1dB from its low power value. The decrease is due to nonlinearities in the amplifier, an indication of the point of transition between small-signal operation and the large signal mode. The saturated output power is a measure of the amplifier’s ability to deliver power into an external load. It is the value of the amplifier’s output power when the input is heavily overdriven. This includes the sum of the power in all harmonics. INTERMODULATION INTERCEPT TESTS 1DB GAIN COMPRESSION AND SATURATED OUTPUT POWER The 1dB gain compression is a measurement of the output power level where the small-signal insertion gain magnitude decreases 1992 Feb 25 10 The intermodulation intercept is an expression of the low level linearity of the amplifier. The intermodulation ratio is the difference in dB between the fundamental output signal level and the generated distortion product level. The relationship between intercept and Philips Semiconductors Product specification Wide-band high-frequency amplifier NE/SA5204A intermodulation ratio is illustrated in Figure 22, which shows product output levels plotted versus the level of the fundamental output for two equal strength output signals at different frequencies. The upper line shows the fundamental output plotted against itself with a 1dB to 1dB slope. The second and third order products lie below the fundamentals and exhibit a 2:1 and 3:1 slope, respectively. The intercept point for either product is the intersection of the extensions of the product curve with the fundamental output. The intercept point is determined by measuring the intermodulation ratio at a single output level and projecting along the appropriate product slope to the point of intersection with the fundamental. When the intercept point is known, the intermodulation ratio can be determined by the reverse process. The second order IMR is equal to the difference between the second order intercept and the fundamental output level. The third order IMR is equal to twice the difference between the third order intercept and the fundamental output level. These are expressed as: IP2=POUT+IMR2 IP3=POUT+IMR3/2 where POUT is the power level in dBm of each of a pair of equal level fundamental output signals, IP2 and IP3 are the second and third order output intercepts in dBm, and IMR2 and IMR3 are the 2.0 1.9 1.8 INPUT VSWR 1.7 1.6 1.5 1.4 1.3 1.2 1.1 1.0 101 ZO = 75Ω ZO = 50Ω TA = 25oC VCC = 6V second and third order intermodulation ratios in dB. The intermodulation intercept is an indicator of intermodulation performance only in the small signal operating range of the amplifier. Above some output level which is below the 1dB compression point, the active device moves into large-signal operation. At this point the intermodulation products no longer follow the straight line output slopes, and the intercept description is no longer valid. It is therefore important to measure IP2 and IP3 at output levels well below 1dB compression. One must be careful, however, not to select too low levels because the test equipment may not be able to recover the signal from the noise. For the NE/SA5204A we have chosen an output level of –10.5dBm with fundamental frequencies of 100.000 and 100.01MHz, respectively. ADDITIONAL READING ON SCATTERING PARAMETERS For more information regarding S-parameters, please refer to High-Frequency Amplifiers by Ralph S. Carson of the University of Missouri, Rolla, Copyright 1985; published by John Wiley & Sons, Inc. “S-Parameter Techniques for Faster, More Accurate Network Design”, HP App Note 95-1, Richard W. Anderson, 1967, HP Journal. “S-Parameter Design”, HP App Note 154, 1972. 2.0 1.9 1.8 . INPUT VSWR 1.7 1.6 1.5 1.4 1.3 1.2 1.1 ZO = 50Ω 1.0 101 ZO = 75Ω Tamb = 25oC VCC = 6V 2 4 6 8 102 2 FREQUENCY—MHz 4 6 8 103 2 4 6 8 102 2 FREQUENCY—MHz 4 6 8 103 SR00213 a. Input VSWR vs Frequency b. Output VSWR vs Frequency Figure 21. Input/Output VSWR vs Frequency +30 THIRD ORDER INTERCEPT POINT 1dB COMPRESSION POINT 2ND ORDER INTERCEPT POINT +20 +10 OUTPUT LEVEL dBm 0 FUNDAMENTAL RESPONSE -10 2ND ORDER RESPONSE 3RD ORDER RESPONSE -20 -30 -40 -60 -50 -40 -30 -20 -10 0 +10 +20 +30 +40 INPUT LEVEL dBm SR00214 Figure 22. 1992 Feb 25 11 Philips Semiconductors Product specification Wide-band high-frequency amplifier NE/SA5204A SO8: plastic small outline package; 8 leads; body width 3.9mm SOT96-1 1992 Feb 25 12 Philips Semiconductors Product specification Wide-band high-frequency amplifier NE/SA5204A DIP8: plastic dual in-line package; 8 leads (300 mil) SOT97-1 1992 Feb 25 13 Philips Semiconductors Product specification Wide-band high-frequency amplifier NE/SA5204A DEFINITIONS Data Sheet Identification Objective Specification Product Status Formative or in Design Definition This data sheet contains the design target or goal specifications for product development. Specifications may change in any manner without notice. This data sheet contains preliminary data, and supplementary data will be published at a later date. Philips Semiconductors reserves the right to make changes at any time without notice in order to improve design and supply the best possible product. This data sheet contains Final Specifications. Philips Semiconductors reserves the right to make changes at any time without notice, in order to improve design and supply the best possible product. Preliminary Specification Preproduction Product Product Specification Full Production Philips Semiconductors and Philips Electronics North America Corporation reserve the right to make changes, without notice, in the products, including circuits, standard cells, and/or software, described or contained herein in order to improve design and/or performance. Philips Semiconductors assumes no responsibility or liability for the use of any of these products, conveys no license or title under any patent, copyright, or mask work right to these products, and makes no representations or warranties that these products are free from patent, copyright, or mask work right infringement, unless otherwise specified. Applications that are described herein for any of these products are for illustrative purposes only. Philips Semiconductors makes no representation or warranty that such applications will be suitable for the specified use without further testing or modification. LIFE SUPPORT APPLICATIONS Philips Semiconductors and Philips Electronics North America Corporation Products are not designed for use in life support appliances, devices, or systems where malfunction of a Philips Semiconductors and Philips Electronics North America Corporation Product can reasonably be expected to result in a personal injury. Philips Semiconductors and Philips Electronics North America Corporation customers using or selling Philips Semiconductors and Philips Electronics North America Corporation Products for use in such applications do so at their own risk and agree to fully indemnify Philips Semiconductors and Philips Electronics North America Corporation for any damages resulting from such improper use or sale. Philips Semiconductors 811 East Arques Avenue P.O. Box 3409 Sunnyvale, California 94088–3409 Telephone 800-234-7381 Philips Semiconductors and Philips Electronics North America Corporation register eligible circuits under the Semiconductor Chip Protection Act. © Copyright Philips Electronics North America Corporation 1993 All rights reserved. Printed in U.S.A. 1992 Feb 25 14