TOP252-262
TOPSwitch-HX Family
™
Enhanced EcoSmart™, Integrated Off-Line Switcher with
Advanced Feature Set and Extended Power Range
Product Highlights
Lower System Cost, Higher Design Flexibility
• Multi-mode operation maximizes efficiency at all loads
• New eSIP-7F and eSIP-7C packages
• Low thermal impedance junction-to-case (2 °C per watt)
• Low height is ideal for adapters where space is limited
• Simple mounting using a clip to aid low cost manufacturing
• Horizontal eSIP-7F package ideal for ultra low height adapter
and monitor applications
• Extended package creepage distance from DRAIN pin to
adjacent pin and to heat sink
• No heat sink required up to 35 W using P, G and M packages
with universal input voltage and up to 48 W at 230 VAC
• Output overvoltage protection (OVP) is user programmable for
latching/non-latching shutdown with fast AC reset
• Allows both primary and secondary sensing
• Line undervoltage (UV) detection prevents turn-off glitches
• Line overvoltage (OV) shutdown extends line surge limit
• Accurate programmable current limit
• Optimized line feed-forward for line ripple rejection
• 132 kHz frequency (254Y-258Y and all E/L packages) reduces
transformer and power supply size
• Half frequency option for video applications
• Frequency jittering reduces EMI filter cost
+
DC
OUT
-
AC
IN
V
D
CONTROL
TOPSwitch-HX
S
X
C
F
PI-4510-100206
Figure 1.
•
•
•
•
•
Typical Flyback Application.
Heat sink is connected to SOURCE for low EMI
Improved auto-restart delivers 600 mW output at 110 VAC input
• >500 mW output at 265 VAC input
Y Package Option for TOP259-261
In order to improve noise-immunity on large TOPSwitch-HX
Y package parts, the F pin has been removed (TOP259-261YN
are fixed at 66 kHz switching frequency) and replaced with a
SIGNAL GROUND (G) pin. This pin acts as a low noise path for
the C pin capacitor and the X pin resistor. It is only required for
the TOP259-261YN package parts.
Description
TOPSwitch-HX cost effectively incorporates a 700 V power
MOSFET, high voltage switched current source, PWM control,
oscillator, thermal shutdown circuit, fault protection and other
control circuitry onto a monolithic device.
Notes for Table 1:
1. Minimum continuous power in a typical non-ventilated
enclosed adapter measured at +50 °C ambient. Use of an
external heat sink will increase power capability.
2. Minimum continuous power in an open frame design at
+50 °C ambient.
3. Peak power capability in any design at +50 °C ambient.
4. 230 VAC or 110/115 VAC with doubler.
5. Packages: P: DIP-8C, G: SMD-8C, M: SDIP-10C,
Y: TO-220-7C, E: eSIP-7C, L: eSIP-7F.
See part ordering information.
6. TOP261 and TOP262 have the same current limit set point. In
some applications TOP262 may run cooler than TOP261 due
to a lower RDS(ON) for the larger device.
+
DC
OUT
-
AC
IN
V
D
CONTROL
TOPSwitch-HX
S
X
C
G
PI-4973-122607
Figure 2. Typical Flyback Application TOP259YN, TOP260YN and TOP261YN.
2
Rev. H 06/13
www.powerint.com
TOP252-262
Section List
Functional Block Diagram ........................................................................................................................................ 4
Pin Functional Description ....................................................................................................................................... 6
TOPSwitch-HX Family Functional Description ........................................................................................................ 7
CONTROL (C) Pin Operation..................................................................................................................................... 8
Oscillator and Switching Frequency........................................................................................................................... 8
Pulse Width Modulator ............................................................................................................................................. 9
Maximum Load Cycle............................................................................................................................................... 9
Error Amplifier........................................................................................................................................................... 9
On-Chip Current Limit with External Programmability................................................................................................ 9
Line Undervoltage Detection (UV)............................................................................................................................ 10
Line Overvoltage Shutdown (OV)............................................................................................................................. 11
Hysteretic or Latching Output Overvoltage Protection (OVP)................................................................................... 11
Line Feed-Forward with DCMAX Reduction............................................................................................................... 13
Remote ON/OFF and Synchronization..................................................................................................................... 13
Soft-Start................................................................................................................................................................ 13
Shutdown/Auto-Restart.......................................................................................................................................... 13
Hysteretic Over-Temperature Protection.................................................................................................................. 13
Bandgap Reference................................................................................................................................................ 13
High-Voltage Bias Current Source........................................................................................................................... 13
Typical Uses of FREQUENCY (F) Pin ....................................................................................................................... 15
Typical Uses of VOLTAGE MONITOR (V) and EXTERNAL CURRENT LIMIT (X) Pins ........................................... 16
Typical Uses of MULTI-FUNCTION (M) Pin ............................................................................................................ 18
Application Examples ............................................................................................................................................... 21
A High Efficiency, 35 W, Dual Output – Universal Input Power Supply...................................................................... 21
A High Efficiency, 150 W, 250-380 VDC Input Power Supply................................................................................... 22
A High Efficiency, 20 W Continuous – 80 W Peak, Universal Input Power Supply.................................................... 23
A High Efficiency, 65 W, Universal Input Power Supply............................................................................................ 24
Key Application Considerations ............................................................................................................................... 25
TOPSwitch-HX vs.TOPSwitch-GX........................................................................................................................ . 25
TOPSwitch-HX Design Considerations ................................................................................................................... 26
TOPSwitch-HX Layout Considerations.................................................................................................................... 27
Quick Design Checklist........................................................................................................................................... 31
Design Tools........................................................................................................................................................... 31
Product Specifications and Test Conditions .......................................................................................................... 32
Typical Performance Characteristics ..................................................................................................................... 39
Package Outlines ..................................................................................................................................................... 43
Part Ordering Information ........................................................................................................................................ 47
3
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Rev. H 06/13
TOP252-262
VC
CONTROL (C)
0
ZC
1
-
SHUNT REGULATOR/
ERROR AMPLIFIER
+
5.8 V
4.8 V
-
5.8 V
+
+
SOFT START
-
INTERNAL UV
COMPARATOR
IFB
KPS(UPPER)
+
VI (LIMIT)
CURRENT
LIMIT
ADJUST
STOP LOGIC
OVP
SOFT
START
OSCILLATOR
WITH JITTER
DCMAX
DCMAX
CURRENT LIMIT
COMPARATOR
HYSTERETIC
THERMAL
SHUTDOWN
STOP
OV/
UV
+
SHUTDOWN/
AUTO-RESTART
MULTIFUNCTION (M)
LINE
SENSE
KPS(LOWER)
√ 16
ON/OFF
VBG + VT
V
DRAIN (D)
INTERNAL
SUPPLY
SOURCE (S)
CONTROLLED
TURN-ON
GATE DRIVER
DMAX
CLOCK
F REDUCTION
S
Q
LEADING
EDGE
BLANKING
R
F REDUCTION
KPS(UPPER)
KPS(LOWER)
SOFT START
IFB
PWM
IPS(UPPER)
IPS(LOWER)
OFF
SOURCE (S)
PI-4508-120307
Figure 3a. Functional Block Diagram (P and G Packages).
VC
CONTROL (C)
0
ZC
1
-
SHUNT REGULATOR/
ERROR AMPLIFIER
+
5.8 V
4.8 V
-
5.8 V
+
EXTERNAL
CURRENT
LIMIT (X)
SOFT START
-
KPS(UPPER)
VI (LIMIT)
OVP OV/
UV
DCMAX
DCMAX
CURRENT LIMIT
COMPARATOR
HYSTERETIC
THERMAL
SHUTDOWN
STOP
SOFT
START
OSCILLATOR
WITH JITTER
+
SHUTDOWN/
AUTO-RESTART
1V
LINE
SENSE
KPS(LOWER)
√ 16
ON/OFF
STOP LOGIC
V
+
VBG + VT
VOLTAGE
MONITOR (V)
+
INTERNAL UV
COMPARATOR
IFB
CURRENT
LIMIT
ADJUST
DRAIN (D)
INTERNAL
SUPPLY
SOURCE (S)
CONTROLLED
TURN-ON
GATE DRIVER
DMAX
CLOCK
S
F REDUCTION
R
Q
LEADING
EDGE
BLANKING
F REDUCTION
KPS(UPPER)
KPS(LOWER)
SOFT START
IFB
PWM
IPS(UPPER)
IPS(LOWER)
OFF
SOURCE (S)
PI-4643-082907
Figure 3b. Functional Block Diagram (M Package).
4
Rev. H 06/13
www.powerint.com
TOP252-262
VC
CONTROL (C)
0
ZC
1
-
SHUNT REGULATOR/
ERROR AMPLIFIER
+
5.8 V
4.8 V
5.8 V
+
EXTERNAL
CURRENT
LIMIT (X)
+
SOFT START
-
INTERNAL UV
COMPARATOR
IFB
CURRENT
LIMIT
ADJUST
KPS(UPPER)
VI (LIMIT)
OVP OV/
UV
SOFT
START
OSCILLATOR
WITH JITTER
DCMAX
DCMAX
CURRENT LIMIT
COMPARATOR
HYSTERETIC
THERMAL
SHUTDOWN
STOP
66k/132k
+
SHUTDOWN/
AUTO-RESTART
STOP LOGIC
LINE
SENSE
KPS(LOWER)
√ 16
ON/OFF
1V
V
+
VBG + VT
VOLTAGE
MONITOR (V)
DRAIN (D)
INTERNAL
SUPPLY
SOURCE (S)
CONTROLLED
TURN-ON
GATE DRIVER
DMAX
CLOCK
S
F REDUCTION
Q
LEADING
EDGE
BLANKING
R
FREQUENCY
(F)
F REDUCTION
KPS(UPPER)
KPS(LOWER)
SOFT START
IFB
PWM
IPS(UPPER)
IPS(LOWER)
OFF
SOURCE (S)
PI-4511-082907
Figure 3c. Functional Block Diagram (TOP254-258 YN Package and all eSIP Packages).
VC
CONTROL (C)
0
ZC
1
-
SHUNT REGULATOR/
ERROR AMPLIFIER
+
5.8 V
4.8 V
-
5.8 V
+
EXTERNAL
CURRENT
LIMIT (X)
SOFT START
-
KPS(UPPER)
VI (LIMIT)
OVP OV/
UV
DCMAX
DCMAX
CURRENT LIMIT
COMPARATOR
HYSTERETIC
THERMAL
SHUTDOWN
STOP
SOFT
START
OSCILLATOR
WITH JITTER
+
SHUTDOWN/
AUTO-RESTART
STOP LOGIC
LINE
SENSE
KPS(LOWER)
√ 16
ON/OFF
1V
V
+
VBG + VT
VOLTAGE
MONITOR (V)
+
INTERNAL UV
COMPARATOR
IFB
CURRENT
LIMIT
ADJUST
DRAIN (D)
INTERNAL
SUPPLY
SOURCE (S)
CONTROLLED
TURN-ON
GATE DRIVER
DMAX
CLOCK
S
F REDUCTION
R
Q
LEADING
EDGE
BLANKING
SOURCE (S)
F REDUCTION
KPS(UPPER)
KPS(LOWER)
SOFT START
IFB
PWM
IPS(UPPER)
IPS(LOWER)
OFF
PI-4974-122607
SIGNAL
GROUND (G)
Figure 3d. Functional Block Diagram TOP259YN, TOP260YN, TOP261YN.
5
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Rev. H 06/13
TOP252-262
DRAIN (D) Pin:
High-voltage power MOSFET DRAIN pin. The internal start-up
bias current is drawn from this pin through a switched highvoltage current source. Internal current limit sense point for
drain current.
CONTROL (C) Pin:
Error amplifier and feedback current input pin for duty cycle
control. Internal shunt regulator connection to provide internal
bias current during normal operation. It is also used as the
connection point for the supply bypass and auto-restart/
compensation capacitor.
EXTERNAL CURRENT LIMIT (X) Pin (Y, M, E and L package):
Input pin for external current limit adjustment and remote
ON/OFF. A connection to SOURCE pin disables all functions on
this pin.
Note: Y package for TOP259-261
Exposed Pad
(Hidden)
Internally
Connected to
SOURCE Pin
+
4 MΩ
RLS
DC
Input
Voltage
CONTROL
S
1
2
9
3
8
S
S
7
S
D
5
6
S
DC
Input
Voltage
1
8
S
C
2
7
S
6
S
5
S
VUV = 102.8 VDC
VOV = 451 VDC
DCMAX@100 VDC = 76%
DCMAX@375 VDC = 41%
V
CONTROL
X
For RLS = 4 MΩ
C
For RIL = 12 kΩ
ILIMIT = 61%
G
See Figure 55b for
other resistor values
(RIL) to select different
ILIMIT values.
RIL
12 kΩ
-
Figure 6. TOP259-261 Y Package Line Sense and External Current Limit.
+
VUV = IUV × RLS + VM (IM = IUV)
VOV = IOV × RLS + VM (IM = IOV)
RLS
4 MΩ
DC
Input
Voltage
D
12345 7
VXCS F D
PI-4644-091108
M
CONTROL
-
For RLS = 4 MΩ
VUV = 102.8 VDC
VOV = 451 VDC
DCMAX@100 VDC = 76%
DCMAX@375 VDC = 41%
C
S
PI-4712-120307
Figure 4. Pin Configuration (Top View).
Figure 7. P/G Package Line Sense.
6
Rev. H 06/13
www.powerint.com
PI-4983-021308
4 MΩ
D
Tab Internally
Connected to
SOURCE Pin
M
See Figure 55b for
other resistor values
(RIL) to select different
ILIMIT values.
Note: Y package for TOP254-258
10 S
P and G Package
For RIL = 12 kΩ
ILIMIT = 61%
VUV = IUV × RLS + VV (IV = IUV)
VOV = IOV × RLS + VV (IV = IOV)
S
Y Package (TO-220-7C)
V
C
RIL
12 kΩ
-
Lead Bend
Outward from Drawing
(Refer to eSIP-7F Package
Outline Drawing)
X
C
VUV = 102.8 VDC
VOV = 451 VDC
X
RLS
M Package
For RLS = 4 MΩ
DCMAX@100 VDC = 76%
DCMAX@375 VDC = 41%
V
D
+
12345 7
V XCSG D
12345 7
VXCF S D
VUV = IUV × RLS + VV (IV = IUV)
VOV = IOV × RLS + VV (IV = IOV)
Figure 5. TOP254-258 Y and All M/E/L Package Line Sense and Externally Set
Current Limit.
Tab Internally
Connected to
SOURCE Pin
12345 7
VXCF S D
L Package (eSIP-7F)
4
MULTI-FUNCTION (M) Pin (P & G packages only):
This pin combines the functions of the VOLTAGE MONITOR (V)
and EXTERNAL CURRENT LIMIT (X) pins of the Y package into
one pin. Input pin for OV, UV, line feed forward with DCMAX
Y Package (TO-220-7C)
E Package (eSIP-7C)
D
VOLTAGE MONITOR (V) Pin (Y & M package only):
Input for OV, UV, line feed forward with DCMAX reduction, output
overvoltage protection (OVP), remote ON/OFF and device reset.
A connection to the SOURCE pin disables all functions on this pin.
PI-4711-021308
Pin Functional Description
TOP252-262
Auto-Restart
+
78
For RIL = 19 kΩ
ILIMIT = 37%
DC
Input
Voltage
D
-
M
CONTROL
RIL
See Figure 55b for other
resistor values (RIL) to
select different ILIMIT values.
Slope = PWM Gain
(constant over load range)
Duty Cycle (%)
For RIL = 12 kΩ
ILIMIT = 61%
C
CONTROL
Current
S
reduction, output overvoltage protection (OVP), external current
limit adjustment, remote ON/OFF and device reset. A
connection to SOURCE pin disables all functions on this pin
and makes TOPSwitch-HX operate in simple three terminal
mode (like TOPSwitch-II).
FREQUENCY (F) Pin (TOP254-258Y, and all E and L packages):
Input pin for selecting switching frequency 132 kHz if connected
to SOURCE pin and 66 kHz if connected to CONTROL pin.
The switching frequency is internally set for fixed 66 kHz
operation in the P, G, M package and TOP259YN, TOP260YN
and TOP261YN.
SIGNAL GROUND (G) Pin (TOP259YN, TOP260YN &
TOP261YN only):
Return for C pin capacitor and X pin resistor.
SOURCE (S) Pin:
Output MOSFET source connection for high voltage power
return. Primary side control circuit common and reference point.
100
55
25
CONTROL
Current
Full Frequency Mode
132
Frequency (kHz)
Figure 8. P/G Package Externally Set Current Limit.
Drain Peak Current
To Current Limit Ratio (%)
PI-4713-021308
Low
Frequency
Mode
Variable
Frequency
Mode
66
Multi-Cycle
Modulation
Jitter
30
ICD1 IB
IC01
IC02
IC03 ICOFF CONTROL
Current
PI-4645-041107
TOPSwitch-HX Family Functional Description
Like TOPSwitch-GX, TOPSwitch-HX is an integrated switched
mode power supply chip that converts a current at the control
input to a duty cycle at the open drain output of a high voltage
power MOSFET. During normal operation the duty cycle of the
power MOSFET decreases linearly with increasing CONTROL
pin current as shown in Figure 9.
In addition to the three terminal TOPSwitch features, such as
the high voltage start-up, the cycle-by-cycle current limiting,
loop compensation circuitry, auto-restart and thermal
shutdown, the TOPSwitch-HX incorporates many additional
functions that reduce system cost, increase power supply
performance and design flexibility. A patented high voltage
CMOS technology allows both the high-voltage power MOSFET
and all the low voltage control circuitry to be cost effectively
integrated onto a single monolithic chip.
Three terminals, FREQUENCY, VOLTAGE-MONITOR, and
EXTERNAL CURRENT LIMIT (available in Y and E/L packages),
Figure 9. Control Pin Characteristics (Multi-Mode Operation).
two terminals, VOLTAGE-MONITOR and EXTERNAL CURRENT
LIMIT (available in M package) or one terminal MULTI-FUNCTION
(available in P and G package) have been used to implement
some of the new functions. These terminals can be connected
to the SOURCE pin to operate the TOPSwitch-HX in a
TOPSwitch-like three terminal mode. However, even in this three
terminal mode, the TOPSwitch-HX offers many transparent
features that do not require any external components:
1. A fully integrated 17 ms soft-start significantly reduces or
eliminates output overshoot in most applications by sweeping
both current limit and frequency from low to high to limit the
peak currents and voltages during start-up.
2. A maximum duty cycle (DCMAX) of 78% allows smaller input
storage capacitor, lower input voltage requirement and/or
higher power capability.
3. Multi-mode operation optimizes and improves the power
supply efficiency over the entire load range while maintaining
good cross regulation in multi-output supplies.
7
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Rev. H 06/13
TOP252-262
4. Switching frequency of 132 kHz reduces the transformer size
with no noticeable impact on EMI.
5. Frequency jittering reduces EMI in the full frequency mode at
high load condition.
6. Hysteretic over-temperature shutdown ensures automatic
recovery from thermal fault. Large hysteresis prevents circuit
board overheating.
7. Packages with omitted pins and lead forming provide large
drain creepage distance.
8. Reduction of the auto-restart duty cycle and frequency to
improve the protection of the power supply and load during
open loop fault, short circuit, or loss of regulation.
9. Tighter tolerances on I2f power coefficient, current limit
reduction, PWM gain and thermal shutdown threshold.
The VOLTAGE-MONITOR (V) pin is usually used for line sensing
by connecting a 4 MW resistor from this pin to the rectified DC
high voltage bus to implement line overvoltage (OV), undervoltage (UV) and dual-slope line feed-forward with DCMAX
reduction. In this mode, the value of the resistor determines the
OV/UV thresholds and the DCMAX is reduced linearly with a dual
slope to improve line ripple rejection. In addition, it also
provides another threshold to implement the latched and
hysteretic output overvoltage protection (OVP). The pin can
also be used as a remote ON/OFF using the IUV threshold.
The EXTERNAL CURRENT LIMIT (X) pin can be used to reduce
the current limit externally to a value close to the operating peak
current, by connecting the pin to SOURCE through a resistor.
This pin can also be used as a remote ON/OFF input.
For the P and G package the VOLTAGE-MONITOR and
EXTERNAL CURRENT LIMIT pin functions are combined on
one MULTI-FUNCTION (M) pin. However, some of the functions
become mutually exclusive.
The FREQUENCY (F) pin in the TOP254-258 Y and E/L packages
set the switching frequency in the full frequency PWM mode to
the default value of 132 kHz when connected to SOURCE pin. A
half frequency option of 66 kHz can be chosen by connecting
this pin to the CONTROL pin instead. Leaving this pin open is
not recommended. In the P, G and M packages and the
TOP259-261 Y packages, the frequency is set internally at
66 kHz in the full frequency PWM mode.
CONTROL (C) Pin Operation
The CONTROL pin is a low impedance node that is capable of
receiving a combined supply and feedback current. During
normal operation, a shunt regulator is used to separate the
feedback signal from the supply current. CONTROL pin voltage
VC is the supply voltage for the control circuitry including the
MOSFET gate driver. An external bypass capacitor closely
connected between the CONTROL and SOURCE pins is
required to supply the instantaneous gate drive current. The
total amount of capacitance connected to this pin also sets the
auto-restart timing as well as control loop compensation.
When rectified DC high voltage is applied to the DRAIN pin
during start-up, the MOSFET is initially off, and the CONTROL
pin capacitor is charged through a switched high voltage
current source connected internally between the DRAIN and
CONTROL pins. When the CONTROL pin voltage VC reaches
approximately 5.8 V, the control circuitry is activated and the
soft-start begins. The soft-start circuit gradually increases the
drain peak current and switching frequency from a low starting
value to the maximum drain peak current at the full frequency
over approximately 17 ms. If no external feedback/supply
current is fed into the CONTROL pin by the end of the soft-start,
the high voltage current source is turned off and the CONTROL
pin will start discharging in response to the supply current
drawn by the control circuitry. If the power supply is designed
properly, and no fault condition such as open loop or shorted
output exists, the feedback loop will close, providing external
CONTROL pin current, before the CONTROL pin voltage has
had a chance to discharge to the lower threshold voltage of
approximately 4.8 V (internal supply undervoltage lockout
threshold). When the externally fed current charges the
CONTROL pin to the shunt regulator voltage of 5.8 V, current in
excess of the consumption of the chip is shunted to SOURCE
through an NMOS current mirror as shown in Figure 3. The
output current of that NMOS current mirror controls the duty
cycle of the power MOSFET to provide closed loop regulation.
The shunt regulator has a finite low output impedance ZC that
sets the gain of the error amplifier when used in a primary
feedback configuration. The dynamic impedance ZC of the
CONTROL pin together with the external CONTROL pin
capacitance sets the dominant pole for the control loop.
When a fault condition such as an open loop or shorted output
prevents the flow of an external current into the CONTROL pin,
the capacitor on the CONTROL pin discharges towards 4.8 V.
At 4.8 V, auto-restart is activated, which turns the output
MOSFET off and puts the control circuitry in a low current
standby mode. The high-voltage current source turns on and
charges the external capacitance again. A hysteretic internal
supply undervoltage comparator keeps VC within a window of
typically 4.8 V to 5.8 V by turning the high-voltage current
source on and off as shown in Figure 11. The auto-restart
circuit has a divide-by-sixteen counter, which prevents the
output MOSFET from turning on again until sixteen discharge/
charge cycles have elapsed. This is accomplished by enabling
the output MOSFET only when the divide-by-sixteen counter
reaches the full count (S15). The counter effectively limits
TOPSwitch-HX power dissipation by reducing the auto-restart
duty cycle to typically 2%. Auto-restart mode continues until
output voltage regulation is again achieved through closure of
the feedback loop.
Oscillator and Switching Frequency
The internal oscillator linearly charges and discharges an
internal capacitance between two voltage levels to create a
triangular waveform for the timing of the pulse width modulator.
This oscillator sets the pulse width modulator/current limit latch
at the beginning of each cycle.
The nominal full switching frequency of 132 kHz was chosen to
minimize transformer size while keeping the fundamental EMI
frequency below 150 kHz. The FREQUENCY pin (available only
in TOP254-258 Y and E, L packages), when shorted to the
CONTROL pin, lowers the full switching frequency to 66 kHz
8
Rev. H 06/13
www.powerint.com
Switching
Frequency
PI-4530-041107
TOP252-262
fOSC +
fOSC 4 ms
VDRAIN
Time
Figure 10. Switching Frequency Jitter (Idealized VDRAIN Waveforms).
(half frequency), which may be preferable in some cases such
as noise sensitive video applications or a high efficiency
standby mode. Otherwise, the FREQUENCY pin should be
connected to the SOURCE pin for the default 132 kHz. In the
M, P and G packages and the TOP259-261 Y package option,
the full frequency PWM mode is set at 66 kHz, for higher
efficiency and increased output power in all applications.
To further reduce the EMI level, the switching frequency in the
full frequency PWM mode is jittered (frequency modulated) by
approximately ±2.5 kHz for 66 kHz operation or ±5 kHz for
132 kHz operation at a 250 Hz (typical) rate as shown in
Figure 10. The jitter is turned off gradually as the system is
entering the variable frequency mode with a fixed peak drain
current.
Pulse Width Modulator
The pulse width modulator implements multi-mode control by
driving the output MOSFET with a duty cycle inversely
proportional to the current into the CONTROL pin that is in
excess of the internal supply current of the chip (see Figure 9).
The feedback error signal, in the form of the excess current, is
filtered by an RC network with a typical corner frequency of
7 kHz to reduce the effect of switching noise in the chip supply
current generated by the MOSFET gate driver.
To optimize power supply efficiency, four different control
modes are implemented. At maximum load, the modulator
operates in full frequency PWM mode; as load decreases, the
modulator automatically transitions, first to variable frequency
PWM mode, then to low frequency PWM mode. At light load,
the control operation switches from PWM control to multi-cyclemodulation control, and the modulator operates in multi-cyclemodulation mode. Although different modes operate differently
to make transitions between modes smooth, the simple
relationship between duty cycle and excess CONTROL pin
current shown in Figure 9 is maintained through all three PWM
modes. Please see the following sections for the details of the
operation of each mode and the transitions between modes.
Full Frequency PWM mode: The PWM modulator enters full
frequency PWM mode when the CONTROL pin current (IC)
reaches IB. In this mode, the average switching frequency is
kept constant at fOSC (66 kHz for P, G and M packages and
TOP259-261 Y, pin selectable 132 kHz or 66 kHz for Y and E/L
packages). Duty cycle is reduced from DCMAX through the
reduction of the on-time when IC is increased beyond IB. This
operation is identical to the PWM control of all other TOPSwitch
families. TOPSwitch-HX only operates in this mode if the
cycle-by-cycle peak drain current stays above kPS(UPPER)*ILIMIT(set),
where kPS(UPPER) is 55% (typical) and ILIMIT(set) is the current limit
externally set via the X or M pin.
Variable Frequency PWM mode: When peak drain current is
lowered to kPS(UPPER)* ILIMIT(set) as a result of power supply load
reduction, the PWM modulator initiates the transition to variable
frequency PWM mode, and gradually turns off frequency jitter.
In this mode, peak drain current is held constant at kPS(UPPER)*
ILIMIT(set) while switching frequency drops from the initial full
frequency of fOSC (132 kHz or 66 kHz) towards the minimum
frequency of fMCM(MIN) (30 kHz typical). Duty cycle reduction is
accomplished by extending the off-time.
Low Frequency PWM mode: When switching frequency
reaches fMCM(MIN) (30 kHz typical), the PWM modulator starts to
transition to low frequency mode. In this mode, switching
frequency is held constant at fMCM(MIN) and duty cycle is reduced,
similar to the full frequency PWM mode, through the reduction
of the on-time. Peak drain current decreases from the initial
value of kPS(UPPER)* ILIMIT(set) towards the minimum value of
kPS(LOWER)*ILIMIT(set), where kPS(LOWER) is 25% (typical) and ILIMIT(set) is
the current limit externally set via the X or M pin.
Multi-Cycle-Modulation mode: When peak drain current is
lowered to kPS(LOWER)*ILIMIT(set), the modulator transitions to
multi-cycle-modulation mode. In this mode, at each turn-on,
the modulator enables output switching for a period of TMCM(MIN)
at the switching frequency of fMCM(MIN) (4 or 5 consecutive pulses
at 30 kHz) with the peak drain current of kPS(LOWER)*ILIMIT(set), and
stays off until the CONTROL pin current falls below IC(OFF). This
mode of operation not only keeps peak drain current low but
also minimizes harmonic frequencies between 6 kHz and
30 kHz. By avoiding transformer resonant frequency this way,
all potential transformer audible noises are greatly suppressed.
Maximum Duty Cycle
The maximum duty cycle, DCMAX, is set at a default maximum
value of 78% (typical). However, by connecting the VOLTAGEMONITOR or MULTI-FUNCTION pin (depending on the
package) to the rectified DC high voltage bus through a resistor
with appropriate value (4 MW typical), the maximum duty cycle
can be made to decrease from 78% to 40% (typical) when input
line voltage increases from 88 V to 380 V, with dual gain slopes.
Error Amplifier
The shunt regulator can also perform the function of an error
amplifier in primary side feedback applications. The shunt
regulator voltage is accurately derived from a temperaturecompensated bandgap reference. The CONTROL pin dynamic
impedance ZC sets the gain of the error amplifier. The
CONTROL pin clamps external circuit signals to the VC voltage
level. The CONTROL pin current in excess of the supply current
is separated by the shunt regulator and becomes the feedback
current Ifb for the pulse width modulator.
9
www.powerint.com
Rev. H 06/13
TOP252-262
~
~
~
~
VUV
~
~
~
~
~
~
VLINE
0V
S15 S14
S13
S12
S0
S14
S15
S13
S12
~
~
S0
S0
S15
S15
5.8 V
4.8 V
~
~
~
~
0V
S13 S12
~
~
S14
~
~
S15
VC
~
~
VDRAIN
0V
VOUT
1
2
3
~
~
~
~
~
~
0V
2
Note: S0 through S15 are the output states of the auto-restart counter
4
PI-4531-121206
Figure 11. Typical Waveforms for (1) Power Up (2) Normal Operation (3) Auto-Restart (4) Power Down.
On-Chip Current Limit with External Programmability
The cycle-by-cycle peak drain current limit circuit uses the
output MOSFET ON-resistance as a sense resistor. A current
limit comparator compares the output MOSFET on-state drain
to source voltage VDS(ON) with a threshold voltage. High drain
current causes VDS(ON) to exceed the threshold voltage and turns
the output MOSFET off until the start of the next clock cycle.
The current limit comparator threshold voltage is temperature
compensated to minimize the variation of the current limit due
to temperature related changes in RDS(ON) of the output MOSFET.
The default current limit of TOPSwitch-HX is preset internally.
However, with a resistor connected between EXTERNAL
CURRENT LIMIT (X) pin (Y, E/L and M packages) or MULTIFUNCTION (M) pin (P and G package) and SOURCE pin (for
TOP259-261 Y, the X pin is connected to the SIGNAL GROUND
(G) pin), current limit can be programmed externally to a lower
level between 30% and 100% of the default current limit. By
setting current limit low, a larger TOPSwitch-HX than necessary
for the power required can be used to take advantage of the
lower RDS(ON) for higher efficiency/smaller heat sinking
requirements. TOPSwitch-HX current limit reduction initial
tolerance through the X pin (or M pin) has been improved
significantly compare with previous TOPSwitch-GX. With a
second resistor connected between the EXTERNAL CURRENT
LIMIT (X) pin (Y, E/L and M packages) or MULTI-FUNCTION (M)
pin (P and G package) and the rectified DC high voltage bus,
the current limit is reduced with increasing line voltage, allowing
a true power limiting operation against line variation to be
implemented. When using an RCD clamp, this power limiting
technique reduces maximum clamp voltage at high line. This
allows for higher reflected voltage designs as well as reducing
clamp dissipation.
The leading edge blanking circuit inhibits the current limit
comparator for a short time after the output MOSFET is turned
on. The leading edge blanking time has been set so that, if a
power supply is designed properly, current spikes caused by
primary-side capacitances and secondary-side rectifier reverse
recovery time should not cause premature termination of the
switching pulse.
The current limit is lower for a short period after the leading
edge blanking time. This is due to dynamic characteristics of
the MOSFET. During startup and fault conditions the controller
prevents excessive drain currents by reducing the switching
frequency.
Line Undervoltage Detection (UV)
At power up, UV keeps TOPSwitch-HX off until the input line
voltage reaches the undervoltage threshold. At power down,
UV prevents auto-restart attempts after the output goes out of
regulation. This eliminates power down glitches caused by slow
discharge of the large input storage capacitor present in
applications such as standby supplies. A single resistor
connected from the VOLTAGE-MONITOR pin (Y, E/L and M
packages) or MULTI-FUNCTION pin (P and G packages) to the
rectified DC high voltage bus sets UV threshold during power
up. Once the power supply is successfully turned on, the UV
threshold is lowered to 44% of the initial UV threshold to allow
extended input voltage operating range (UV low threshold). If
the UV low threshold is reached during operation without the
power supply losing regulation, the device will turn off and stay
off until UV (high threshold) has been reached again. If the
power supply loses regulation before reaching the UV low
threshold, the device will enter auto-restart. At the end of each
auto-restart cycle (S15), the UV comparator is enabled. If the
UV high threshold is not exceeded, the MOSFET will be
disabled during the next cycle (see Figure 11). The UV feature
can be disabled independent of the OV feature.
10
Rev. H 06/13
www.powerint.com
TOP252-262
Line Overvoltage Shutdown (OV)
The same resistor used for UV also sets an overvoltage
threshold, which, once exceeded, will force TOPSwitch-HX to
stop switching instantaneously (after completion of the current
switching cycle). If this condition lasts for at least 100 ms, the
TOPSwitch-HX output will be forced into off state. Unlike with
TOPSwitch-GX, however, when the line voltage is back to
normal with a small amount of hysteresis provided on the OV
threshold to prevent noise triggering, the state machine sets to
S13 and forces TOPSwitch-HX to go through the entire autorestart sequence before attempting to switch again. The ratio
of OV and UV thresholds is preset at 4.5, as can be seen in
Figure 12. When the MOSFET is off, the rectified DC high
voltage surge capability is increased to the voltage rating of the
MOSFET (700 V), due to the absence of the reflected voltage
and leakage spikes on the drain. The OV feature can be
disabled independent of the UV feature.
In order to reduce the no-load input power of TOPSwitch-HX
designs, the V-pin (or M-pin for P Package) operates at very low
currents. This requires careful layout considerations when
designing the PCB to avoid noise coupling. Traces and
components connected to the V-pin should not be adjacent to
any traces carrying switching currents. These include the drain,
clamp network, bias winding return or power traces from other
converters. If the line sensing features are used, then the sense
resistors must be placed within 10 mm of the V-pin to minimize
the V-pin node area. The DC bus should then be routed to the
line sense resistors. Note that external capacitance must not
be connected to the V-pin as this may cause misoperation of
the V pin related functions.
Hysteretic or Latching Output Overvoltage Protection (OVP)
The detection of the hysteretic or latching output overvoltage
protection (OVP) is through the trigger of the line overvoltage
threshold. The V-pin or M-pin voltage will drop by 0.5 V, and
the controller measures the external attached impedance
immediately after this voltage drops. If IV or IM exceeds IOV(LS)
(336 mA typical) longer than 100 ms, TOPSwitch-HX will latch
into a permanent off state for the latching OVP. It only can be
reset if V V or VM goes below 1 V or VC goes below the powerup-reset threshold (VC(RESET)) and then back to normal.
If IV or IM does not exceed IOV(LS) or exceeds no longer than
100 ms, TOPSwitch-HX will initiate the line overvoltage and the
hysteretic OVP. Their behavior will be identical to the line
overvoltage shutdown (OV) that has been described in detail in
the previous section.
Voltage Monitor and External Current Limit Pin Table*
Figure Number
16
Three Terminal Operation
3
Line Undervoltage
Line Overvoltage
Line Feed-Forward (DCMAX)
17
18
19
20
3
3
3
3
3
3
3
3
3
3
3
3
Output Overvoltage Protection
21
22
23
24
25
26
27
3
3
3
3
3
3
3
3
3
3
Overload Power Limiting
External Current Limit
3
3
3
Remote ON/OFF
3
3
3
Device Reset
28
3
*This table is only a partial list of many VOLTAGE MONITOR and EXTERNAL CURRENT LIMIT Pin Configurations that are possible.
Table 2.
VOLTAGE MONITOR (V) Pin and EXTERNAL CURRENT LIMIT (X) Pin Configuration Options.
Multi-Function Pin Table*
Figure Number
29
Three Terminal Operation
3
Line Undervoltage
Line Overvoltage
Line Feed-Forward (DCMAX)
30
31
32
33
3
3
3
3
3
3
3
3
3
3
3
3
Output Overvoltage Protection
34
35
36
3
3
3
Remote ON/OFF
Device Reset
38
39
3
3
3
3
3
40
3
Overload Power Limiting
External Current Limit
37
3
*This table is only a partial list of many MULTI-FUNCTIONAL Pin Configurations that are possible.
Table 3.
MULTI-FUNCTION (M) Pin Configuration Options.
11
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Rev. H 06/13
TOP252-262
M Pin
X Pin
V Pin
IUV
IREM(N)
IOV
IOV(LS)
(Enabled)
Output
MOSFET
Switching
(Non-Latching)
(Latching)
(Disabled)
Disabled when supply
output goes out of
regulation
I
ILIMIT (Default)
Current
Limit
I
DCMAX (78%)
Maximum
Duty Cycle
I
VBG
Pin Voltage
-250
-200
-150
-100
-50
0
25
50
75
100
125
336
I
X and V Pins (Y, E, L and M Packages) and M Pin (P and G Packages) Current (µA)
Note: This figure provides idealized functional characteristics with typical performance values. Please refer to the parametric
table and typical performance characteristics sections of the data sheet for measured data. For a detailed description of
each functional pin operation refer to the Functional Description section of the data sheet.
PI-4646-071708
Figure 12. MULTI-FUNCTION (P and G package). VOLTAGE MONITOR and EXTERNAL CURRENT LIMIT (Y, E/L and M package) Pin Characteristics.
The circuit examples shown in Figures 41, 42 and 43 show a
simple method for implementing the primary sensed overvoltage protection.
During a fault condition resulting from loss of feedback, output
voltage will rapidly rise above the nominal voltage. The increase
in output voltage will also result in an increase in the voltage at
the output of the bias winding. A voltage at the output of the
bias winding that exceeds of the sum of the voltage rating of the
Zener diode connected from the bias winding output to the
V-pin (or M-pin) and V-pin (or M-pin) voltage, will cause a current
in excess of IV or IM to be injected into the V-pin
(or M-pin), which will trigger the OVP feature.
The primary sensed OVP protection circuit shown in Figures 41,
42 and 43 is triggered by a significant rise in output voltage (and
therefore bias winding voltage). If the power supply is operating
under heavy load or low input line conditions when an open
loop occurs, the output voltage may not rise significantly.
Under these conditions, a latching shutdown will not occur until
load or line conditions change. Nevertheless, the operation
provides the desired protection by preventing significant rise in
the output voltage when the line or load conditions do change.
Primary side OVP protection with the TOPSwitch-HX in a typical
application will prevent a nominal 12 V output from rising above
approximately 20 V under open loop conditions. If greater
accuracy is required, a secondary sensed OVP circuit is
recommended.
12
Rev. H 06/13
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TOP252-262
Line Feed-Forward with DCMAX Reduction
The same resistor used for UV and OV also implements line voltage
feed-forward, which minimizes output line ripple and reduces
power supply output sensitivity to line transients. Note that for the
same CONTROL pin current, higher line voltage results in smaller
operating duty cycle. As an added feature, the maximum duty
cycle DCMAX is also reduced from 78% (typical) at a voltage slightly
lower than the UV threshold to 36% (typical) at the OV threshold.
DCMAX of 36% at high line was chosen to ensure that the power
capability of the TOPSwitch-HX is not restricted by this feature
under normal operation. TOPSwitch-HX provides a better fit to the
ideal feed-forward by using two reduction slopes: -1% per mA for all
bus voltage less than 195 V (typical for 4 MW line impedance) and
-0.25% per mA for all bus voltage more than 195 V. This dual
slope line feed-forward improves the line ripple rejection
significantly compared with the TOPSwitch-GX.
Remote ON/OFF
TOPSwitch-HX can be turned on or off by controlling the
current into the VOLTAGE-MONITOR pin or out from the
EXTERNAL CURRENT LIMIT pin (Y, E/L and M packages) and
into or out from the MULTI-FUNCTION pin (P and G package,
see Figure 12). In addition, the VOLTAGE-MONITOR pin has a
1 V threshold comparator connected at its input. This voltage
threshold can also be used to perform remote ON/OFF control.
When a signal is received at the VOLTAGE-MONITOR pin or the
EXTERNAL CURRENT LIMIT pin (Y, E/L and M packages) or the
MULTI-FUNCTION pin (P and G package) to disable the output
through any of the pin functions such as OV, UV and remote
ON/OFF, TOPSwitch-HX always completes its current switching
cycle before the output is forced off.
As seen above, the remote ON/OFF feature can also be used as
a standby or power switch to turn off the TOPSwitch-HX and
keep it in a very low power consumption state for indefinitely
long periods. If the TOPSwitch-HX is held in remote off state for
long enough time to allow the CONTROL pin to discharge to the
internal supply undervoltage threshold of 4.8 V (approximately
32 ms for a 47 µF CONTROL pin capacitance), the CONTROL
pin goes into the hysteretic mode of regulation. In this mode,
the CONTROL pin goes through alternate charge and discharge
cycles between 4.8 V and 5.8 V (see CONTROL pin operation
section above) and runs entirely off the high voltage DC input,
but with very low power consumption (160 mW typical at
230 VAC with M or X pins open). When the TOPSwitch-HX is
remotely turned on after entering this mode, it will initiate a
normal start-up sequence with soft-start the next time the
CONTROL pin reaches 5.8 V. In the worst case, the delay from
remote on to start-up can be equal to the full discharge/charge
cycle time of the CONTROL pin, which is approximately 125 ms
for a 47 µF CONTROL pin capacitor. This reduced
consumption remote off mode can eliminate expensive and
unreliable in-line mechanical switches. It also allows for
microprocessor controlled turn-on and turn-off sequences that
may be required in certain applications such as inkjet and laser
printers.
Soft-Start
The 17 ms soft-start sweeps the peak drain current and
switching frequency linearly from minimum to maximum value
by operating through the low frequency PWM mode and the
variable frequency mode before entering the full frequency
mode. In addition to start-up, soft-start is also activated at each
restart attempt during auto-restart and when restarting after
being in hysteretic regulation of CONTROL pin voltage (VC), due
to remote OFF or thermal shutdown conditions. This effectively
minimizes current and voltage stresses on the output MOSFET,
the clamp circuit and the output rectifier during start-up. This
feature also helps minimize output overshoot and prevents
saturation of the transformer during start-up.
Shutdown/Auto-Restart
To minimize TOPSwitch-HX power dissipation under fault
conditions, the shutdown/auto-restart circuit turns the power
supply on and off at an auto-restart duty cycle of typically 2% if
an out of regulation condition persists. Loss of regulation
interrupts the external current into the CONTROL pin. VC
regulation changes from shunt mode to the hysteretic autorestart mode as described in CONTROL pin operation section.
When the fault condition is removed, the power supply output
becomes regulated, VC regulation returns to shunt mode, and
normal operation of the power supply resumes.
Hysteretic Over-Temperature Protection
Temperature protection is provided by a precision analog circuit
that turns the output MOSFET off when the junction temperature
exceeds the thermal shutdown temperature (142 °C typical).
When the junction temperature cools to below the lower
hysteretic temperature point, normal operation resumes, thus
providing automatic recovery. A large hysteresis of 75 °C
(typical) is provided to prevent overheating of the PC board due
to a continuous fault condition. VC is regulated in hysteretic
mode, and a 4.8 V to 5.8 V (typical) triangular waveform is
present on the CONTROL pin while in thermal shutdown.
Bandgap Reference
All critical TOPSwitch-HX internal voltages are derived from a
temperature-compensated bandgap reference. This voltage
reference is used to generate all other internal current
references, which are trimmed to accurately set the switching
frequency, MOSFET gate drive current, current limit, and the line
OV/UV/OVP thresholds. TOPSwitch-HX has improved circuitry
to maintain all of the above critical parameters within very tight
absolute and temperature tolerances.
High-Voltage Bias Current Source
This high-voltage current source biases TOPSwitch-HX from the
DRAIN pin and charges the CONTROL pin external capacitance
during start-up or hysteretic operation. Hysteretic operation
occurs during auto-restart, remote OFF and over-temperature
shutdown. In this mode of operation, the current source is
switched on and off, with an effective duty cycle of approximately 35%. This duty cycle is determined by the ratio of
CONTROL pin charge (IC) and discharge currents (ICD1 and ICD2).
This current source is turned off during normal operation when
the output MOSFET is switching. The effect of the current
source switching will be seen on the DRAIN voltage waveform
as small disturbances and is normal.
13
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Rev. H 06/13
TOP252-262
Y, E/L and M Package
CONTROL (C)
TOPSwitch-HX
200 µA
(Negative Current Sense - ON/OFF,
Current Limit Adjustment)
VBG + VT
EXTERNAL CURRENT LIMIT (X)
(Voltage Sense)
VOLTAGE MONITOR (V)
VREF
1V
(Positive Current Sense - Undervoltage,
Overvoltage, ON/OFF, Maximum Duty
Cycle Reduction, Output Overvoltage Protection)
400 µA
PI-4714-071408
Figure 13a. VOLTAGE MONITOR (V) and EXTERNAL CURRENT LIMIT (X) Pin Input Simplified Schematic.
P and G Package
CONTROL (C)
TOPSwitch-HX
200 µA
(Negative Current Sense - ON/OFF,
Current Limit Adjustment)
VBG + VT
MULTI-FUNCTION (M)
VREF
(Positive Current Sense - Undervoltage,
Overvoltage, Maximum Duty Cycle Reduction,
Output Overvoltage Protection)
400 µA
PI-4715-071408
Figure 13b. MULTI-FUNCTION (M) Pin Input Simplified Schematic.
14
Rev. H 06/13
www.powerint.com
TOP252-262
Typical Uses of FREQUENCY (F) Pin
+
+
DC
Input
Voltage
DC
Input
Voltage
D
CONTROL
S
C
D
CONTROL
S
F
F
-
-
PI-2655-071700
PI-2654-071700
Figure 14. Full Frequency Operation (132 kHz).
C
Figure 15. Half Frequency Operation (66 kHz).
15
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Rev. H 06/13
TOP252-262
Typical Uses of VOLTAGE MONITOR (V) and EXTERNAL CURRENT LIMIT (X) Pins
TOP252-258M
+
D
S
DC
Input
Voltage
S
C
X
V
S
S
S
D
VXCS F
+
TOP259-261Y
D
DC
Input
Voltage
S C
VXCSG
C
CONTROL
S
X
S D
CONTROL
C
F
S
-
X
C
CS
+
eSIP L Package
VXC FS
DC
Input
Voltage
C
D
eSIP E Package
VXC FS
PI-4984-020708
C
+
S D
X
4 MΩ
DC
Input
Voltage
V
CONTROL
C
F
-
For RLS = 4 MΩ
VUV = 102.8 VDC
VOV = 451 VDC
DCMAX@100 VDC = 76%
DCMAX@375 VDC = 41%
D
CONTROL
S
VUV = IUV × RLS + VV (IV = IUV)
VOV = IOV × RLS + VV (IV = IOV)
RLS
V
D
-
Figure 16b. Three Terminal Operation (VOLTAGE MONITOR and EXTERNAL
CURRENT LIMIT Features Disabled for TOP259-261 Y Packages.
D
S D
C
S
PI-4717-120307
PI-4956-071708
Figure 16c. Three Terminal Operation (VOLTAGE MONITOR and EXTERNAL
CURRENT LIMIT Features Disabled. FREQUENCY Pin Tied to
SOURCE or CONTROL Pin) for TOP252-262 L and E Packages.
+
VUV = IUV × RLS + VV (IV = IUV)
VOV = IOV × RLS + VV (IV = IOV)
RLS
4 MΩ
DC
Input
Voltage
D
10 kΩ
Reset
-
V
CONTROL
QR
For RLS = 4 MΩ
VUV = 102.8 VDC
VOV = 451 VDC
Sense Output Voltage
DCMAX @ 100 VDC = 76%
DCMAX @ 375 VDC = 41%
Figure 17. Line-Sensing for Undervoltage, Overvoltage and Line Feed-Forward.
VUV = IUV × RLS + VV (IV = IUV)
VOV = IOV × RLS + VV (IV = IOV)
+
RLS
DC
Input
Voltage
4 MΩ
D
C
-
For RLS = 4 MΩ
VUV = 102.8 VDC
VOV = 451 VDC
Sense Output Voltage
ROVP
VROVP
DCMAX @ 100 VDC = 76%
DCMAX @ 375 VDC = 41%
V
CONTROL
S
D
G
PI-4716-020508
Figure 16a. Three Terminal Operation (VOLTAGE MONITOR and EXTERNAL
CURRENT LIMIT Features Disabled. FREQUENCY Pin Tied to
SOURCE or CONTROL Pin) for TOP254-258 Y Packages.
D
V
D
V
D
-
TOP254-258Y
C
ROVP >3kΩ
S
PI-4756-121007
PI-4719-120307
Figure 18. Line-Sensing for Undervoltage, Overvoltage, Line Feed-Forward and
Latched Output Overvoltage Protection.
Figure 19. Line-Sensing for Undervoltage, Overvoltage, Line Feed-Forward and
Hysteretic Output Overvoltage Protection.
16
Rev. H 06/13
www.powerint.com
TOP252-262
Typical Uses of VOLTAGE MONITOR (V) and EXTERNAL CURRENT LIMIT (X) Pins (cont.)
+
VUV = RLS × IUV + VV (IV = IUV)
4 MΩ
+
For Values Shown
VUV = 103.8 VDC
RLS
DC
Input
Voltage
55 kΩ
CONTROL
1N4148
V
D
V
6.2 V
-
RLS
DC
Input
Voltage
40 kΩ
D
VOV = IOV × RLS + VV (IV = IOV)
4 MΩ
For Values Shown
VOV = 457.2 VDC
CONTROL
C
S
-
S
C
PI-4720-120307
Figure 20. Line Sensing for Undervoltage Only (Overvoltage Disabled).
+
For RIL = 12 kΩ
ILIMIT = 61%
PI-4721-120307
Figure 21. Line-Sensing for Overvoltage Only (Undervoltage Disabled). Maximum
Duty Cycle Reduced at Low Line and Further Reduction with
Increasing Line Voltage.
+
RLS
ILIMIT = 100% @ 100 VDC
ILIMIT = 53% @ 300 VDC
2.5 MΩ
TOP259-261YN would
use the G pin as the
return for RIL.
For RIL = 19 kΩ
ILIMIT = 37%
DC
Input
Voltage
See Figure 55b for other
resistor values (RIL).
D
CONTROL
S
D
CONTROL
C
TOP259-261YN would
use the G pin as the
return for RIL.
X
RIL
-
DC
Input
Voltage
S
C
X
RIL
6 kΩ
-
PI-4723-011008
PI-4722-021308
Figure 22. External Set Current Limit.
Figure 23. Current Limit Reduction with Line Voltage.
+
DC
Input
Voltage
QR can be an optocoupler
output or can be replaced by
a manual switch.
TOP259-261YN would
use the G pin as the
return for QR.
D
CONTROL
S
-
+
For RIL = 12 kΩ
DC
Input
Voltage
ILIMIT = 61%
D
CONTROL
C
X
QR
QR can be an optocoupler
output or can be replaced
by a manual switch.
S
PI-2625-011008
Figure 24. Active-on (Fail Safe) Remote ON/OFF.
C
ILIMIT = 37%
TOP259-261YN would
use the G pin as the
return for QR.
X
RIL
ON/OFF
47 KΩ
For RIL = 19 kΩ
QR
16 kΩ
ON/OFF
PI-4724-011008
Figure 25. Active-on Remote ON/OFF with Externally Set Current Limit.
17
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Rev. H 06/13
TOP252-262
+
VUV = IUV × RLS + VV (IV = IUV)
VOV = IOV × RLS + VV (IV = IoV)
TOP259-261YN would
use the G pin as the
return for QR.
CONTROL
C
DC
Input
Voltage
For RIL = 12 kΩ
ILIMIT = 61%
-
QR
16 kΩ
ON/OFF
V
D
CONTROL
TOP259-261YN would
use the G pin as the
return for RIL.
X
RIL
4 MΩ
RLS
QR can be an optocoupler
output or can be replaced
by a manual switch.
V
D
S
VUV = IUV x RLS + VV (IV = IUV)
VOV = IOV x RLS + VV (IV = IoV)
4 MΩ DCMAX@100 VDC = 76%
DCMAX@375 VDC = 41%
RLS
DC
Input
Voltage
+
S
For RLS = 4 MΩ
VUV = 102.8 VDC
VOV = 451 VDC
DCMAX @ 100 VDC = 76%
DCMAX @ 375 VDC = 41%
C
For RIL = 12 kΩ
ILIMIT = 61%
X
See Figure 55b for
other resistor values
(RIL) to select different
ILIMIT values.
RIL
12 kΩ
-
PI-4725-011008
Figure 26. Active-on Remote ON/OFF with Line-Sense and External
Current Limit.
Figure 27. Line Sensing and Externally Set Current Limit.
+
VUV = IUV × RLS + VV (IV = IUV)
VOV = IOV × RLS + VV (IV = IOV)
RLS
4 MΩ
DC
Input
Voltage
D
10 kΩ
Reset
DCMAX @ 100 VDC = 76%
DCMAX @ 375 VDC = 41%
V
CONTROL
QR
For RLS = 4 MΩ
VUV = 102.8 VDC
VOV = 451 VDC
Sense Output Voltage
C
S
-
PI-4756-121007
Figure 28. Line-Sensing for Undervoltage, Overvoltage, Line Feed-Forward and
Latched Output Overvoltage Protection with Device Reset.
Typical Uses of MULTI-FUNCTION (M) Pin
+
+
D
C
VUV = IUV × RLS + VM (IM = IUV)
VOV = IOV × RLS + VM (IM = IOV)
M
RLS
S
DC
Input
Voltage
D
M
CONTROL
-
D
S
S
S
4 MΩ
DC
Input
Voltage
S
DCMAX @ 100 VDC = 76%
DCMAX @ 375 VDC = 41%
D
C
PI-4727-061207
Figure 29. Three Terminal Operation (MULTI-FUNCTION Features Disabled).
M
CONTROL
C
S
For RLS = 4 MΩ
VUV = 102.8 VDC
VOV = 451 VDC
C
S
PI-4728-120307
Figure 30. Line Sensing for Undervoltage, Overvoltage and Line Feed-Forward.
18
Rev. H 06/13
www.powerint.com
PI-4726-021308
Typical Uses of VOLTAGE MONITOR (V) and EXTERNAL CURRENT LIMIT (X) Pins (cont.)
TOP252-262
Typical Uses of MULTI-FUNCTION (M) Pin (cont.)
+
VUV = IUV × RLS + VM (IM = IUV)
VOV = IOV × RLS + VM (IM = IOV)
RLS
4 MΩ
DC
Input
Voltage
D
M
CONTROL
For RLS = 4 MΩ
VUV = 102.8 VDC
VOV = 451 VDC
Sense Output Voltage
DCMAX @ 100 VDC = 76%
DCMAX @ 375 VDC = 41%
RLS
DC
Input
Voltage
4 MΩ
For RLS = 4 MΩ
VUV = 102.8 VDC
VOV = 451 VDC
Sense Output Voltage
ROVP
VROVP
D
DCMAX @ 100 VDC = 76%
DCMAX @ 375 VDC = 41%
M
CONTROL
C
S
-
VUV = IUV × RLS + VM (IM = IUV)
VOV = IOV × RLS + VM (IM = IOV)
+
C
ROVP >3kΩ
S
PI-4729-120307
PI-4730-120307
Figure 31. Line Sensing for Undervoltage, Overvoltage, Line Feed-Forward and
Latched Output Overvoltage Protection.
Figure 32. Line Sensing for Undervoltage, Overvoltage, Line Feed-Forward and
Hysteretic Output Overvoltage Protection.
+
VUV = RLS × IUV + VM (IM = IUV)
4 MΩ
+
VOV = IOV × RLS + VM (IM = IOV)
4 MΩ
For Values Shown
VUV = 103.8 VDC
RLS
RLS
DC
Input
Voltage
DC
Input
Voltage
40 kΩ
D
CONTROL
D
1N4148
M
CONTROL
C
S
-
55 kΩ
M
6.2 V
For Values Shown
VOV = 457.2 VDC
C
S
-
PI-4732-120307
PI-4731-120307
Figure 33. Line Sensing for Undervoltage Only (Overvoltage Disabled).
+
For RIL = 12 kΩ
ILIMIT = 61%
Figure 34. Line Sensing for Overvoltage Only (Undervoltage Disabled). Maximum
Duty Cycle Reduced at Low Line and Further Reduction with
Increasing Line Voltage.
+
RLS
For RIL = 19 kΩ
ILIMIT = 37%
DC
Input
Voltage
D
-
M
CONTROL
RIL
See Figures 55b for other
resistor values (RIL) to
select different ILIMIT values.
D
C
PI-4733-021308
Figure 35. Externally Set Current Limit (Not Normally Required – See M Pin
Operation Description).
2.5 MΩ
DC
Input
Voltage
RIL
S
ILIMIT = 100% @ 100 VDC
ILIMIT = 53% @ 300 VDC
M
CONTROL
6 kΩ
C
S
PI-4734-092107
Figure 36. Current Limit Reduction with Line Voltage (Not Normally Required –
See M Pin Operation Description).
19
www.powerint.com
Rev. H 06/13
TOP252-262
Typical Uses of MULTI-FUNCTION (M) Pin (cont.)
+
+
QR can be an optocoupler
output or can be replaced
by a manual switch.
QR can be an optocoupler
output or can be replaced
by a manual switch.
For RIL = 12 kΩ
DC
Input
Voltage
M
D
CONTROL
C
ON/OFF
47 kΩ
S
-
For RIL = 19 kΩ
16 kΩ
ILIMIT = 37%
M
D
RIL
CONTROL
QR
ON/OFF
ILIMIT = 61%
DC
Input
Voltage
C
QR
S
-
PI-2519-040501
Figure 38. Active-on Remote ON/OFF with Externally Set Current Limit
(see M Pin Operation Description).
Figure 37. Active-on (Fail Safe) Remote ON/OFF.
+
DC
Input
Voltage
+
QR can be an optocoupler
output or can be replaced
by a manual switch.
ON/OFF
RIL
VUV = IUV × RLS + VM (IM = IUV)
VOV = IOV × RLS + VM (IM = IOV)
RLS
7 kΩ
D
-
PI-4735-092107
QR
M
CONTROL
12 kΩ
RMC
24 kΩ
RMC = 2RIL
DC
Input
Voltage
D
10 kΩ
Reset
C
S
PI-4736-060607
Figure 39. Active-off Remote ON/OFF with Externally Set Current Limit
(see M Pin Operation Description).
For RLS = 4 MΩ
4 MΩ VUV = 102.8 VDC
VOV = 451 VDC
Sense Output Voltage
M
CONTROL
QR
DCMAX @ 100 VDC = 76%
DCMAX @ 375 VDC = 41%
C
S
PI-4757-120307
Figure 40. Line-Sensing for Undervoltage, Overvoltage, Line Feed-Forward and
Latched Output Overvoltage Protection with Device Reset.
20
Rev. H 06/13
www.powerint.com
TOP252-262
Application Examples
A High Efficiency, 35 W, Dual Output - Universal Input
Power Supply
The circuit in Figure 41 takes advantage of several of the
TOPSwitch-HX features to reduce system cost and power
supply size and to improve efficiency. This design delivers
35 W total continuous output power from a 90 VAC to 265 VAC
input at an ambient of 50 ºC in an open frame configuration. A
nominal efficiency of 84% at full load is achieved using
TOP258P. With a DIP-8 package, this design provides 35 W
continuous output power using only the copper area on the
circuit board underneath the part as a heat sink. The different
operating modes of the TOPSwitch-HX provide significant
improvement in the no-load, standby, and light load performance
of the power supply as compared to the previous generations of
the TOPSwitch.
Resistors R3 and R4 provide line sensing, setting line UV at
100 VDC and line OV at 450 VDC.
Diode D5, together with resistors R6, R7, capacitor C6 and TVS
VR1, forms a clamp network that limits the drain voltage of the
TOPSwitch after the integrated MOSFET turns off. TVS VR1
provides a defined maximum clamp voltage and typically only
conducts during fault conditions such as overload. This allows
the RCD clamp (R6, R7, C6 and D5) to be sized for normal
operation, thereby maximizing efficiency at light load. Should
the feedback circuit fail, the output of the power supply may
exceed regulation limits. This increased voltage at output will
also result in an increased voltage at the output of the bias
Resistors R20, R21 and R18 form a voltage divider network.
The output of this divider network is primarily dependent on the
divider circuit formed using R20 and R21 and will vary to some
extent for changes in voltage at the 15 V output due to the
connection of resistor R18 to the output of the divider network.
Resistor R19 and Zener VR3 improve cross regulation in case
only the 5 V output is loaded, which results in the 12 V output
operating at the higher end of the specification.
R11
33 Ω
VR1
P6KE200A
R7
20 Ω
1/2 W
3
11
4
9
C16
470 pF
100 V
L1
6.8 mH
L
C14
C15
680 µF L2
220 µF
25 V 3.3 µH 25 V
L3
3.3 µH
C4
100 µF
400 V
R2
1 MΩ
R10
D6
FR106 4.7 Ω
5
C10
10 µF
50 V
RT1
10 Ω
t
O
D
E
N
R13
330 Ω
S
90 - 265
VAC
C8
100 nF
50 V
R8
6.8 Ω
C9
47 µF
16 V
R19
10 Ω
R14
22 Ω
C19
1.0 µF
50 V
VR3
BZX55B8V2
8.2 V
2%
R15
1 kΩ
U2B
PS25011-H-A
TOPSwitch-HX
U1
CONTROL TOP258PN
C
M
RTN
+5 V,
2.2 A
RTN
C17
2200 µF
10 V
C11
2.2 nF
250 VAC
VR2
1N5250B
R5
20 V
5.1 kΩ
C3
220 nF
275 VAC
+12 V,
2A
C18
220 µF
10 V
D8
SB530
R4
2.0 MΩ
R1
1 MΩ
F1
3.15 A
D5
FR106
R3
2.0 MΩ
D4
1N4007
C13
680 µF
25 V
R12
33 Ω
6
D3
1N4937
C12
470 pF
100 V
D7
SB560
T1
2 EER28 7
R6
22 kΩ
2W
D2
1N4007
The output voltage is controlled using the amplifier TL431.
Diode D9, capacitor C20 and resistor R16 form the soft finish
circuit. At startup, capacitor C20 is discharged. As the output
voltage starts rising, current flows through the optocoupler diode
inside U2A, resistor R13 and diode D9 to charge capacitor C20.
This provides feedback to the circuit on the primary side. The
current in the optocoupler diode U2A gradually decreases as the
capacitor C20 becomes charged and the control amplifier IC U3
becomes operational. This ensures that the output voltage
increases gradually and settles to the final value without any
overshoot. Resistor R16 ensures that the capacitor C20 is
maintained charged at all times after startup, which effectively
isolates C20 from the feedback circuit after startup. Capacitor
C20 discharges through R16 when the supply shuts down.
C7
2.2 nF
250 VAC
C6
3.9 nF
1 kV
D1
1N4937
winding. Zener VR2 will break down and current will flow into
the “M” pin of the TOPSwitch initiating a hysteretic overvoltage
protection with automatic restart attempts. Resistor R5 will limit
the current into the M pin to < 336 mA, thus setting hysteretic
OVP. If latching OVP is desired, the value of R5 can be reduced
to 20 W.
U2A
PS25011-H-A
R16
10 kΩ
C20
10 µF
50 V
D9
1N4148
U3
TL431
2%
R17
10 kΩ
R18
196 kΩ
1%
R20
12.4 kΩ
1%
C21
220 nF
50 V
R21
10 kΩ
1%
PI-4747-020508
Figure 41. 35 W Dual Output Power Supply using TOP258PN.
21
www.powerint.com
Rev. H 06/13
TOP252-262
dissipated by VR1 and VR3, the leakage energy instead being
dissipated by R1 and R2. However, VR1 and VR3 are essential
to limit the peak drain voltage during start-up and/or overload
conditions to below the 700 V rating of the TOPSwitch-HX
MOSFET. The schematic shows an additional turn-off snubber
circuit consisting of R20, R21, R22, D5 and C18. This reduces
turn-off losses in the TOPSwitch-HX.
A High Efficiency, 150 W, 250 – 380 VDC Input
Power Supply
The circuit shown in Figure 42 delivers 150 W (19 V @ 7.7 A) at
84% efficiency using a TOP258Y from a 250 VDC to 380 VDC
input. A DC input is shown, as typically at this power level a
power factor correction stage would precede this supply,
providing the DC input. Capacitor C1 provides local decoupling,
necessary when the supply is remote from the main PFC output
capacitor.
The secondary is rectified and smoothed by D2, D3 and C5,
C6, C7 and C8. Two windings are used and rectified with
separate diodes D2 and D3 to limit diode dissipation. Four
capacitors are used to ensure their maximum ripple current
specification is not exceeded. Inductor L1 and capacitors C15
and C16 provide switching noise filtering.
The flyback topology is still usable at this power level due to the
high output voltage, keeping the secondary peak currents low
enough so that the output diode and capacitors are reasonably
sized. In this example, the TOP258YN is at the upper limit of its
power capability.
Output voltage is controlled using a TL431 reference IC and
R15, R16 and R17 to form a potential divider to sense the
output voltage. Resistor R12 and R24 together limit the
optocoupler LED current and set overall control loop DC gain.
Control loop compensation is achieved using components C12,
C13, C20 and R13. Diode D6, resistor R23 and capacitor C19
form a soft finish network. This feeds current into the control
pin prior to output regulation, preventing output voltage
overshoot and ensuring startup under low line, full load
conditions.
Resistors R3, R6 and R7 provide output power limiting,
maintaining relatively constant overload power with input voltage.
Line sensing is implemented by connecting a 4 MW resistor from
the V pin to the DC rail. Resistors R4 and R5 together form the
4 MW line sense resistor. If the DC input rail rises above
450 VDC, then TOPSwitch-HX will stop switching until the
voltage returns to normal, preventing device damage.
Due to the high primary current, a low leakage inductance
transformer is essential. Therefore, a sandwich winding with a
copper foil secondary was used. Even with this technique, the
leakage inductance energy is beyond the power capability of a
simple Zener clamp. Therefore, R1, R2 and C3 are added in
parallel to VR1 and VR3, two series TVS diodes being used to
reduce dissipation. During normal operation, very little power is
2.2 nF
250 VAC
C4
R2
R1
68 kΩ 68 kΩ
2W 2W
250 - 380
VDC
F1
4A
Sufficient heat sinking is required to keep the TOPSwitch-HX
device below 110 °C when operating under full load, low line
and maximum ambient temperature. Airflow may also be
required if a large heat sink area is not acceptable.
RT1 O
5Ωt
R6
4.7 MΩ
R4
2.0 MΩ
R7
4.7 MΩ
R5
2.0 ΜΩ
C3
4.7 nF
1 kV
D1
BYV26C
11
12
4
C18
120 pF
1 kV
X
RTN
D3
MBR20100CT
R18
22 Ω
0.5 W
R8
4.7 Ω
C17
47 pF
1 kV
R12
240 Ω
0.125 W
C9
10 µF
50 V
R23
15 kΩ
0.125 W
C11
100 nF
50 V
R16
31.6 kΩ
1%
U2
PC817A
R11
C12
1 kΩ
4.7 nF
0.125 W 50 V
U2
PC817B
C
F
C20
1.0 µF
50 V
R24
30 Ω
0.125 W
TOPSwitch-HX
U1
TOP258YN
CONTROL
S
R3
8.06 kΩ
1%
VR2
1N5258B
36 V
R19
4.7 Ω
R22
1.5 kΩ
2W
+19 V,
7.7 A
9,10
D5
1N4937
V
C15-C16
820 µF
25 V
L1
3.3 µH
T1
EI35
D
C5-C8
820 µF
25 V
D2
MBR20100CT
7
D4
1N4148
5
R20
1.5 kΩ
2W
R21
1.5 kΩ
2W
C14
47 pF
1 kV
13,14
1
VR1, VR3
P6KE100A
C1
22 µF
400 V
R14
22 Ω
0.5 W
R10
6.8 Ω
C10
47 µF
10 V
C19
10 µF
50 V
R13
56 kΩ
0.125 W
D6
1N4148
U3
TL431
2%
R17
562 Ω
1%
C13
100 nF
50 V
R15
4.75 kΩ
1%
PI-4795-092007
Figure 42. 150 W, 19 V Power Supply using TOP258YN.
22
Rev. H 06/13
www.powerint.com
TOP252-262
TOPSwitch-HX and R20, C9, R22 and VR5. Should the bias
winding output voltage across C13 rise due to output overload
or an open loop fault (opto coupler failure), then VR5 conducts
triggering the latching shutdown. To prevent false triggering
due to short duration overload, a delay is provided by R20, R22
and C9.
A High Efficiency, 20 W continuous – 80 W Peak, Universal
Input Power Supply
The circuit shown in Figure 43 takes advantage of several of
TOPSwitch-HX features to reduce system cost and power
supply size and to improve power supply efficiency while
delivering significant peak power for a short duration. This
design delivers continuous 20 W and peak 80 W at 32 V from
an 90 VAC to 264 VAC input. A nominal efficiency of 82% at full
load is achieved using TOP258MN. The M-package part has an
optimized current limit to enable design of power supplies
capable of delivering high power for a short duration.
To reset the supply following a latching shutdown, the V pin
must fall below the reset threshold. To prevent the long reset
delay associated with the input capacitor discharging, a fast AC
reset circuit is used. The AC input is rectified and filtered by
D13 and C30. While the AC supply is present, Q3 is on and Q1
is off, allowing normal device operation. However when AC is
removed, Q1 pulls down the V pin and resets the latch. The supply
will then return to normal operation when AC is again applied.
Resistor R12 sets the current limit of the part. Resistors R11
and R14 provide line feed forward information that reduces the
current limit with increasing DC bus voltage, thereby maintaining
a constant overload power level with increasing line voltage.
Resistors R1 and R2 implement the line undervoltage and
overvoltage function and also provide feed forward compensation
for reducing line frequency ripple at the output. The overvoltage
feature inhibits TOPSwitch-HX switching during a line surge
extending the high voltage withstand to 700 V without device
damage.
Transistor Q2 provides an additional lower UV threshold to the
level programmed via R1, R2 and the V pin. At low input AC
voltage, Q2 turns off, allowing the X pin to float and thereby
disabling switching.
A simple feedback circuit automatically regulates the output
voltage. Zener VR3 sets the output voltage together with the
voltage drop across series resistor R8, which sets the DC gain
of the circuit. Resistors R10 and C28 provide a phase boost to
improve loop bandwidth.
The snubber circuit comprising of VR7, R17, R25, C5 and D2
limits the maximum drain voltage and dissipates energy stored in
the leakage inductance of transformer T1. This clamp configuration
maximizes energy efficiency by preventing C5 from discharging
below the value of VR7 during the lower frequency operating
modes of TOPSwitch-HX. Resistor R25 damps high frequency
ringing for reduced EMI.
Diodes D6 and D7 are low-loss Schottky rectifiers, and
capacitor C20 is the output filter capacitor. Inductor L3 is a
common mode choke to limit radiated EMI when long output
cables are used and the output return is connected to safety
earth ground. Example applications where this occurs include
PC peripherals, such as inkjet printers.
A combined output overvoltage and over power protection
circuit is provided via the latching shutdown feature of
R19 C26
68 Ω 100 pF
0.5 W 1 kV
C8
1 nF
250 VAC
1
C20
330 µF
50 V
10
C31
22 µF
50 V
L2
L3
32 V
625 mA, 2.5 APK
3.3 µH
D8
1N4007
D9
1N4007
C3
120 µF
400 V
R1
2 MΩ
to
D11
1N4007
D10
1N4007
RT1
10 Ω
L1
5.3 mH
R23
R24
1 MΩ
1 MΩ
C1
220 nF
275 VAC
F1
3.15 A
90 - 264
VAC
R17
1 kΩ
0.5 W
C5
10 nF
1 kV
2
NC
4
T1
EF25
D
R21
1 MΩ
0.125 W
VR5
1N5250B
20 V
V
R22
2 MΩ
C30
100 nF
400 V
R26
68 kΩ
D5
LL4148
R10
56 Ω
R12
7.5 kΩ
1%
TOPSwitch-HX
U4
TOP258MN
C29
220 nF
50 V
R8
1.5 kΩ
C9
1 µF
100 V
R20
130 kΩ
U2A
PC817D
VR3
1N5255B
28 V
PI-4833-092007
X
C6
100 nF
50 V
Q2
2N3904
Q3
2N3904
C10
1 nF
250 VAC
R9
2 kΩ
C
Q1
2N3904
C13
10 µF
50 V
C28
330 nF
50 V
CONTROL
S
RTN
47 µH
D2
FR107
R14
3.6 MΩ
R15
1 kΩ
D6-D7
STPS3150
9
5
3
R3
2 MΩ
R4
2 MΩ
R25
100 Ω
R11
3.6 MΩ
R2
2 MΩ
D13
1N4007
VR7
BZY97C150
150 V
R18
39 kΩ
R6
6.8 Ω
C7
47 µF
16 V
Figure 43. 20 W Continuous, 80 W Peak, Universal Input Power Supply using TOP258MN.
23
www.powerint.com
Rev. H 06/13
TOP252-262
A High Efficiency, 65 W, Universal Input Power Supply
The circuit shown in Figure 44 delivers 65 W (19 V @ 3.42 A) at
88% efficiency using a TOP260EN operating over an input
voltage range of 90 VAC to 265 VAC.
The secondary output from the transformer is rectified by diode
D2 and filtered by capacitors C13 and C14. Ferrite Bead L3 and
capacitors C15 form a second stage filter and effectively reduce
the switching noise to the output.
Capacitors C1 and C6 and inductors L1 and L2 provide
common mode and differential mode EMI filtering. Capacitor C2
is the bulk filter capacitor that ensures low ripple DC input to the
flyback converter stage. Capacitor C4 provides decoupling for
switching currents reducing differential mode EMI.
Output voltage is controlled using a LM431 reference IC.
Resistor R19 and R20 form a potential divider to sense the
output voltage. Resistor R16 limits the optocoupler LED current
and sets the overall control loop DC gain. Control loop
compensation is achieved using C18 and R21. The components
connected to the control pin on the primary side C8, C9 and
R15 set the low frequency pole and zero to further shape the
control loop response. Capacitor C17 provides a soft finish
during startup. Optocoupler U2 is used for isolation of the
feedback signal.
In this example, the TOP260EN is used at reduced current limit
to improve efficiency.
Resistors R5, R6 and R7 provide power limiting, maintaining
relatively constant overload power with input voltage. Line
sensing is implemented by connecting a 4 MW impedance from
the V pin to the DC rail. Resistors R3 and R4 together form the
4 MW line sense resistor. If the DC input rail rises above
450 VDC, then TOPSwitch-HX will stop switching until the
voltage returns to normal, preventing device damage.
Diode D4 and capacitor C10 form the bias winding rectifier and
filter. Should the feedback loop break due to a defective
component, a rising bias winding voltage will cause the Zener
VR2 to break down and trigger the over voltage protection
which will inhibit switching.
This circuit features a high efficiency clamp network consisting
of diode D1, zener VR1, capacitor C5 together with resistors R8
and R9. The snubber clamp is used to dissipate the energy of
the leakage reactance of the transformer. At light load levels,
very little power is dissipated by VR1 improving efficiency as
compared to a conventional RCD clamp network.
An optional secondary side over voltage protection feature
which offers higher precision (as compared to sensing via the
bias winding) is implemented using VR3, R18 and U3. Excess
voltage at the output will cause current to flow through the
optocoupler U3 LED which in turn will inject current in the V-pin
through resistor R13, thereby triggering the over voltage
protection feature.
C6
2.2 nF
250 VAC
C5
VR1
2.2 nF BZY97C180
1 kV
180 V
3KBP08M
BR1
C13
C14
470 µF 470 µF
25 V
25 V
T1
4 RM10 FL1
5
R8
100 Ω
C12
1 nF R16
100 V 33 Ω
R9
1 kΩ
C2
120 µF
400 V
R1
R2
2.2 MΩ 2.2 MΩ
F1
4A
D1
DL4937
R4
2.0 MΩ
2
R6
6.8 MΩ
D5
BAV19WS
C4
100 nF
400 V
C1
330 nF
275 VAC
D3
BAV19WS
TOPSwitch-HX
U1
V TOP260EN
D
L
CONTROL
E
N
90 - 265
VAC
S
C3
470 pF
250 VAC
R7
15 kΩ
1%
L2
Ferrite Bead
X
C
F
C8
100 nF
50 V
VR3
BZX79-C22
22 V
C11
100 nF
50 V
R11
2 MΩ
D4 BAV19WS
L1
12 mH
19 V, 3.42 A
RTN
C10
VR2
R10
22 µF
1N5248B
50 V 73.2 kΩ 18 V
3
R5
5.1 MΩ
C15
47 µF
25 V
D2
MBR20100CT
FL2
6
R3
2.0 MΩ
L3
Ferrite
Bead
R12
5.1 kΩ
R16
680 Ω
C7
100 nF
25 V
U2B
LTY817C
R13
5.1 Ω
U3A
PC357A
U2A
LTY817C
R14
100 Ω
R15
6.8 Ω
C9
47 µF
16 V
R18
47 Ω
U3B
PC357A
D6
1N4148
C16
1 µF
50 V
C18
100 nF
R19
68.1 kΩ
R21
1 kΩ
C17
33 µF
35 V
U4
LM431
2%
R20
10 kΩ
PI-4998-021408
Figure 44. 65 W, 19 V Power Supply Using TOP260EN.
24
Rev. H 06/13
www.powerint.com
TOP252-262
Key Application Considerations
TOPSwitch-HX vs. TOPSwitch-GX
Table 4 compares the features and performance differences
between TOPSwitch-HX and TOPSwitch-GX. Many of the new
features eliminate the need for additional discrete components.
Other features increase the robustness of design, allowing cost
savings in the transformer and other power components.
TOPSwitch-HX vs. TOPSwitch-GX
Function
TOPSwitch-GX
TOPSwitch-HX
TOPSwitch-HX Advantages
EcoSmart
Linear frequency reduction to Multi-mode operation with
30 kHz (@ 132 kHz) for
linear frequency reduction to
duty cycles < 10%
30 kHz (@ 132 kHz) and
multi-cycle modulation
(virtually no audible noise)
• Improved efficiency over load (e.g. at 25% load
Output Overvoltage
Protection (OVP)
Not available
User programmable primary
or secondary hysteretic or
latching OVP
• Protects power supply output during open loop fault
Line Feed-Forward with Duty
Cycle Reduction
Linear reduction
Dual slope reduction with
lower, more accurate onset
point
• Improved line ripple rejection
Switching Frequency DIP-8
Package
132 kHz
66 kHz
• Increased output power for given MOSFET size due
Lowest MOSFET On
Resistance in DIP-8 Package
3.0 W (TOP246P)
I2f Trimming
Not available
point)
• Improved standby efficiency
• Improved no-load consumption
• Maximum design flexibility
• Smaller DC bus capacitor
to higher efficiency
1.8 W (TOP258P)
• Increased output power in designs without external
heat sink
-10% / +20%
• Increased output power for given core size
• Reduced over-load power
Auto-restart Duty Cycle
5.6%
2%
• Reduced delivered average output power during
open loop faults
Frequency Jitter
±4 kHz @ 132 kHz
±2 kHz @ 66 kHz
±5 kHz @ 132 kHz
±2.5 kHz @ 66 kHz
• Reduced EMI filter cost
Thermal Shutdown
130 °C to 150 °C
135 °C to 150 °C
• Increased design margin
External Current Limit
30%-100% of ILIMIT
30%-100% of ILIMIT, additional • Reduced tolerances when current limit is set
trim at 0.7 × ILIMIT
externally
Line UV Detection Threshold
50 mA (2 MW sense
impedance)
25 mA (4 MW sense
impedance)
Soft-Start
10 ms duty cycle and current 17 ms sweep through
limit ramp
multi-mode characteristic
• Reduced dissipation for lower no-load consumption
• Reduced peak current and voltage component
stress at startup
• Smooth output voltage rise
Table 4.
Comparison Between TOPSwitch-GX and TOPSwitch-HX.
25
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Rev. H 06/13
TOP252-262
TOPSwitch-HX Design Considerations
Power Table
The data sheet power table (Table 1) represents the maximum
practical continuous output power based on the following
conditions:
1. 12 V output.
2. Schottky or high efficiency output diode.
3. 135 V reflected voltage (VOR) and efficiency estimates.
4. A 100 VDC minimum for 85-265 VAC and 250 VDC minimum for 230 VAC.
5. Sufficient heat sinking to keep device temperature ≤100 °C.
6. Power levels shown in the power table for the M/P package
device assume 6.45 cm2 of 610 g/m2 copper heat sink area
in an enclosed adapter, or 19.4 cm2 in an open frame.
The provided peak power depends on the current limit for the
respective device.
TOPSwitch-HX Selection
Selecting the optimum TOPSwitch-HX depends upon required
maximum output power, efficiency, heat sinking constraints,
system requirements and cost goals. With the option to
externally reduce current limit, an Y, E/L or M package
TOPSwitch-HX may be used for lower power applications
where higher efficiency is needed or minimal heat sinking is
available.
Input Capacitor
The input capacitor must be chosen to provide the minimum
DC voltage required for the TOPSwitch-HX converter to
maintain regulation at the lowest specified input voltage and
maximum output power. Since TOPSwitch-HX has a high
DCMAX limit and an optimized dual slope line feed forward for
ripple rejection, it is possible to use a smaller input capacitor.
For TOPSwitch-HX, a capacitance of 2 mF per watt is possible
for universal input with an appropriately designed transformer.
Primary Clamp and Output Reflected Voltage VOR
A primary clamp is necessary to limit the peak TOPSwitch-HX
drain to source voltage. A Zener clamp requires few parts and
takes up little board space. For good efficiency, the clamp
Zener should be selected to be at least 1.5 times the output
reflected voltage VOR, as this keeps the leakage spike conduction
time short. When using a Zener clamp in a universal input
application, a VOR of less than 135 V is recommended to allow
for the absolute tolerances and temperature variations of the
Zener. This will ensure efficient operation of the clamp circuit
and will also keep the maximum drain voltage below the rated
breakdown voltage of the TOPSwitch-HX MOSFET. A high VOR
is required to take full advantage of the wider DCMAX of
TOPSwitch-HX. An RCD clamp provides tighter clamp voltage
tolerance than a Zener clamp and allows a VOR as high as 150
V. RCD clamp dissipation can be minimized by reducing the
external current limit as a function of input line voltage (see
Figures 23 and 36). The RCD clamp is more cost effective than
the Zener clamp but requires more careful design (see Quick
Design Checklist).
Output Diode
The output diode is selected for peak inverse voltage, output
current, and thermal conditions in the application (including heat
sinking, air circulation, etc.). The higher DCMAX of TOPSwitch-HX,
along with an appropriate transformer turns ratio, can allow the
use of a 80 V Schottky diode for higher efficiency on output
voltages as high as 15 V (see Figure 41).
Bias Winding Capacitor
Due to the low frequency operation at no-load, a 10 mF bias
winding capacitor is recommended.
Soft-Start
Generally, a power supply experiences maximum stress at
start-up before the feedback loop achieves regulation. For a
period of 17 ms, the on-chip soft-start linearly increases the
drain peak current and switching frequency from their low
starting values to their respective maximum values. This
causes the output voltage to rise in an orderly manner, allowing
time for the feedback loop to take control of the duty cycle.
This reduces the stress on the TOPSwitch-HX MOSFET, clamp
circuit and output diode(s), and helps prevent transformer
saturation during start-up. Also, soft-start limits the amount of
output voltage overshoot and, in many applications, eliminates
the need for a soft-finish capacitor.
EMI
The frequency jitter feature modulates the switching frequency
over a narrow band as a means to reduce conducted EMI peaks
associated with the harmonics of the fundamental switching
frequency. This is particularly beneficial for average detection
mode. As can be seen in Figure 45, the benefits of jitter increase
with the order of the switching harmonic due to an increase in
frequency deviation. Devices in the P, G or M package and
TOP259-261YN operate at a nominal switching frequency of
66 kHz. The FREQUENCY pin of devices in the TOP254-258 Y
and E packages offer a switching frequency option of 132 kHz or
66 kHz. In applications that require heavy snubber on the drain
node for reducing high frequency radiated noise (for example,
video noise sensitive applications such as VCRs, DVDs, monitors,
TVs, etc.), operating at 66 kHz will reduce snubber loss, resulting
in better efficiency. Also, in applications where transformer size is
not a concern, use of the 66 kHz option will provide lower EMI
and higher efficiency. Note that the second harmonic of 66 kHz
is still below 150 kHz, above which the conducted EMI
specifications get much tighter. For 10 W or below, it is possible
to use a simple inductor in place of a more costly AC input
common mode choke to meet worldwide conducted EMI limits.
Transformer Design
It is recommended that the transformer be designed for
maximum operating flux density of 3000 Gauss and a peak flux
density of 4200 Gauss at maximum current limit. The turns
ratio should be chosen for a reflected voltage (VOR) no greater
than 135 V when using a Zener clamp or 150 V (max) when
using an RCD clamp with current limit reduction with line
voltage (overload protection). For designs where operating
current is significantly lower than the default current limit, it is
recommended to use an externally set current limit close to the
operating peak current to reduce peak flux density and peak
power (see Figures 22 and 35). In most applications, the tighter
current limit tolerance, higher switching frequency and soft-start
features of TOPSwitch-HX contribute to a smaller transformer
when compared to TOPSwitch-GX.
26
Rev. H 06/13
www.powerint.com
TOP252-262
PI-2576-010600
80
70
Amplitude (dBµV)
60
50
40
30
20
-10
0
EN55022B (QP)
EN55022B (AV)
-10
-20
0.15
1
10
30
Frequency (MHz)
Figure 45a. Fixed Frequency Operation Without Jitter.
70
PI-2577-010600
80
TOPSwitch-HX (with jitter)
Amplitude (dBµV)
60
50
40
30
20
-10
0
EN55022B (QP)
EN55022B (AV)
-10
-20
0.15
1
10
30
Frequency (MHz)
Figure 45b. TOPSwitch-HX Full Range EMI Scan (132 kHz With Jitter) With
Identical Circuitry and Conditions.
Standby Consumption
Frequency reduction can significantly reduce power loss at light
or no load, especially when a Zener clamp is used. For very low
secondary power consumption, use a TL431 regulator for
feedback control. A typical TOPSwitch-HX circuit automatically
enters MCM mode at no load and the low frequency mode at
light load, which results in extremely low losses under no-load
or standby conditions.
High Power Designs
The TOPSwitch-HX family contains parts that can deliver up to
333 W. High power designs need special considerations.
Guidance for high power designs can be found in the Design
Guide for TOPSwitch-HX (AN-43).
TOPSwitch-HX Layout Considerations
The TOPSwitch-HX has multiple pins and may operate at
high power levels. The following guidelines should be
carefully followed.
Primary Side Connections
Use a single point (Kelvin) connection at the negative terminal of
the input filter capacitor for the TOPSwitch-HX SOURCE pin
and bias winding return. This improves surge capabilities by
returning surge currents from the bias winding directly to the
input filter capacitor. The CONTROL pin bypass capacitor
should be located as close as possible to the SOURCE and
CONTROL pins, and its SOURCE connection trace should not
be shared by the main MOSFET switching currents. All
SOURCE pin referenced components connected to the
MULTI-FUNCTION (M-pin), VOLTAGE MONITOR (V-pin) or
EXTERNAL CURRENT LIMIT (X-pin) pins should also be located
closely between their respective pin and SOURCE. Once again,
the SOURCE connection trace of these components should not
be shared by the main MOSFET switching currents. It is very
critical that SOURCE pin switching currents are returned to the
input capacitor negative terminal through a separate trace that
is not shared by the components connected to CONTROL,
MULTI-FUNCTION, VOLTAGE MONITOR or EXTERNAL
CURRENT LIMIT pins. This is because the SOURCE pin is also
the controller ground reference pin. Any traces to the M, V, X or
C pins should be kept as short as possible and away from the
DRAIN trace to prevent noise coupling. VOLTAGE MONITOR
resistors (R1 and R2 in Figures 46, 47, 48, R3 and R4 in
Figure 49, and R14 in Figure 50) should be located close to the
M or V pin to minimize the trace length on the M or V pin side.
Resistors connected to the M, V or X pin should be connected
as close to the bulk cap positive terminal as possible while
routing these connections away from the power switching
circuitry. In addition to the 47 μF CONTROL pin capacitor, a
high frequency bypass capacitor in parallel may be used for
better noise immunity. The feedback optocoupler output
should also be located close to the CONTROL and SOURCE
pins of TOPSwitch-HX and away from the drain and clamp
component traces.
Y Capacitor
The Y capacitor should be connected close to the secondary
output return pin(s) and the positive primary DC input pin of the
transformer.
Heat Sinking
The tab of the Y package (TO-220C) and E package (eSIP-7C)
and L package (eSIP-7F) are internally electrically tied to the
SOURCE pin. To avoid circulating currents, a heat sink
attached to the tab should not be electrically tied to any primary
ground/source nodes on the PC board. When using a P (DIP-8),
G (SMD-8) or M (DIP-10) package, a copper area underneath
the package connected to the SOURCE pins will act as an
effective heat sink. On double sided boards, topside and bottom
side areas connected with vias can be used to increase the
effective heat sinking area. In addition, sufficient copper area
should be provided at the anode and cathode leads of the
output diode(s) for heat sinking. In Figures 46 to 50 a narrow
trace is shown between the output rectifier and output filter
capacitor. This trace acts as a thermal relief between the rectifier
and filter capacitor to prevent excessive heating of the capacitor.
27
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Rev. H 06/13
TOP252-262
Isolation Barrier
Optional PCB slot for external
heatsink in contact with
SOURCE pins
C2
VR1
Input Filter
Capacitor
Y1Capacitor
C6
R4
T1
C10
R3
R9
Output
Rectifier
D1
J1
D3
Transformer
+
S
-
S
C1
D
U1
C
S
JP1
L1
M
C4
C3
R8
C5
R2
C8
J2
R8
R7
D2
R6
R1
Maximize hatched copper
areas (
) for optimum
heat sinking
JP2
R14
S
R13
HV
Output Filter
Capacitor
C7
U3
R10
C9
U2
VR2
R11
R12
-
DC
+
Out
PI-4753-070307
Figure 46. Layout Considerations for TOPSwitch-HX Using P Package.
Isolation Barrier
C2
Optional PCB slot for external
heatsink in contact with
SOURCE pins
Y1Capacitor
C6
R6
VR1
Input Filter
Capacitor
T1
R5
J1
D1
+
HV
-
D3
Transformer
R7
C3
R1
R2
R3
R4
Maximize hatched copper
areas (
) for optimum
heat sinking
R9
VR2
Output Filter
Capacitor
L1
C
X
V
C4
C9
C5
R8
R13
R14
D2
C8
U3
R11
JP1
C7
D
R10
S
S
S U1
S
S
C1
Output
Rectifier
R12
R15
JP2
U2
J2
R16
R17
- DC +
Out
PI-4752-070307
Figure 47. Layout Considerations for TOPSwitch-HX Using M Package.
28
Rev. H 06/13
www.powerint.com
TOP252-262
Isolation Barrier
C2
R4
T1
VR1
Input Filter
Capacitor
Y1Capacitor
C6
R3
R12
Output
Rectifier
C10
D1
J1
HS1
+
HV
-
S
U1
Transformer
D
Output Filter
Capacitor
D3
C7
F
L1
C
V
X
JP1
C4
R10
D2
R4
R9
R3
C5
U3
JP2
U2
VR2
R12
R13
C9
C8
R14
R2
R8
R1
R16
R7
R11
C1
J2
R15
R17
- DC +
Out
PI-4751-070307
Figure 48. Layout Considerations for TOPSwitch-HX Using TOP254-258 Y Package.
Isolation Barrier
C6
Y1Capacitor
C7
R7
T1
Input Filter
Capacitor
R6
Output Filter
Capacitor
R12
D8
HS1
S
R3
R11
Transformer
D
G
C
U5
X
V
C8
R4
C9
C17
R22
R14
D6
R8
R9
R5
VR2
C10
R21
JP2
U2
L3
R20
U4 C21
R15
C4
+
HV
-
JP1
C16
R10
J1
VR1
D5
C18
R17
R13
J2
- DC +
Out
PI-4977-021408
Figure 49. Layout Considerations for TOPSwitch-HX Using TOP259-261 Y Package.
29
www.powerint.com
Rev. H 06/13
TOP252-262
Isolation Barrier
Input Filter
Capacitor
J1
+
HV
-
C6 R7
C4
T1
R6 D5
HS1
Y1Capacitor
C7
C16
R12
D8
H52
C8
S
F
X
C
R4
D
U1
Transformer
VR1
R22
Output Filter
Capacitor
C17
R8
V
R3
Output
Rectifier
L3
R11
R5
C10
D6
R14
C9
C18
U4
R10
C19
C21
VR2
R9
J2
R17
U2
JP2
R20
R13
R15
R21
- DC +
Out
PI-4975-022108
Figure 50a. Layout Considerations for TOPSwitch-HX Using E Package and Operating at 66 kHz.
Isolation Barrier
Input Filter
Capacitor
+
HV
-
J1
C6 R7
T1
R6
HS1
Y1Capacitor
C7
C16
R12
D5
C4
D8
H52
C8
S
D
C9
F
X
C
R4
Output Filter
Capacitor
Transformer
VR1
U1
Output
Rectifier
C17
R22
L3
V
R3
R11
R5
R8
R14
C10
D6
C18
U4
R10
VR2
R9
JP2
C19
C21
R20
J2
R17
U2
R13
R15
R21
- DC +
Out
PI-4976-011410
Figure 50b. Layout Considerations for TOPSwitch-HX Using E Package and Operating at 132 kHz.
30
Rev. H 06/13
www.powerint.com
TOP252-262
Isolation Barrier
C6
Input Filter
Capacitor
C4
R6
R7
T1
Y1Capacitor
C7
R12
C16
VR1
D8
J1
+
HV
R22
X
Y
JP1
F
C S
Output
Rectifier
Transformer
D5
D
HS2
Output Filter
Capacitor
C17
C8
U1
R5
R14
R4
R3
R11
R8
L3
C9
D6
HS1
C10
C18
U4
R10
VR2
Note: Components U1, R8, C8, C9 and R22
are under heat sink HS1.
C19
C21
R9
JP2
R20
J2
R17
U2
R13
R15
R21
- DC +
Out
PI-5216-091508
Figure 50c. Layout Considerations for TOPSwitch-HX Using L Package and Operating at 132 kHz.
Quick Design Checklist
In order to reduce the no-load input power of TOPSwitch-HX
designs, the V-pin (or M-pin for P Package) operates at very
low current. This requires careful layout considerations when
designing the PCB to avoid noise coupling. Traces and
components connected to the V-pin should not be adjacent to
any traces carrying switching currents. These include the drain,
clamp network, bias winding return or power traces from other
converters. If the line sensing features are used, then the sense
resistors must be placed within 10 mm of the V-pin to minimize
the V pin node area. The DC bus should then be routed to the
line sense resistors. Note that external capacitance must not
be connected to the V-pin as this may cause misoperation of
the V pin related functions.
As with any power supply design, all TOPSwitch-HX designs
should be verified on the bench to make sure that components
specifications are not exceeded under worst-case conditions.
The following minimum set of tests is strongly recommended:
1. Maximum drain voltage – Verify that peak VDS does not
exceed 675 V at highest input voltage and maximum
overload output power. Maximum overload output power
occurs when the output is overloaded to a level just before
the power supply goes into auto-restart (loss of regulation).
2. Maximum drain current – At maximum ambient temperature,
maximum input voltage and maximum output load, verify
drain current waveforms at start-up for any signs of transformer saturation and excessive leading edge current spikes.
TOPSwitch-HX has a leading edge blanking time of 220 ns
to prevent premature termination of the ON-cycle. Verify that
the leading edge current spike is below the allowed current
limit envelope (see Figure 53) for the drain current waveform
at the end of the 220 ns blanking period.
3. Thermal check – At maximum output power, both minimum
and maximum voltage and ambient temperature; verify that
temperature specifications are not exceeded for
TOPSwitch-HX, transformer, output diodes and output
capacitors. Enough thermal margin should be allowed for
the part-to-part variation of the RDS(ON) of TOPSwitch-HX, as
specified in the data sheet. The margin required can either
be calculated from the values in the parameter table or it can
be accounted for by connecting an external resistance in
series with the DRAIN pin and attached to the same heat
sink, having a resistance value that is equal to the difference
between the measured RDS(ON) of the device under test and
the worst case maximum specification.
Design Tools
Up-to-date information on design tools can be found at the
Power Integrations website: www.powerint.com
31
www.powerint.com
Rev. H 06/13
TOP252-262
Absolute Maximum Ratings(2)
DRAIN Peak Voltage............................................ -0.3 V to 700 V
DRAIN Peak Current: TOP252.......................................... 0.68 A
DRAIN Peak Current: TOP253.......................................... 1.37 A
DRAIN Peak Current: TOP254.......................................... 2.08 A
DRAIN Peak Current: TOP255.......................................... 2.72 A
DRAIN Peak Current: TOP256.......................................... 4.08 A
DRAIN Peak Current: TOP257.......................................... 5.44 A
DRAIN Peak Current: TOP258.......................................... 6.88 A
DRAIN Peak Current: TOP259.......................................... 7.73 A
DRAIN Peak Current: TOP260.......................................... 9.00 A
DRAIN Peak Current: TOP261........................................ 11.10 A
DRAIN Peak Current: TOP262........................................ 11.10 A
CONTROL Voltage.................................................. -0.3 V to 9 V
CONTROL Current......................................................... 100 mA
VOLTAGE MONITOR Pin Voltage............................ -0.3 V to 9 V
CURRENT LIMIT Pin Voltage............................... -0.3 V to 4.5 V
MULTI-FUNCTION Pin Voltage................................ -0.3 V to 9 V
FREQUENCY Pin Voltage .......................................-0.3 V to 9 V
Storage Temperature .......................................-65 °C to 150 °C
Operating Junction Temperature.......................-40 °C to 150 °C
Lead Temperature(1).........................................................260 °C
Notes:
1. 1/16 in. from case for 5 seconds.
2. Maximum ratings specified may be applied one at a time
without causing permanent damage to the product.
Exposure to Absolute Maximum Rating conditions for
extended periods of time may affect product reliability.
Thermal Impedance
Thermal Impedance: Y Package:
(qJA) ............................................ 80 °C/W(1)
(qJC) .............................................. 2 °C/W(2)
P, G and M Packages:
(qJA) ...........................70 °C/W(3); 60 °C/W(4).
(qJC) ........................................... .11 °C/W(5)
E/L Package:
(qJA) ............................................105 °C/W(1)
(qJC) .............................................. 2 °C/W(2)
Parameter
Symbol
Notes:
1. Free standing with no heat sink.
2. Measured at the back surface of tab.
3. Soldered to 0.36 sq. in. (232 mm2), 2 oz. (610 g/m2) copper clad.
4. Soldered to 1 sq. in. (645 mm2), 2 oz. (610 g/m2) copper clad.
5. Measured on the SOURCE pin close to plastic interface.
Conditions
SOURCE = 0 V; TJ = -40 to 125 °C
See Figure 54
(Unless Otherwise Specified)
Min
Typ
Max
119
132
145
Units
Control Functions
Switching Frequency
in Full Frequency
Mode (average)
Frequency Jitter
Deviation
Frequency Jitter
Modulation Rate
Maximum Duty Cycle
Soft-Start Time
PWM Gain
fOSC
Df
kHz
59.4
66
72.6
59.4
66
72.6
fM
DCMAX
IC = ICD1
tSOFT
DCreg
PWM Gain
Temperature Drift
External Bias Current
TJ = 25 °C
FREQUENCY Pin
Connected to SOURCE
TOP252-258Y
TOP255-262L
TOP252-262E
FREQUENCY Pin
Connected to CONTROL
TOP252-258Y
TOP255-262L
TOP252-262E
TOP252-258P/G/M
TOP259-261Y
132 kHz Operation
66 kHz Operation
IV ≤ IV(DC) or IM ≤ IM(DC) or
VV, VM = 0 V
75
IV or IM = 95 mA
30
TJ = 25 °C
TOP252-255
TOP256-258
TOP259-262
TJ = 25 °C
66 kHz Operation
TOP252-255
TOP256-258
TOP259-262
kHz
250
Hz
78
83
17
-31
-27
-25
See Note A
IB
±5
±2.5
-25
-22
-20
ms
-20
-17
-15
-0.01
0.9
1.0
1.1
1.5
1.6
1.7
%
%/mA
%/mA/°C
2.1
2.2
2.4
mA
32
Rev. H 06/13
www.powerint.com
TOP252-262
Parameter
Symbol
Conditions
SOURCE = 0 V; TJ = -40 to 125 °C
(Unless Otherwise Specified)
Min
Typ
Max
TOP252-255
TOP256-258
TOP259-262
TOP252-255
TOP256-258
TOP259-262
TOP252-255
TOP256-258
TOP259-262
1.0
1.3
1.6
1.6
1.9
2.2
4.4
4.7
5.1
4.6
5.1
6.0
2.2
2.5
2.9
5.8
6.1
6.5
6.0
6.5
7.4
IC = 4 mA; TJ = 25 °C, See Figure 52
10
18
22
Units
Control Functions (cont.)
External Bias Current
IB
132 kHz Operation
66 kHz Operation
CONTROL Current at
0% Duty Cycle
IC(OFF)
132 kHz Operation
Dynamic Impedance
ZC
Dynamic Impedance
Temperature Drift
CONTROL Pin Internal
Filter Pole
mA
mA
W
0.18
%/°C
7
kHz
Upper Peak Current to
Set Current Limit Ratio
kPS(UPPER)
TJ = 25 °C
See Note B
Lower Peak Current to
Set Current Limit Ratio
kPS(LOWER)
TJ = 25 °C
See Note B
25
%
Multi-CycleModulation Switching
Frequency
fMCM(MIN)
TJ = 25 °C
30
kHz
Minimum Multi-CycleModulation On Period
TMCM(MIN)
TJ = 25 °C
135
ms
50
55
60
%
Shutdown/Auto-Restart
Control Pin
Charging Current
IC(CH)
TJ = 25 °C
Charging Current
Temperature Drift
VC = 0 V
-5.0
-3.5
-1.0
VC = 5 V
-3.0
-1.8
-0.6
See Note A
Auto-Restart
Upper Threshold
Voltage
VC(AR)U
Auto-Restart Lower
Threshold Voltage
VC(AR)L
mA
0.5
%/°C
5.8
V
4.5
4.8
0.8
1.0
5.1
V
Multi-Function (M), Voltage Monitor (V) and External Current Limit (X) Inputs
Auto-Restart
Hysteresis Voltage
VC(AR)hyst
Auto-Restart Duty
Cycle
DC(AR)
2
Auto-Restart
Frequency
f(AR)
0.5
Line Undervoltage
Threshold Current and
Hysteresis (M or V Pin)
IUV
TJ = 25 °C
Line Overvoltage
Threshold Current and
Hysteresis (M or V Pin)
IOV
TJ = 25 °C
Threshold
22
Hysteresis
Threshold
Hysteresis
25
V
4
Hz
27
14
107
112
4
%
mA
mA
117
mA
mA
33
www.powerint.com
Rev. H 06/13
TOP252-262
Parameter
Symbol
Conditions
SOURCE = 0 V; TJ = -40 to 125 °C
(Unless Otherwise Specified)
Min
Typ
Max
Units
Multi-Function (M), Voltage Monitor (V) and External Current Limit (X) Inputs
Output Overvoltage
Latching Shutdown
Threshold Current
V or M Pin Reset Voltage
IOV(LS)
TJ = 25 °C
269
336
403
mA
VV(TH) or
VM(TH)
TJ = 25 °C
0.8
1.0
1.6
V
-35
-27
-20
Remote ON/OFF
Negative Threshold
Current and Hysteresis
(M or X Pin)
IREM (N)
TJ = 25 °C
V or M Pin Short Circuit
Current
IV(SC) or
IM(SC)
TJ = 25 °C
X or M Pin Short Circuit
Current
IX(SC) or
Threshold
IM(SC)
V or M Pin Voltage
(Positive Current)
VV or VM
V or M Pin Voltage
Hysteresis (Positive
Current)
VV(hyst) or
VM(hyst)
X or M Pin Voltage
(Negative Current)
VX or VM
Maximum Duty Cycle
Reduction Onset
Threshold Current
IV(DC) or
IM(DC)
Maximum Duty Cycle
Reduction Slope
mA
Hysteresis
VX, VM = 0 V
IV or IM = IOV
VV, VM = VC
300
400
500
Normal Mode
-260
-200
-140
Auto-Restart Mode
-95
-75
-55
IV or IM = IUV
2.10
2.8
3.20
TOP252-TOP257
2.79
3.0
3.21
TOP258-TOP262
2.83
3.0
3.25
IV or IM = IOV
0.2
0.5
IX or IM = -50 mA
1.23
1.30
1.37
IX or IM = -150 mA
1.15
1.22
1.29
18.9
22.0
24.2
IC ≥ IB, TJ = 25 °C
TJ = 25 °C
5
-1.0
IV or IM ≥48 mA
-0.25
X, V or M Pin
Floating
0.6
1.0
V or M Pin Shorted to
CONTROL
1.0
1.6
ID(RMT)
VDRAIN = 150 V
Remote ON Delay
tR(ON)
From Remote ON to Drain
Turn-On
See Note B
Remote OFF
Setup Time
tR(OFF)
Minimum Time Before Drain
Turn-On to Disable Cycle
See Note B
mA
V
V
IV(DC) < IV
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