ISL6567IRZS2698

DATASHEET ISL6567 FN9243 Rev 4.00 August 9, 2011 Multipurpose Two-Phase Buck PWM Controller with Integrated MOSFET Drivers The ISL6567 two-phase synchronous buck PWM control IC provides a precision voltage regulation system for point-of-load and other high-current applications requiring an efficient and compact implementation. Multi-phase power conversion is a marked departure from single phase converter configurations employed to satisfy the increasing current demands of various electronic circuits. By distributing the power and load current, implementation of multi-phase converters utilize smaller and lower cost transistors with fewer input and output capacitors. These reductions accrue from the higher effective conversion frequency with higher frequency ripple current resulting from the phase interleaving inherent to this topology. Outstanding features of this controller IC include an internal 0.6V reference with a system regulation accuracy of 0.6%, an optional external reference input, and user-adjustable switching frequency. Precision regulation is further enhanced by the available unity-gain differential amplifier targeted at remote voltage sensing capability, while output regulation is monitored and its quality is reported via a PGOOD pin. Also included is an internal shunt regulator with optional external connection capability which extends the operational input voltage range. For applications requiring voltage tracking or sequencing, the ISL6567 offers a host of possibilities, including coincidental, ratiometric, or offset tracking, as well as sequential start-ups, user adjustable for a wide range of applications. Protection features of this controller IC include overvoltage and overcurrent protection. Overvoltage results in the converter turning the lower MOSFETs ON to clamp the rising output voltage. The ISL6567 uses cost and space-saving rDS(ON) sensing for channel current balance, dynamic voltage positioning, and overcurrent protection. Channel current balancing is automatic and accurate with the integrated current-balance control system. Overcurrent protection can be tailored to various application with no need for additional parts. Features • Integrated Two-Phase Power Conversion - Integrated 4A Drivers for High Efficiency • Shunt Regulator for Wide Input Power Conversion - 5V and Higher Bias - Up to 20V Power-Down-Conversion • Precision Channel Current Sharing - Loss-Less Current Sampling - Uses rDS(ON) • Precision Output Voltage Regulation - 0.6% System Accuracy Over-Temperature (Commercial Range) • 0.6V Internal Reference • Full Spectrum Voltage Tracking - Coincidental, Ratiometric, or Offset • Sequential Start-up Control • Adjustable Switching Frequency - 150kHz to 1.5MHz • Fast Transient Recovery Time • Unity-Gain Differential Amplifier - Increased Voltage Sensing Accuracy • Overcurrent Protection • Overvoltage Protection • Start-up into Pre-Charged Output • Small, QFN Package Footprint • Pb-Free (RoHS Compliant) Ordering Information PART NUMBER (Notes 1, 2, 3) PART MARKING TEMP. RANGE (°C) PACKAGE (Pb-Free) PKG. DWG. # ISL6567CRZ 65 67CRZ 0 to +70 24 Ld 4x4 QFN L24.4x4C ISL6567IRZ 65 67IRZ -40 to +85 24 Ld 4x4 QFN L24.4x4C NOTE: 1. Add “-T*” suffix for tape and reel. Please refer to TB347 for details on reel specifications. 2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pbfree requirements of IPC/JEDEC J STD-020. 3. For Moisture Sensitivity Level (MSL), please see device information page for ISL6567. For more information on MSL please see techbrief TB363. FN9243 Rev 4.00 August 9, 2011 Page 1 of 26 ISL6567 Pin Configuration FN9243 Rev 4.00 August 9, 2011 REFTRK SS FS PGOOD BOOT1 UGATE1 ISL6567 (24 LD QFN) TOP VIEW 24 23 22 21 20 19 RGND 1 18 PHASE1 VSEN 2 17 ISEN1 MON 3 VDIFF 4 FB 5 14 LGATE2 COMP 6 13 ISEN2 16 LGATE1 25 GND 7 8 9 10 11 12 VREG VCC EN BOOT2 UGATE2 PHASE2 15 PVCC Page 2 of 26 FS PGOOD MON VCC EN PVCC ISL6567 FN9243 Rev 4.00 August 9, 2011 Block Diagram BOOT1 20µA 300mV + 110% - 90% OSCILLATOR VCC 10µA UGATE1 POWER-ON RESET (POR) 10µA GATE CONTROL OVP PHASE1 COMP LGATE1 120% REFTRK SOFT-START AND FAULT LOGIC REFERENCE PWM1 GND CONTROL LOGIC EA FB BOOT2 PWM2 + OC - VDIFF  VCC UGATE2  RGND 20µA X1 VSEN VREG VCC CURRENT CORRECTION 20µA/1mA Page 3 of 26 ISEN2 PHASE2 LGATE2 SHUNT LINEAR REGULATOR ISEN1 GATE CONTROL SS ISL6567 Simplified Power System Diagram VIN Q1 EN CHANNEL1 Q2 PGOOD VOUT Q3 CHANNEL2 Q4 ISL6567 Typical Application VIN LIN CHFIN1 RSHUNT CBIN1 CF2 CF1 VCC VREG PVCC BOOT1 CBOOT1 REFTRK UGATE1 FS Q1 PHASE1 LOUT1 RFS ISEN1 SS RISEN1 CSS Q2 MON LGATE1 RPG BOOT2 PGOOD CBOOT2 ISL6567 R4 VOUT CHFIN2 EN UGATE2 CBIN2 CHFOUT CBOUT Q3 R5 PHASE2 ISEN2 C1 LGATE2 R2 RISEN2 C2 GND RS LOUT2 COMP Q4 RP FB VDIFF VSEN RGND R1 FN9243 Rev 4.00 August 9, 2011 Page 4 of 26 ISL6567 Absolute Maximum Ratings Thermal Information Supply Voltage, VCC, PVCC . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6.5V Shunt Regulator Voltage, VVREG . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6.5V Boot Voltage, VBOOT . . . . . . . . . . . . . . . . . . . . . PGND - 0.3V to PGND + 27V PHASE Voltage . . . . . . . . . . . . . . . . . . . . . GND - 0.3V (DC) to VBOOT + 0.3V . . . . . . . . . . . . . . . . GND - 5V ( (EQ. 5) CHANNEL-CURRENT BALANCE FIGURE 5. NORMAL SOFT-START WAVEFORMS FOR ISL6567BASED MULTI-PHASE CONVERTER Another benefit of multi-phase operation is the thermal advantage gained by distributing the dissipated heat over multiple devices and greater area. By doing this, the designer avoids the complexity of driving multiple parallel MOSFETs and the expense of using expensive heat sinks and exotic magnetic materials. Whenever the ISL6567’s power-on reset falling threshold is tripped, or it is disabled via the EN pin, the SS capacitor is quickly discharged via an internal pull-down device (represented as the 1mA, typical, current source). All things being equal, in order to fully realize the thermal advantage, it is important that each channel in a multi-phase converter be controlled to deliver about the same current at any load level. Intersil’s ISL6567 ensure current balance by comparing each channel’s current to the average current delivered by both channels and making appropriate adjustments to each channel’s pulse width based on the resultant error. The error signal modifies the pulse width to correct any unbalance and force the error toward zero. Should OC protection be tripped while the ISL6567 is operating in internal-reference mode and the SS pin not be allowed to fully discharge the SS capacitor, the ISL6567 cannot continue the normal SS cycling. Conversely, should a channel-to-channel imbalance be desired, such imbalance can be created by adjusting the individual channel’s RISEN resistor. Asymmetrical layouts, where one phase of the converter is naturally carrying more current than the other, or where one of the two phases is subject to a more stringent thermal environment limiting its current-carrying capability, are instances where this adjustment is particularly useful, helping to cancel out the design-intrinsic thermal or current imbalances. As the SS pin’s positive excursion is internally clamped to about 3.5V, insure that any external pull-up device does not force more than 3mA into this pin. OVERCURRENT PROTECTION The individual channel currents, as sensed via the PHASE pins and scaled via the ISEN resistors, as well as their combined average are continuously monitored and compared with an internal overcurrent (OC) reference. If the combined channel current average exceeds the reference current, the overcurrent comparator triggers an overcurrent event. Similarly, an OC event is also triggered if either channel’s current exceeds the OC reference for 7 consecutive switching cycles. SOFT-START The soft-start function allows the converter to bring up the output voltage in a controlled fashion, resulting in a linear ramp-up. As soon as the controller is fully enabled for operation, the SS pin starts to output a small current which charges the external capacitor, CSS, connected to this pin. An internal reference clamp controlled by the potential at the SS pin releases the reference to the input of the error amplifier with a 1:1 correspondence for SS potential exceeding 0.7V FN9243 Rev 4.00 August 9, 2011 Page 11 of 26 ISL6567 SETTING THE OUTPUT VOLTAGE OUTPUT CURRENT EN GND> SS The ISL6567 uses a precision internal reference voltage to set the output voltage. Based on this internal reference, the output voltage can thus be set from 0.6V up to a level determined by the input voltage, the maximum duty cycle, and the conversion efficiency of the circuit; Equation 7 estimates this maximum amplitude the output voltage can be regulated to. Obviously, insure that the input voltage and all the voltages sampled by the ISL6567 do not exceed the controller’s absolute maximum limits, or any other limits specified in this document. ISL6567 EXTERNAL CIRCUIT VREF GND> OUTPUT VOLTAGE FIGURE 6. OVERCURRENT BEHAVIOR WHILE IN INTERNAL REFERENCE MODE As a result of an OC event, output drives turn off both upper and lower MOSFETs, and the SS capacitor is discharged via a 20µA current source. The behavior following this standard response varies depending whether the controller is operating in internal (using internal reference; MON > 3.5V) or external reference mode (using external reference; MON < 3.5V). As shown in Figure 6, the soft-start capacitor discharge prompted by the OC event is followed by two SS cycles, during which the ISL6567 stays off. Following the dormant SS cycles, the controller attempts to re-establish the output. Should the OC condition been removed, the output voltage is ramped up and operation resumes as normal. Should the OC condition still be present and result in another OC event, the entire behavior repeats until the OC condition is removed or the IC is disabled. Figure 7 details the OC behavior while in external reference mode. Following the OC event, the output drives are turned off, the ISL6567 latches off, and the SS capacitor is discharged to ground. Resetting the OC latch involves removal of bias power or cycling of the EN pin (pictured in Figure 7). Should the OC event been removed, the controller initiates a new SS cycle and restores the output voltage. OUTPUT CURRENT EN GND> GND> + FB ERROR AMPLIFIER R1 VDIFF X1 DIFFERENTIAL AMPLIFIER + - RGND RP + To VOUT - FIGURE 8. SETTING THE OUTPUT VOLTAGE AT THE INPUT OF THE DIFFERENTIAL AMPLIFIER (EQ. 7) V OUTMAX = d MAX  V IN  Efficiency The output voltage can be set via a simple resistor divider, as shown in Figure 8. It is recommended this resistor divider is connected at the input of the differential amplifier (as this amplifier is powered from the IC’s 5V bias and has limited input range). To avoid degradation of DC regulation tolerance due to the differential amplifier’s input resistance, a size requirement is placed on the combined value of RP and RS. Consider R to be the parallel combination of these two resistors, and use a value of 2k or less for R; use the following equations to determine the value of RP and RS, based on the desired output voltage, the reference voltage, and the chosen value of R. V OUT R P = R  -------------------------------V OUT – V REF SS RS VSEN V OUT R S = R  ------------V REF (EQ. 8) GND> OUTPUT VOLTAGE FIGURE 7. OVERCURRENT BEHAVIOR WHILE IN EXTERNAL REFERENCE MODE FN9243 Rev 4.00 August 9, 2011 Page 12 of 26 ISL6567 ISL6567 minimal impact on the output voltage setting, follow the guidelines presented in “Setting the Output Voltage” on page 12. EXTERNAL CIRCUIT VREF + ERROR AMPLIFIER EXTERNAL REFERENCE OPERATION RS(R1) FB + RP To VOUT VDIFF + VSEN - RGND X1 DIFFERENTIAL AMPLIFIER FIGURE 9. SETTING THE OUTPUT VOLTAGE AT THE FB PIN The differential amplifier can be used even if remote output sensing is not desired or not feasible, simply connect RGND to the local ground and connect VSEN to the output voltage being monitored. Should one desire to bypass the differential amplifier, the circuit in Figure 9 is recommended as the proper implementation. Since its output is monitored for OVP and PGOOD purposes, the differential amplifier needs to be connected to the feedback circuit at all times, hence its input connections to FB and local ground. However, its output, VDIFF, can be left open. The resistor divider setting the output voltage is calculated in a manner identical to that already revealed. ISL6567 ZIN ~2µA DIFFERENTIAL AMPLIFIER VSEN + RGND ZIN > 240k OVER PROCESS, TEMPERATURE, AND 0 < VSEN-RGND < 2.5V FIGURE 10. SETTING THE OUTPUT VOLTAGE AT THE INPUT OF THE DIFFERENTIAL AMPLIFIER DIFFERENTIAL AMPLIFIER’S UNITY GAIN NETWORK The differential amplifier on the ISL6567 utilizes a typical resistive network along with active compensating circuitry to set its unity gain. This resistive network can affect the DC regulation setpoint in proportion to its relative magnitude compared to the external output voltage setting resistor divider. Figure 10 details the internal resistive network. For FN9243 Rev 4.00 August 9, 2011 ISL6567 IH (10µA) EXTERNAL CIRCUIT VDIFF VCC MON COMPARATOR RS MON + - RP VREF (300mV) + - EXTERNAL CIRCUIT VDIFF - The ISL6567 is capable of accepting an external voltage and using it as a reference for its output regulation. To enable this mode of operation, the MON pin potential has to be below 3.5V and the reference voltage has to be connected to the REFTRK pin. The internal or external reference mode of operation is latched in every time the POR is released or the ISL6567 is enabled. The highest magnitude external reference fed to the REFTRK pin that the ISL6567 can follow is limited to 2.3V. The ISL6567 utilizes a small initial negative offset (typically about 50mV) in the voltage loop at the beginning of it soft-start, to counteract any positive offsets that may have undesirable effects. As this initial offset is phased out as the reference is ramped up to around 200mV, in order to avoid an error in the output regulation level, it is recommended the external reference has an amplitude (final, DC level) exceeding 300mV. FIGURE 11. SETTING THE MONITORING THRESHOLD AND HYSTERESIS While in external reference (ER) mode, the threshold sensitive MON pin can be used to control when the ISL6567 starts to monitor the output for PGOOD and OVP protection purposes. As shown in Figure 11, connect the MON pin to the voltage to be monitored via a resistor divider. An internal current source helps set a user-adjustable monitor threshold hysteresis. Choose resistor values according to desired hysteresis voltage, VH, and desired rising threshold, V T. Make note that, in these equations, VREF refers to the reference of the MON comparator (300mV). VH R S = -----IH R S  V REF R P = ------------------------V T – V REF (EQ. 9) Intersil recommends the MON threshold is set to be tripped when the external reference voltage reaches at least 90% of the final DC value. Since PGOOD and OVP monitoring are relative to the external reference magnitude, it is important to understand that the PGOOD and OVP thresholds will move in proportion to the moving reference (externally soft-started reference). Thus, the absolute thresholds of PGOOD and OVP. Page 13 of 26 ISL6567 ISL6567 POWER-GOOD OPERATION VOLTAGE TRACKING AND SEQUENCING The open-collector PGOOD output reports on the quality of the regulated output voltage. Once the ISL6567 is enabled and the MON pin is above its threshold, PGOOD goes open circuit when the output enters the power-good window (see the “Electrical Specifications” table starting on page 5, and stays open for as long as the output remains within the specified window. The PGOOD is immediately pulled low if the ISL6567 is disabled by removal of bias, toggling of the EN pin, or upon encountering of an overcurrent event; PGOOD is allowed to report the status of the output as soon as operation is resumed following any of these events. By making creative use of the reference clamps at the SS and REFTRK pins, and/or the available external reference input, as well as the functionality of the EN pin, the ISL6567 can accommodate the full spectrum of tracking and sequencing options. The following figures offer some implementation suggestions for a few typical situations. VTARGET VOUT VTARGET ISL6567 VOUT REFTRK + To VTARGET R S RP ISL6567 REFTRK + To VTARGET R S RP + - FB R1 VDIFF E/A + FB VSEN + R1 X1 VDIFF - RP V REF ------------ = ------------------V OUT RP + RS VSEN + X1 - RGND RP RGND RS - + To VOUT FIGURE 13. COINCIDENTAL VOLTAGE TRACKING - + To VOUT FIGURE 12. RATIOMETRIC VOLTAGE TRACKING The MON pin is used in external-reference configurations, where the reference is controlled by a circuit external to the ISL6567. As such, the ISL6567 has no way of ‘knowing’ when the external reference has stabilized to its full value, or is within a certain percentage of its final value. Thus, the MON pin’s functionality can be used to indicate when a desired threshold has been reached (either by monitoring the reference itself, or the output voltage controlled by the ISL6567). By default, when operating in external-reference mode and desiring PGOOD monitoring as shown in this datasheet, it is recommended the MON is set to trip its threshold when the output voltage (or reference) reaches 92% of the final set value, choosing the resistor divider as to achieve a 2% hysteresis. When operating in internal-reference mode, the value of the reference is known to the ISL6567, so the MON pin function can be bypassed by tying it to VCC potential. FN9243 Rev 4.00 August 9, 2011 E/A EXT CIRCUIT VREF - RP V OUT ---------------------- = ------------------V TARGET RP + RS EXT CIRCUIT VREF Simple ratiometric external voltage tracking, such as that required by the termination voltage regulator for double data rate (DDR) memory can be implemented by feeding a reference voltage equal to 0.5 of the memory core voltage (VDDQ) to the reference input of the ISL6567, as shown in Figure 12. The resistor divider at the REFTRK pin sets the VOUT level. Select a suitable SS capacitor, such that the SS clamp does not interfere with the desired ramp-up time or slope of VOUT. Coincidental tracking using the internal reference results in a behavior similar to that presented in Figure 13. The resistor divider at the input of the differential amplifier sets the output voltage, VOUT, to the desired regulation level. The same resistor divider used at the REFTRK pin divides down the voltage to be tracked, effectively scaling it to the magnitude of the internal reference. As a result, the output voltage ramps up at the same rate as the target voltage, its ramp-up leveling off at the programmed regulation level established by the RS/RP resistor divider. Page 14 of 26 ISL6567 threshold. For as long as the ISL6567 is biased, OVP has the highest priority, bypassing all other control mechanisms and acting directly onto the lower MOSFETs, as described. Disabling the IC via the EN pin does not turn off OVP protection. VTARGET VOUT VOFS START-UP INTO A PRE-CHARGED OUTPUT ISL6567 REFTRK EXT CIRCUIT VREF - + + VOFS The ISL6567 also has the ability to start up into a pre-charged output, without causing any unnecessary disturbance. The FB pin is monitored during soft-start, and should it be higher than the equivalent internal ramping reference voltage, the output drives hold both MOSFETs off. Once the internal ramping reference exceeds the FB pin potential, the output drives are enabled, allowing the output to ramp from the pre-charged level to the final level dictated by the circuit setting. To VTARGET RS RP E/A + - FB R1 VOFS + - + + OUTPUT PRE-CHARGED: ABOVE INTERNAL REFERENCE ABOVE EXTERNAL REFERENCE BELOW REFERENCE VDIFF - VSEN + X1 - VOUT (1.0V/DIV) RP RGND RP V REF ------------ = ------------------V OUT RP + RS RS - + To VOUT SS (1V/DIV) GND> OUTPUT INITIALLY DISCHARGED EN (5V/DIV) FIGURE 14. OFFSET VOLTAGE TRACKING GND> Offset tracking can be accomplished via a circuit similar to that used for coincidental tracking (see Figure 14). The desired offset can be implemented via a voltage source inserted in line with the resistor divider present at the REFTRK pin. Since most offset tracking requirements are subject to fairly broad tolerances, simple voltage drop sources can be used. Figure 14 exemplifies the use of various counts of forward-biased diodes or that of a Schottky, although other options are available. Sequential start-up control is easily implemented via the EN pin, using either a logic control signal or the ISL6567’s own EN threshold as a power-good detector for the tracked, or sequence-triggering, voltage. See Figure 15 for details of control using the EN pin. OVERVOLTAGE PROTECTION Although the normal feedback loop operation naturally counters overvoltage (OV) events the ISL6567 benefits from a secondary, fixed threshold overvoltage protection. Should the output voltage exceed 120% of the reference, the lower MOSFETs are turned on. Once turned on, the lower MOSFETs are only turned off when the sensed output voltage drops below the 110% falling threshold of the OC comparator. The OVP behavior repeats for as long as the ISL6567 is biased, should the sensed output voltage rise back above the designated threshold. The occurrence of an OVP event does not latch the controller; should the phenomenon be transitory, the controller resumes normal operation following such an event. FIGURE 15. SOFT-START WAVEFORMS INTO A PRE-CHARGED OUTPUT CAPACITOR BANK As shown in Figure 15, while operating in internal reference mode, should the output be pre-charged to a level exceeding the circuit’s output voltage setting, the output drives are enabled at the conclusion of the internal reference ramp, leading to an abrupt correction in the output voltage down to the set level. When operating in external reference mode, should the output voltage be pre-charged above the regulation level driven by the external reference, the output drives are fully enabled when the SS pin levels out at the top of its range. CONTROL OF ISL6567 OPERATION The internal power-on reset circuit (POR) prevents the ISL6567 from starting before the bias voltage at VCC and PVCC reach the rising POR thresholds, as defined in the “Electrical Specifications” table starting on page 5. The POR levels are sufficiently high to guarantee that all parts of the ISL6567 can perform their functions properly once bias is applied to the part. While bias is below the rising POR thresholds, the controlled MOSFETs are kept in an off state. When operating in external-reference mode, the OVP monitoring is enabled when the MON pin exceeds its rising FN9243 Rev 4.00 August 9, 2011 Page 15 of 26 ISL6567 ISL6567 EXTERNAL CIRCUIT ISL6567 +5V POR CIRCUIT EXTERNAL CIRCUIT PVCC POR CIRCUIT VIN VCC VIN VCC VREF (0.61V) RBIAS RS E/A - + EN COMP Q1 EN OFF RP + VREF ON IH (20µA) VREG SHUNT REG VCC FIGURE 16. START-UP COORDINATION USING THE EN PIN A secondary disablement feature is available via the thresholdsensitive enable input, the EN. This optional feature prevents the ISL6567 from operating until a certain chosen voltage rail is available and above some selectable threshold. One example would be the input voltage: it may be desirable the ISL6567-based converter does not start up until the input voltage is sufficiently high. The schematic in Figure 16 demonstrates coordination of the ISL6567 start-up with such a rail. The internal current source, IH, provides the means to a user-adjustable hysteresis. The resistor value selection process follows the same equations as those presented in “External Reference Operation” on page 13. Additionally, an open-drain or open-collector device (Q1) can be used to wire-AND a second (or multiple) control signal. To defeat the threshold-sensitive enable, connect EN to VCC directly or via a pull-up resistor. In summary, for the ISL6567 to operate, VCC and PVCC must be greater than their respective POR thresholds and the voltage at EN must be greater than the comparator’s reference (see typical threshold in the ”Electrical Specifications” table starting on page 5 ). Once these conditions are met, the controller immediately initiates a soft-start sequence. General Application Design Guide This design guide is intended to provide a high-level explanation of the steps necessary to create a multi-phase power converter. It is assumed that the reader is familiar with many of the basic skills and techniques referenced in the following. In addition to this guide, Intersil provides complete reference designs that include schematics, bills of materials, and example board layouts for typical applications. FN9243 Rev 4.00 August 9, 2011 FIGURE 17. INTERNAL SHUNT REGULATOR USE WITH EXTERNAL RESISTOR (PASSIVE CONFIGURATION) BIAS SUPPLY CONSIDERATIONS The ISL6567 features an on-board shunt regulator capable of sinking up to 100mA (minimally). This integrated regulator can be used to produce the necessary bias voltage for the controller and the MOSFETs. The integrated regulator can be utilized directly, via a properly sized resistor, as shown in Figure 17, or via an external NPN transistor and additional resistors when either the current needed or the power being dissipated becomes too large to be handled inside the ISL6567 in the given operating environment. A first step in determining the feasibility and selecting the proper bias regulator configuration consists in determining the maximum bias current required by the circuit. While the bias current required by the ISL6567 is listed in the “Electrical Specifications” table starting on page 5, the bias current required by the controlled MOSFETs needs be calculated. Equation 10 helps determine this bias current function of the sum of the gate charge of all the controlled MOSFETs at 5V VGS, QGTOTAL, and the switching frequency, FSW: I B  Q GTOTAL  F SW (EQ. 10) I BIAS = I VCC + I B Total required bias current, IBIAS, sums up the ISL6567’s bias current, IVCC, to that required by the MOSFETs, IB. Page 16 of 26 ISL6567 70 7 VIN = 1V VIN = 2V VIN = 3V VIN = 4V 6 60 50 40 30 VIN = 5V VIN = 6V VIN = 7V VIN = 8V 20 10 0 6 7 8 9 VINmin (V) 10 (EQ. 11) VCC To exemplify the use, for an input voltage ranging from 10V to 14V, find the intersection of the VIN = 4V curve with the VINmin = 10V mark and project the result on the Y axis to find the maximum bias current obtainable (approximately 56% of the maximum current obtainable via the integrated shunt regulator, IVREG_MAX). Once the maximum obtainable bias current, IBIAS_MAX, is determined, and providing it is greater than the bias current, IBIAS, required by the circuit, RSHUNT can be determined based on the lowest input voltage, VINMIN: V INMIN – V VCC R BIAS = -------------------------------------I BIAS 4 3 1 0 6 The maximum bias current, IBIAS, that can be obtained via the internal shunt regulator and a simple external resistor is characterized in Figure 18 and can also be determined using Equation 11. INMAX 5 2 11 FIGURE 18. NORMALIZED MAXIMUM BIAS CURRENT OBTAINABLE IN PASSIVE CONFIGURATION vs INPUT VOLTAGE RANGE CHARACTERISTIC; VVCC = 5V V INMIN – V VCC I BIASMAX = I VREGMAX  --------------------------------------V –V RBIASmin (k•mA) IBIAS_MAX (% of IVREG_MAX) 80 (EQ. 12) 7 8 9 VINmin (V) 11 10 FIGURE 19. NORMALIZED RESISTOR VALUE IN PASSIVE CONFIGURATION; VVCC = 5V Figure 20 details the normalized maximum power dissipation RBIAS will be subject to in the given application. To use the graph provided, find the power dissipation level corresponding to the minimum input voltage and the input voltage range and multiply it by the maximum desired bias current to obtain the maximum power RBIAS will dissipate. 150 VIN = 1V VIN = 2V VIN = 3V VIN = 4V VIN = 5V VIN = 6V VIN = 7V VIN = 8V 135 120 PMAX_RBIAS (mW/mA) 90 105 90 75 60 45 30 15 Figure 19 helps with a quick resistor selection based on the previous guidelines presented. Divide the value thus obtained by the maximum desired bias current, IBIAS, to obtain the actual resistor value to be used. 0 6 7 8 9 VINmin (V) 10 11 FIGURE 20. NORMALIZED RESISTOR POWER DISSIPATION (AS SELECTED IN FIGURE 17) vs MINIMUM INPUT VOLTAGE; VVCC = 5V Alternately, the maximum power dissipation inside RBIAS can be calculated using Equation 13. P MAXRBIAS =  V INMAX – V VCC   I BIAS (EQ. 13) Maximum power dissipation in the bias resistor will take place at the upper end of the input voltage range. Select a resistor with a power dissipation rating above the calculated value and pay attention to design aspects related to the power dissipation level of this component. Although Figures 18 FN9243 Rev 4.00 August 9, 2011 Page 17 of 26 ISL6567 through 20 assume a VCC voltage of 5V, the design aid curves can be translated to a different VCC voltage by translating them in the amount of the voltage differential, to the left for a lower VCC voltage, or to the right for a higher VCC voltage. Should the simple series bias resistor configuration fall short of providing the necessary bias current, the internal shunt regulator can be used in conjunction with an external BJT transistor to increase the shunt regulator current. Figure 21 details such an implementation utilizing a PNP transistor. Selection of R1 can be based on the graphs provided for the passive regulator configuration. Maximum power dissipation inside Q1 will take place when maximum voltage is applied to the circuit and the ISL6567 is disabled; determine IVREGMAX by reverse-use of the graph in Figure 18 and use the obtained number to calculate Q1 power dissipation. ISL6567 VIN POR CIRCUIT R1 VCC Q1 - R2 (optional) + VREF VCC E/A VIN Q1 POR CIRCUIT R1 (optional) + VREF VREG R2 SHUNT REGULATOR FIGURE 22. INTERNAL SHUNT REGULATOR USE WITH EXTERNAL NPN TRANSISTOR (ACTIVE CONFIGURATION) FREQUENCY COMPENSATION The ISL6567 multi-phase converter behaves in a similar manner to a voltage-mode controller. This section highlights the design consideration for a voltage-mode controller requiring external compensation. To address a broad range of applications, a type-3 feedback network is recommended (see Figure 23). VREG C2 R2 SHUNT REGULATOR C1 COMP FIGURE 21. INTERNAL SHUNT REGULATOR USE WITH EXTERNAL PNP TRANSISTOR (ACTIVE CONFIGURATION) FB C3 R1 An NPN transistor can also be used to increase the maximum available bias current, as shown in Figure 22. Used as a series pass element, Q1 will dissipate the most power when the circuit is enabled and operational, and the input voltage, VIN, is at its highest level. With the series pass element configuration shown in Figure 22, the difference between the input and the regulation level at the VCC pin has to be higher than the lowest acceptable VCE of Q1 (may choose to run Q1 into saturation, but must consider the reduced gain). Thus, R2 has to be chosen such that it will provide appropriate base current at lowest VCE of Q1. Next, ensure the ISL6567’s IVREGMAX is not exceeded when the input voltage swings to its highest extreme (assume base current goes to 0 when the IC is disabled). R1 is an optional circuit element: it can be added to offset some of the power dissipation in Q1, but it also reduces the available VCE for Q1. If utilizing such a series resistor, check that it does not impede on the proper operation at the lowest input voltage and choose a power rating corresponding to the highest bias current that the ISL6567 may require to drive the switching MOSFETs. FN9243 Rev 4.00 August 9, 2011 EXTERNAL CIRCUIT PVCC EXTERNAL CIRCUIT PVCC E/A ISL6567 ISL6567 R3 VDIFF (VOUT) FIGURE 23. COMPENSATION CONFIGURATION FOR ISL6567 CIRCUIT Figure 24 highlights the voltage-mode control loop for a synchronous-rectified buck converter, applicable, with a small number of adjustments, to the multi-phase ISL6567 circuit. The output voltage (VOUT) is regulated to the reference voltage, VREF, level. The error amplifier output (COMP pin voltage) is compared with the oscillator (OSC) modified saw-tooth wave to provide a pulse-width modulated wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter (L and C). The output filter capacitor bank’s equivalent series resistance is represented by the series resistor E. Page 18 of 26 ISL6567 pin, RO in Figure 24, the design procedure can be followed as presented. However, when setting the output voltage via a resistor divider placed at the input of the differential amplifier, in order to compensate for the attenuation introduced by the resistor divider, the obtained R2 value needs be multiplied by a factor of (RP+RS)/RP. The remainder of the calculations remain unchanged, as long as the compensated R2 value is used. C2 R2 COMP C3 R3 C1 - E/A R1 FB + V OSC  R1  F 0 R 2 = ----------------------------------------d MAX  V IN  F LC Ro VREF 2. Calculate C1 such that FZ1 is placed at a fraction of the FLC, at 0.1 to 0.75 of FLC (to adjust, change the 0.5 factor to desired number). The higher the quality factor of the output filter and/or the higher the ratio FCE/FLC, the lower the FZ1 frequency (to maximize phase boost at FLC). VDIFF - RGND + VSEN VOUT OSCILLATOR HALF-BRIDGE DRIVE L D PHASE LGATE ISL6567 C E EXTERNAL CIRCUIT FIGURE 24. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN The modulator transfer function is the small-signal transfer function of VOUT /VCOMP. This function is dominated by a DC gain, given by dMAXVIN /VOSC , and shaped by the output filter, with a double pole break frequency at FLC and a zero at FCE . For the purpose of this analysis, L and D represent the individual channel inductance and its DCR divided by 2 (equivalent parallel value of the two output inductors), while C and E represents the total output capacitance and its equivalent series resistance. 1 F LC = -------------------------2  L  C 1 F CE = ---------------------2  C  E (EQ. 14) The compensation network consists of the error amplifier (internal to the ISL6567) and the external R1 to R3, C1 to C3 components. The goal of the compensation network is to provide a closed loop transfer function with high 0dB crossing frequency (F0; typically 0.1 to 0.3 of FSW) and adequate phase margin (better than 45 °). Phase margin is the difference between the closed loop phase at F0dB and 180°. The equations that follow relate the compensation network’s poles, zeros and gain to the components (R1 , R2 , R3 , C1 , C2 , and C3) in Figure 23. Use the following guidelines for locating the poles and zeros of the compensation network: 1. Select a value for R1 (1k to 5k, typically). Calculate value for R2 for desired converter bandwidth (F0). If setting the output voltage via an offset resistor connected to the FB FN9243 Rev 4.00 August 9, 2011 (EQ. 16) C1 C 2 = ----------------------------------------------------2  R 2  C 1  F CE – 1 VOSC UGATE 1 C 1 = --------------------------------------------2  R 2  0.5  F LC 3. Calculate C2 such that FP1 is placed at FCE. VIN PWM CIRCUIT (EQ. 15) (EQ. 17) 4. Calculate R3 such that FZ2 is placed at FLC. Calculate C3 such that FP2 is placed below FSW (typically, 0.5 to 1.0 times FSW). FSW represents the per-channel switching frequency. Change the numerical factor to reflect desired placement of this pole. Placement of FP2 lower in frequency helps reduce the gain of the compensation network at high frequency, in turn reducing the HF ripple component at the COMP pin and minimizing resultant duty cycle jitter. R1 R3 = --------------------F SW ----------- – 1 F LC 1 C3 = -----------------------------------------------2  R3  0.7  F SW (EQ. 18) It is recommended a mathematical model is used to plot the loop response. Check the loop gain against the error amplifier’s open-loop gain. Verify phase margin results and adjust as necessary. The following equations describe the frequency response of the modulator (GMOD), feedback compensation (GFB) and closed-loop response (GCL): d MAX  V IN 1 + s f  E  C G MOD  f  = ---------------------------  ------------------------------------------------------------------------------------2 V OSC 1 + sf  E + D  C + s f  L  C 1 + s  f   R2  C 1 G FB  f  = ---------------------------------------------------  s  f   R1   C 1 + C 2  (EQ. 19) 1 + s  f    R1 + R3   C3  ----------------------------------------------------------------------------------------------------------------------  C1  C2    1 + s  f   R 3  C 3    1 + s  f   R 2   -------------------    C 1 + C 2  G CL  f  = G MOD  f   G FB  f  where s  f  = 2  f  j COMPENSATION BREAK FREQUENCY EQUATIONS 1 F Z1 = -----------------------------2  R 2  C 1 1 F Z2 = -----------------------------------------------2   R 1 + R 3   C 3 1 F P1 = ------------------------------------------C1  C2 2  R 2  ------------------C1 + C2 1 F P2 = -----------------------------2  R 3  C 3 (EQ. 20) Figure 25 shows an asymptotic plot of the DC/DC converter’s gain vs frequency. The actual Modulator Gain has a high gain Page 19 of 26 ISL6567 peak dependent on the quality factor (Q) of the output filter, which is not shown. Using the above guidelines should yield a compensation gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 against the capabilities of the error amplifier. The closed loop gain, GCL, is constructed on the log-log graph of Figure 25 by adding the modulator gain, GMOD (in dB), to the feedback compensation gain, GFB (in dB). This is equivalent to multiplying the modulator transfer function and the compensation transfer function and then plotting the resulting gain. A stable control loop has a gain crossing with close to a -20dB/decade slope and a phase margin greater than 45°. Include worst case component variations when determining phase margin. The mathematical model presented makes a number of approximations and is generally not accurate at frequencies approaching or exceeding half the switching frequency. When designing compensation networks, select target crossover frequencies in the range of 10% to 30% of the perchannel switching frequency, FSW. FP1 FP2 GAIN FZ1 FZ2 R2 20 log  --------  R1 MODULATOR GAIN COMPENSATION GAIN CLOSED LOOP GAIN OPEN LOOP E/A GAIN d MAX  V IN 20 log -------------------------------V OSC 0 GFB LOG GCL GMOD LOG FLC FCE F0 FREQUENCY FIGURE 25. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN OUTPUT FILTER DESIGN The output inductors and the output capacitor bank together form a low-pass filter responsible for smoothing the square wave voltage at the phase nodes. Additionally, the output capacitors must also provide the energy required by a fast transient load during the short interval of time required by the controller and power train to respond. Because it has a low bandwidth compared to the switching frequency, the output filter limits the system transient response leaving the output capacitor bank to supply the load current or sink the inductor currents, all while the current in the output inductors increases or decreases to meet the load demand. In high-speed converters, the output capacitor bank is amongst the costlier (and often the physically largest) parts of the circuit. Output filter design begins with consideration of the critical load parameters: maximum size of the load step, I, the load-current slew rate, di/dt, and the maximum allowable output voltage deviation under transient loading, VMAX. Capacitors are characterized according to their capacitance, ESR, and ESL (equivalent series inductance). FN9243 Rev 4.00 August 9, 2011 At the beginning of the load transient, the output capacitors supply all of the transient current. The output voltage will initially deviate by an amount approximated by the voltage drop across the ESL. As the load current increases, the voltage drop across the ESR increases linearly until the load current reaches its final value. The capacitors selected must have sufficiently low ESL and ESR so that the total output voltage deviation is less than the allowable maximum. Neglecting the contribution of inductor current and regulator response, the output voltage initially deviates according to Equation 21. di V   ESL  ----- +  ESR  I dt (EQ. 21) The filter capacitor must have sufficiently low ESL and ESR so that V < VMAX. Most capacitor solutions rely on a mixture of high-frequency capacitors with relatively low capacitance in combination with bulk capacitors having high capacitance but limited highfrequency performance. Minimizing the ESL of the highfrequency capacitors allows them to support the output voltage as the current increases. Minimizing the ESR of the bulk capacitors allows them to supply the increased current with less output voltage deviation. The ESR of the bulk capacitors is also responsible for the majority of the output-voltage ripple. As the bulk capacitors sink and source the inductor AC ripple current, a voltage develops across the bulk-capacitor ESR equal to IPP. Thus, once the output capacitors are selected and a maximum allowable ripple voltage, VPP(MAX), is determined from an analysis of the available output voltage budget, Equation 22 can be used to determine a lower limit on the output inductance.  V IN – 2  V OUT   V OUT L  ESR  ---------------------------------------------------------f S  V IN  V PP  MAX  (EQ. 22) Since the capacitors are supplying a decreasing portion of the load current while the regulator recovers from the transient, the capacitor voltage becomes slightly depleted. The output inductors must be capable of assuming the entire load current before the output voltage decreases more than VMAX. This places an upper limit on inductance. 4  C  V OUT L  -----------------------------   V MAX – I  ESR  2  I  (EQ. 23) While Equation 23 addresses the leading edge, Equation 24 gives the upper limit on L for cases where the trailing edge of the current transient causes a greater output voltage deviation than the leading edge. 2.5  C L  ----------------   V MAX – I  ESR    V IN – V O  2  I  (EQ. 24) Normally, the trailing edge dictates the selection of L, if the duty cycle is less than 50%. Nevertheless, both inequalities should be evaluated, and L should be selected based on the lower of the two results. In all equations in this paragraph, L is the per-channel inductance and C is the total output bulk capacitance. Page 20 of 26 ISL6567 LAYOUT CONSIDERATIONS There are two sets of critical components in a DC/DC converter using a ISL6567 controller. The power components are the most critical because they switch large amounts of energy. Next are small signal components that connect to sensitive nodes or supply critical bypassing current and signal coupling. MOSFETs switch very fast and efficiently. The speed with which the current transitions from one device to another causes voltage spikes across the interconnecting impedances and parasitic circuit elements. These voltage spikes can degrade efficiency, radiate noise into the circuit and lead to device overvoltage stress. Careful component layout and printed circuit design minimizes the voltage spikes in the converter. Consider, as an example, the turnoff transition of the upper PWM MOSFET. Prior to turnoff, the upper MOSFET was carrying channel current. During the turnoff, current stops flowing in the upper MOSFET and is picked up by the lower MOSFET. Any inductance in the switched current path generates a large voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide circuit traces minimize the magnitude of voltage spikes. +12VIN Although the ISL6567 allows for external adjustment of the channel-to-channel current balancing (via the RISEN resistors), it is desirable to have a symmetrical layout, preferably with the controller equidistantly located from the two power trains it controls. Equally important are the gate drive lines (UGATE, LGATE, PHASE): since they drive the power train MOSFETs using short, high current pulses, it is important to size them accordingly and reduce their overall impedance. Equidistant placement of the controller to the two power trains also helps keeping these traces equally long (equal impedances, resulting in similar driving of both sets of MOSFETs). LIN CBIN1 +5VIN (CHFIN1) (CF2) (CF1) VCC PVCC BOOT1 REFTRK UGATE1 FS RFS CBOOT1 Q1 SS LOCATE NEAR LOAD (MINIMIZE CONNECTION PATH) LOUT1 PHASE1 CSS ISEN1 RISEN1 Q2 MON LGATE1 BOOT2 RPG REN VOUT CBOOT2 ISL6567 (CHFOUT) PGOOD CBIN2 UGATE2 Q3 CBOUT (CHFIN2) EN C2 PHASE2 COMP ISEN2 C1 LGATE2 R2 RISEN2 LOUT2 RS Q4 PGND RP FB R1 GND VDIFF VSEN RGND LOCATE NEAR SWITCHING TRANSISTORS (MINIMIZE CONNECTION PATH) KEY LOCATE CLOSE TO IC (MINIMIZE CONNECTION PATH) HEAVY TRACE ON CIRCUIT PLANE LAYER ISLAND ON POWER PLANE LAYER ISLAND ON CIRCUIT PLANE LAYER VIA CONNECTION TO GROUND PLANE FIGURE 26. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS FN9243 Rev 4.00 August 9, 2011 Page 21 of 26 ISL6567 The power components should be placed first. Locate the input capacitors close to the power switches. Minimize the length of the connections between the input capacitors, CIN, and the power switches. Locate the output inductors and output capacitors between the MOSFETs and the load. Locate all the high-frequency decoupling capacitors (ceramic) as close as practicable to their decoupling target, making use of the shortest connection paths to any internal planes, such as vias to GND immediately next, or even onto the capacitor’s grounded solder pad. The critical small components include the bypass capacitors for VCC and PVCC. Locate the bypass capacitors, CBP, close to the device. It is especially important to locate the components associated with the feedback circuit close to their respective controller pins, since they belong to a high-impedance circuit loop, sensitive to EMI pick-up. It is important to place the RISEN resistors close to the respective terminals of the ISL6567. A multi-layer printed circuit board is recommended. Figure 26 shows the connections of the critical components for one output channel of the converter. Note that capacitors CxxIN and CxxOUT could each represent numerous physical capacitors. Dedicate one solid layer, usually the one underneath the component side of the board, for a ground plane and make all critical component ground connections with vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. Keep the metal runs from the PHASE terminal to inductor LOUT short. The power plane should support the input power and output power nodes. Use copper filled polygons on the top and bottom circuit layers for the phase nodes. Use the remaining printed circuit layers for small signal wiring. Size the trace interconnects commensurate with the signals they are carrying. Use narrow (0.005” to 0.008”) and short traces for the high-impedance, small-signal connections, such as the feedback, compensation, soft-start, frequency set, enable, reference track, etc. The wiring traces from the IC to the MOSFETs’ gates and sources should be wide (0.02” to 0.05”) and short, encircling the smallest area possible. Component Selection Guidelines MOSFETS The selection of MOSFETs revolves closely around the current each MOSFET is required to conduct, the switching characteristics, the capability of the devices to dissipate heat, as well as the characteristics of available heat sinking. Since the ISL6567 drives the MOSFETs with a 5V signal, the selection of appropriate MOSFETs should be done by comparing and evaluating their characteristics at this specific VGS bias voltage. The following paragraphs addressing MOSFET power dissipation ignore the driving losses associated with the gate resistance. The aggressive design of the shoot-through protection circuits, part of the ISL6567 output drivers, is geared toward reducing the deadtime between the conduction of the upper and the lower MOSFET/s. The short deadtimes, coupled with strong pull-up and pull-down output devices driving the UGATE and LGATE pins make the ISL6567 best suited to driving low gate FN9243 Rev 4.00 August 9, 2011 resistance (RG), late-generation MOSFETs. If employing MOSFETs with a nominal gate resistance in excess of 1-2, check for the circuit’s proper operation. A few examples (non exclusive list) of suitable MOSFETs to be used in ISL6567 applications include the D8 (and later) generation from Renesas and the OptiMOS®2 line from Infineon. Along the same line, the use of gate resistors is discouraged, as they may interfere with the circuits just mentioned. LOWER MOSFET POWER CALCULATION Since virtually all of the heat loss in the lower MOSFET is conduction loss (due to current conducted through the channel resistance, rDS(ON)), a quick approximation for heat dissipated in the lower MOSFET can be found using Equation 25: 2 I L ,PP  1 – D   I OUT 2 P LMOS1 = r DS  ON   ----------  1 – D  + -------------------------------12  2  (EQ. 25) where: IM is the maximum continuous output current, IL,PP is the peak-to-peak inductor current, and D is the duty cycle (approximately VOUT/VIN). An additional term can be added to the lower-MOSFET loss equation to account for additional loss accrued during the dead time when inductor current is flowing through the lowerMOSFET body diode. This term is dependent on the diode forward voltage at IM, VD(ON); the switching frequency, fS; and the length of dead times, td1 and td2, at the beginning and the end of the lower-MOSFET conduction interval, respectively.  I OUT I PP  I OUT I  PP- t P LMOS 2 = V D  ON  f S  ---------- + -------- t d1 +  ---------- – ------2  2  d2  2  2 (EQ. 26) Equation 26 assumes the current through the lower MOSFET is always positive; if so, the total power dissipated in each lower MOSFET is approximated by the summation of PLMOS1 and PLMOS2. UPPER MOSFET POWER CALCULATION In addition to rDS(ON) losses, a large portion of the upper-MOSFET losses are switching losses, due to currents conducted through the device while the input voltage is present as VDS. Upper MOSFET losses can be divided into separate components, separating the upper-MOSFET switching losses, the lower-MOSFET body diode reverse recovery charge loss, and the upper MOSFET rDS(ON) conduction loss. In most typical circuits, when the upper MOSFET turns off, it continues to conduct a decreasing fraction of the output inductor current as the voltage at the phase node falls below ground. Once the lower MOSFET begins conducting (via its body diode or enhancement channel), the current in the upper MOSFET decreases to zero. In Equation 27, the required time for this commutation is t1and the associated power loss is PUMOS,1.  I OUT I L ,PP  t 1  P UMOS ,1  V IN  ---------- + ------------  -----  f S 2  2  N (EQ. 27) Page 22 of 26 ISL6567 Similarly, the upper MOSFET begins conducting as soon as it begins turning on. Assuming the inductor current is in the positive domain, the upper MOSFET sees approximately the input voltage applied across its drain and source terminals, while it turns on and starts conducting the inductor current. This transition occurs over a time t2, and the approximate the power loss is PUMOS,2. (EQ. 28) A third component involves the lower MOSFET’s reverserecovery charge, QRR. Since the lower MOSFET’s body diode conducts the full inductor current before it has fully switched to the upper MOSFET, the upper MOSFET has to provide the charge required to turn off the lower MOSFET’s body diode. This charge is conducted through the upper MOSFET across VIN, the power dissipated as a result, PUMOS,3 can be approximated as: P UMOS ,3 = V IN Q rr f S (EQ. 29) Lastly, the conduction loss part of the upper MOSFET’s power dissipation, PUMOS,4, can be calculated using Equation 30. 2 I PP2  I OUT P UMOS ,4 = r DS  ON   ---------- d + ---------12  N  OUTPUT INDUCTOR SELECTION One of the parameters limiting the converter’s response to a load transient is the time required to change the inductor current. In a multi-phase converter, small inductors reduce the response time with less impact to the total output ripple current (as compared to single-phase converters). (EQ. 30) In this case, of course, rDS(ON) is the ON resistance of the upper MOSFET. The total power dissipated by the upper MOSFET at full load can be approximated as the summation of these results. Since the power equations depend on MOSFET parameters, choosing the correct MOSFETs can be an iterative process that involves repetitively solving the loss equations for different MOSFETs and different switching frequencies until converging upon the best solution. OUTPUT CAPACITOR SELECTION The output capacitor is selected to meet both the dynamic load requirements and the voltage ripple requirements. The load transient a microprocessor impresses is characterized by high slew rate (di/dt) current demands. In general, multiple high quality capacitors of different size and dielectric are paralleled to meet the design constraints. Should the load be characterized by high slew rates, attention should be particularly paid to the selection and placement of high-frequency decoupling capacitors (MLCCs, typically multi-layer ceramic capacitors). High frequency capacitors supply the initially transient current and slow the load rate-of-change seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (effective series resistance) and capacitance requirements. High frequency decoupling capacitors should be placed as close to the power pins of the load, or for that reason, to any decoupling target they are meant for, as physically possible. Attention should be paid as not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. FN9243 Rev 4.00 August 9, 2011 Bulk capacitor choices include aluminum electrolytic, OS-Con, Tantalum and even ceramic dielectrics. An aluminum electrolytic capacitor’s ESR value is related to the case size with lower ESR available in larger case sizes. However, the equivalent series inductance (ESL) of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slewrate transient loading. Unfortunately, ESL is not a specified parameter. Consult the capacitor manufacturer and/or measure the capacitor’s impedance with frequency to help select a suitable component. 1.0 CURRENT MULTIPLIER, KCM  I OUT I L ,PP  t 2  P UMOS , 2  V IN  ---------- – ------------  -----  f S 2  2   N Use only specialized low-ESR capacitors intended for switching-regulator applications for the bulk capacitors. The bulk capacitor’s ESR determines the output ripple voltage and the initial voltage drop following a high slew-rate transient’s edge. In most cases, multiple capacitors of small case size perform better than a single large case capacitor. 0.8 0.6 0.4 0.2 0 0 0.1 0.2 0.3 0.4 0.5 DUTY CYCLE (VO/VIN) FIGURE 27. RIPPLE CURRENT vs DUTY CYCLE The output inductor of each power channel controls the ripple current. The control IC is stable for channel ripple current (peak-to-peak) up to twice the average current. A single channel’s ripple current is approximated by: V IN – V OUT V OUT I L PP = ----------------------------  ------------V IN F SW  L (EQ. 31) The current from multiple channels tend to cancel each other and reduce the total ripple current. The total output ripple current can be determined using the curve in Figure 27; it provides the total ripple current as a function of duty cycle and number of active channels, normalized to the parameter KNORM at zero duty cycle. V OUT K NORM = ------------------L  F SW (EQ. 32) where L is the channel inductor value. Page 23 of 26 ISL6567 Find the intersection of the active channel curve and duty cycle for your particular application. The resulting ripple current multiplier from the y-axis is then multiplied by the normalization factor, KNORM, to determine the total output ripple current for the given application. (EQ. 33) I TOTAL = K NORM  K CM INPUT CAPACITOR SELECTION The important parameters for the bulk input capacitors are the voltage rating and the RMS current rating. For reliable operation, select bulk input capacitors with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage. The input RMS current required for a multi-phase converter can be approximated with the aid of Figure 28. For a more exact calculation of the input RMS current use Equation 34. I IN  RMS  = 2 2 2 D I O   D – D  + I L PP  ------12 (EQ. 34) For bulk capacitance, several electrolytic or high-capacity MLC capacitors may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surge-current at power-up. APPLICATION SYSTEM DC TOLERANCE Although the ISL6567 features a tight voltage reference, the overall system DC tolerance can be affected by the tolerance of the other components employed. The resistive divider used to set the output voltage will directly influence the system DC voltage tolerance. Figure 29 details the absolute worst case tolerance stack-up for 1% and 0.1% feedback resistors, and assuming the ISL6567 is regulating at 0.8% above its nominal reference. Other component tolerance stack-ups may be investigated using the following equation, where REFTM, RPTM, and RSTM are the tolerance multipliers corresponding to VREF, RS, and RP, respectively.  k – 1   R STM + R PTM REF TM  --------------------------------------------------------- – 1 k  R PTM TOL = -----------------------------------------------------------------------------------------100 0.2 (EQ. 35) [%] 2.8 2.6 0.1 IL,PP = 0.5 x IO IL,PP = 0.75 x IO 0 RSTM = 1.01 RPTM = 0.99 REFTM = 1.008 2.4 IL,PP = 0 0 0.1 0.2 0.3 0.4 0.5 DUTY CYCLE (VO /VIN) FIGURE 28. NORMALIZED INPUT RMS CURRENT vs DUTY CYCLE FOR A 2-PHASE CONVERTER As the input capacitors are responsible for sourcing the AC component of the input current flowing into the upper MOSFETs, their RMS current capacity must be sufficient to handle the AC component of the current drawn by the upper MOSFETs. Figure 28 can be used to determine the input-capacitor RMS current function of duty cycle, maximum sustained output current (IO), and the ratio of the peak-to-peak inductor current (IL,PP) to the maximum sustained load current, IO. FN9243 Rev 4.00 August 9, 2011 TOLERANCE (%) INPUT CAPACITOR CURRENT (IIN(RMS)/IO) 0.3 Use a mix of input bypass capacitors to control the input voltage ripple. Use ceramic capacitance for the high frequency decoupling and bulk capacitors to supply the RMS current. Minimize the connection path inductance of the high frequency decoupling ceramic capacitors (from drain of upper MOSFET to source of lower MOSFET). 2.2 2.0 1.8 1.6 1.4 RSTM = 1.001 RPTM = 0.999 REFTM = 1.008 1.2 1.0 0.8 1 2 3 4 5 6 7 8 9 10 k = VOUT/VREF FIGURE 29. WORST CASE SYSTEM DC REGULATION TOLERANCE (V REF AT 0.8% ABOVE NOMINAL) Page 24 of 26 ISL6567 © Copyright Intersil Americas LLC 2005-2011. All Rights Reserved. All trademarks and registered trademarks are the property of their respective owners. For additional products, see www.intersil.com/en/products.html Intersil products are manufactured, assembled and tested utilizing ISO9001 quality systems as noted in the quality certifications found at www.intersil.com/en/support/qualandreliability.html Intersil products are sold by description only. Intersil may modify the circuit design and/or specifications of products at any time without notice, provided that such modification does not, in Intersil's sole judgment, affect the form, fit or function of the product. Accordingly, the reader is cautioned to verify that datasheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com FN9243 Rev 4.00 August 9, 2011 Page 25 of 26 ISL6567 Package Outline Drawing L24.4x4C 24 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 2, 10/06 4.00 4X 2.5 A 20X 0.50 B PIN 1 INDEX AREA PIN #1 CORNER (C 0 . 25) 24 19 1 18 4.00 2 . 50 ± 0 . 15 13 0.15 (4X) 12 7 0.10 M C A B 0 . 07 24X 0 . 23 +- 0 . 05 4 24X 0 . 4 ± 0 . 1 TOP VIEW BOTTOM VIEW SEE DETAIL "X" 0.10 C C 0 . 90 ± 0 . 1 BASE PLANE ( 3 . 8 TYP ) SEATING PLANE 0.08 C SIDE VIEW ( 2 . 50 ) ( 20X 0 . 5 ) C 0 . 2 REF 5 ( 24X 0 . 25 ) 0 . 00 MIN. 0 . 05 MAX. ( 24X 0 . 6 ) DETAIL "X" TYPICAL RECOMMENDED LAND PATTERN NOTES: 1. Dimensions are in millimeters. Dimensions in ( ) for Reference Only. 2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994. 3. Unless otherwise specified, tolerance : Decimal ± 0.05 4. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 5. Tiebar shown (if present) is a non-functional feature. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be either a mold or mark feature. FN9243 Rev 4.00 August 9, 2011 Page 26 of 26