RT6204
500mA, 60V, 350kHz Synchronous Step-Down Converter
General Description
Features
The RT6204 is a 60V, 500mA, 350kHz, high-efficiency,
synchronous step-down DC-DC converter with an
input-voltage range of 5.2V to 60V and a programmable
output-voltage range of 0.8V to 50V. It features
current-mode control to simplify external compensation
and to optimize transient response with a wide range of
inductors and output capacitors. High efficiency can be
achieved through integrated N-MOSFETs, and
pulse-skipping mode at light loads. With EN pin,
power-up sequence can be more flexible and shutdown
quiescent current can be reduced to < 3A.
The RT6204 features cycle-by-cycle current limit for
over-current protection against short-circuit outputs,
and user-programmable soft-start time to prevent inrush
current during startup. It also includes input under-voltage
lockout, output under-voltage, and thermal shutdown
protection to provide safe and smooth operation in all
operating conditions.
0.8V Feedback Reference Voltage with 1.5%
Accuracy
Wide Input Voltage Range : 5.2V to 60V
Output Current : 500mA
Integrated N-MOSFETs
Current-Mode Control
Fixed Switching Frequency : 350kHz
Programmable Output Voltage : 0.8V to 50V
Low < 3A Shutdown Quiescent Current
Up to 92% Efficiency
Pulse-Skipping Mode for Light-Load Efficiency
Programmable Soft-Start Time
Cycle-by-Cycle Current Limit Protection
Input Under-Voltage Lockout, Output Under-Voltage
and Thermal Shutdown Protection
Applications
The RT6204 is available in the SOP-8 (Exposed pad)
package.
4-20mA Loop-Powered Sensors
OBD-II Port Power Supplies
Low-Power Standby or Bias Voltage Supplies
Industrial Process Control, Metering, and Security
Systems
High-Voltage LDO Replacement
Telecommunications Systems
Commercial Vehicle Power Supplies
General Purpose Wide Input Voltage Regulation
Simplified Application Circuit
Efficiency vs. Output Current
CBOOT
100
90
VIN
BOOT
CIN
RT6204
EN
Enable
SS
CP
VOUT
R1
FB
CFF
COMP GND
CC
CSS
80
L1
SW
R2
RC
COUT
Efficiency (%)
VIN
70
VIN = 24V
60
VIN = 36V
50
VIN = 48V
40
VIN = 60V
30
20
10
VOUT = 12V
0
0
0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45 0.5
Output Current (A)
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DS6204-00
June 2016
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RT6204
Ordering Information
Marking Information
RT6204
Package Type
SP : SOP-8 (Exposed Pad-Option 2)
RT6204GSP : Product Number
YMDNN : Date Code
RT6204
GSPYMDNN
Lead Plating System
G : Green (Halogen Free and Pb Free)
Pin Configuration
Note :
(TOP VIEW)
Richtek products are :
RoHS compliant and compatible with the current
requirements of IPC/JEDEC J-STD-020.
Suitable for use in SnPb or Pb-free soldering processes.
8
BOOT
VIN
2
SW
3
GND
4
GND
SS
7
EN
6
COMP
5
FB
9
SOP-8 (Exposed Pad)
Functional Pin Description
Pin No.
Pin Name
Pin Function
1
BOOT
Bootstrap capacitor connection node for High-Side Gate Driver. Connect a
0.1F ceramic capacitor from BOOT to SW to power the internal gate driver.
2
VIN
Supply voltage input, 5.2V to 60V. Bypass VIN to GND with a large
high-quality capacitor.
3
SW
Switch node for output inductor connection.
GND
Power ground. The exposed pad must be connected to GND and well
soldered to the input and output capacitors and a large PCB copper area for
maximum power dissipation.
5
FB
Feedback voltage input. Connect FB to the midpoint of the external
feedback resistor divider to sense the output voltage. The device regulates
the FB voltage at 0.8V (typical) Feedback Reference Voltage.
6
COMP
Compensation node for the compensation of the regulation control loop.
Connect a series RC network from COMP to GND. In some cases, another
capacitor from COMP to GND may be required.
EN
Enable control input. A logic High (VEN > 1.35V) enables the device, and a
logic Low (VEN < 0.925V) shuts down the device, reducing the supply
current to 3A or below. Connect EN pin to VIN pin with a 100k pull-up
resistor for automatic startup.
SS
Soft-start capacitor connection node. Connect an external capacitor from SS
to GND to set the soft-start time. Do not leave SS pin unconnected. A
capacitor of capacitance from 10nF to 100nF is recommended, which can
set the soft-start time from 1.33ms to 13.3ms, accordingly.
4, 9
(Exposed Pad)
7
8
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RT6204
Functional Block Diagram
VIN
EN
HV
Protection
Thermal
Shutdown
1.2V
Internal
Regulator
UVLO
+
Shutdown
Comparator
1μA
Current
Sense
BOOT
UVLO
BOOT
0.4V
Logic &
Clamp
Control
+
Gate Driver
with Dead
Time Control
UV
Comparator
FB
0.8V
HS Switch
Current
Comparator
Slope
Compensation
SS
SW
LS
+ EA
+
6μA
HS
LS Switch
Current
Comparator
Current
Sense
GND
Oscillator
COMP
Operation
The RT6204 is a synchronous step-down converter,
Pulse Skipping Operation
integrated with both high-side (HS) and low-side (LS)
MOSFETs to reduce external component count and a
gate driver with dead-time control logic to prevent
shoot-through condition from happening. The RT6204
also features constant frequency and peak
current-mode control with slope compensation. During
PWM operation, output voltage is regulated down, and
is sensed from the FB pin to be compared with an
internal 0.8V reference voltage VREF. In normal
operation, the high-side N-MOSFET is turned on when
an S-R latch is set by the rising edge of an internal
oscillator output as the PWM clock, and is turned off
when the S-R latch is reset by the output of a (high-side)
current comparator, which compares the high-side
sensed current signal with the current signal related to
the COMP voltage. While the high-side N-MOSFET is
At very light-load condition, the RT6204 provides pulse
skipping technique to decrease switching loss for better
efficiency. When load current decreases, the FB
voltage VFB will increase slightly. With VFB 1% higher
than VREF, the COMP voltage will be clamped at a
minimum value and the converter will enter into pulse
skipping mode. When the converter operates in pulse
skipping mode, the internal oscillator will be stopped,
which makes the switching period being extended. In
pulse skipping mode, as the load current decreases,
turned off, the low-side N-MOSFET will be turned on. If
the output voltage is not established, the high-side
power switch will be turned on again and another cycle
begins.
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June 2016
VFB will be discharged more slowly, which in turn will
extend the switching period even more.
Error Amplifier
The RT6204 adopts a transconductance amplifier as
the error amplifier. The error amplifier of a typical
970A/V transconductance (gm) compares the
feedback voltage VFB with the lower one of the
soft-start voltage or the internal reference voltage VREF,
0.8V. As VFB drops due to the load current increase,
the output voltage of the error amplifier will go up so
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RT6204
that the device will supply more inductor current to
match the load current. The frequency compensation
components, such as the series resistor and capacitor,
feedback voltage VFB will be compared with the
soft-start ramp voltage during soft-start time. For the
RT6204, the external capacitor CSS is required, and for
and an optional capacitor, are placed between the
COMP pin and ground.
350kHz as a fixed frequency for PWM operation.
soft-start control, the SS pin should never be left
unconnected, and it is not recommended to be
connected to an external voltage source. The soft-start
time depends on RC time constant; for example, a
0.1F capacitor for programming soft-start time will
result in 13.3ms (typ.) soft-start time.
Slope Compensation
Output Under-Voltage Protection (UVP) with Hiccup
In order to prevent sub-harmonic oscillations that may
occur over all specified load and line conditions when
operating at duty cycle higher than 50%, the RT6204
features an internal slope compensation, which adds a
compensating slope signal to the sensed current signal
to support applications with duty cycle up to 93%.
Mode
Oscillator
The internal oscillator frequency is set to a typical
The RT6204 provides under-voltage protection with
hiccup mode. When the feedback voltage VFB drops
below under-voltage protection threshold VTH-UVP, half
of the feedback reference voltage VREF, the UVP
function will be triggered to turn off the high-side
Internal Regulator
MOSFET immediately. The converter will attempt
auto-recovery soft-start after under-voltage condition
When the VIN is plugged in, the internal regulator will
generate a low voltage to drive internal control circuitry
and to supply the bootstrap power for the high-side
gate driver.
has occurred for a period of time. Once the
under-voltage condition is removed, the converter will
resume switching and be back to normal operation.
Current Limit Protection
Chip Enable
The RT6204 provides an EN pin, as an external chip
enable control, to enable or disable the device. When
VIN is higher than the input under-voltage lockout
threshold (VUVLO) with the EN voltage (VEN) higher
than 1.35V, the converter will be turned on. When VEN
is lower than 0.925V, the converter will enter into
shutdown mode, during which the supply current can
be even reduced to 3A or below.
External Soft-Start
The RT6204 provides external soft-start feature to
reduce input inrush current. The soft-start time can be
programmed by selecting the value of the capacitor
CSS connected from the SS pin to GND. An internal
current source ISS (typically, 6A) charges the external
The RT6204 provides cycle-by-cycle current limit
protection against over-load or short-circuited condition.
When the peak inductor current reaches the current
limit, the high-side MOSFET will be turned off
immediately with no violating minimum on-time tON_MIN
requirement to prevent the device from operating in an
over-current condition.
Thermal Shutdown
The RT6204 provides over-temperature protection
(OTP) function to prevent the chip from damaging due
to over-heating. The over-temperature protection
function will shut down the switching operation when
the junction temperature exceeds 165C. Once the
over-temperature condition is removed, the converter
will resume switching and be back to normal operation.
capacitor CSS to build a soft-start ramp voltage. The
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is a registered trademark of Richtek Technology Corporation.
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June 2016
RT6204
Absolute Maximum Ratings
VIN
SW
(Note 1)
(Note 5) ------------------------------------------------------------------------------------------------- 0.3V to 80V
DC----------------------------------------------------------------------------------------------------------------- 0.3V to (VIN + 0.3V)
0.063
For example, if the VIN = 50V, the VOUT should be set
higher than 3.15V.
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5V
BOOT
CBOOT
100nF
RT6204
SW
Figure 6. External Bootstrap Diode
Inductor Selection
Output inductor plays a very important role in
step-down converters because it stores energy from
input power rail and releases to output load. For better
efficiency, DC resistance (DCR) of the inductor must be
minimized to reduce copper loss. In addition, since the
inductor takes up most of the PCB space, its size also
matters. Low-profile inductors can also save board
space if height limitation exists. However, low-DCR and
low-profile inductors are usually not cost effective.
On the other hand, while larger inductance may lower
ripple current, and then power loss, rise time of the
inductor current, however, increases with inductance,
which degrades the transient responses. Therefore, the
inductor design is a trade-off among performance, size
and cost.
The first thing to consider is inductor ripple current. The
inductor ripple current is recommended in the range of
20% to 40% of full-load current, and thus the
inductance can be calculated using the following
equation.
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RT6204
LMIN =
VIN VOUT VOUT
fSW k IOUT VIN
where k is the ratio of peak-to-peak ripple current to
rated output current. From above, 0.2 to 0.4 of the ratio
k is recommended.
The next thing to consider is inductor saturation current.
Choose an inductor with saturation current rating
greater than maximum inductor peak current. The peak
inductor current can be calculated using the following
equation :
V VOUT VOUT
IL = IN
LMIN fSW VIN
where IL is the inductor peak to peak current, and
IL_PEAK = IOUT +
IL
2
Input Capacitor Selection
A high-quality ceramic capacitor of 4.7F or greater,
such as X5R or X7R, are recommended for the input
decoupling capacitor. X5R and X7R ceramic capacitors
are commonly used in power regulator applications
because the dielectric material has less capacitance
variation and more temperature stability.
Voltage rating and current rating are the key
parameters to select an input capacitor. An input
capacitor with voltage rating 1.5 times greater than the
maximum input voltage is a conservative and safe
design choice. As for current rating, the input capacitor
is used to supply the input RMS current, which can be
approximately calculated using the following equation :
IIN_RMS = IOUT
VOUT
VIN
V
1 OUT
VIN
It is practical to have several capacitors with low
equivalent series resistance (ESR), being paralleled to
form a capacitor bank, to meet size or height
requirements, and to be placed close to the drain of the
high-side MOSFET, which is very helpful in reducing
input voltage ripple at heavy load. Besides, the input
voltage ripple is determined by the input capacitance,
which can be approximately calculated by the following
equation :
VIN
IOUT(MAX) VOUT VOUT
=
1
CIN fSW
VIN
VIN
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DS6204-00
June 2016
Output Capacitor Selection
Output capacitance affects stability of the control
feedback loop, ripple voltage, and transient response.
In steady state condition, inductor ripple current flows
into the output capacitor, which results in voltage ripple.
Output voltage ripple VRIPPLE can be calculated by the
following equation :
1
VRIPPLE = IL ESR +
8 COUT fSW
where IL is the peak-to-peak inductor current.
The output inductor and capacitor form a second-order
low-pass filter for the buck converter.
It takes a few switching cycles to respond to load
transient due to the delay from the control loop. During
the load transient, the output capacitor will supply
current before the inductor can supply current high
enough to output load. Therefore, a voltage drop,
caused by the current change onto output capacitor,
and the current flowing through ESR of the capacitor,
will occur. To meet the transient response requirement,
the output capacitance should be large enough and its
ESR should be as small as possible. The output
voltage drop (V) can be calculated by the equation
below :
V = IOUT ESR +
COUT >
IOUT
tS
COUT
IOUT tS
V IOUT ESR
where IOUT is the size of the output current transient,
and tS is the control-loop delay time. For the worst-case
scenario, from no load to full load, tS is about 1 to 3
switching cycles.
Given that a transient response requirement is 4% for
5V output voltage VOUT, output current transient IOUT
is from 0A to 0.5A, ESR of the ceramic capacitor is
2m, tS is 3 switching cycles for the longest delay, and
switching frequency is 350kHz, a minimum output
capacitance 21.53F can then be calculated from
above.
Another factor for output voltage drop is equivalent
series inductance (ESL). A big change in load current,
i.e. large di/dt, along with the ESL of the capacitor,
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RT6204
causes a drop on the output voltage. A better transient
performance can be obtained by using a capacitor with
low ESL. Generally, using several capacitors connected
in parallel can have better transient performance than
using a single capacitor with the same total ESR.
operating ambient temperature for fixed TJ(MAX) and
thermal resistance, JA. The derating curve in Figure 7
allows the designer to see the effect of rising ambient
temperature on the maximum power dissipation.
Maximum Power Dissipation (W)1
4.0
External Diode Selection
In order to reduce conduction loss, an external diode
between SW pin and GND is recommended. Since a
low forward voltage of a diode may cause low
conduction loss during OFF-time, SCHOTTKY diodes
with current rating greater than maximum inductor peak
current are good design choice for the application.
During the on-time, the diode can prevent the reverse
voltage back to the input voltage. Therefore, the
voltage rating should be higher than maximum input
voltage.
PD(MAX) = (TJ(MAX) TA) / JA
where TJ(MAX) is the maximum junction temperature, TA
is the ambient temperature, and JA is the
junction-to-ambient thermal resistance.
For continuous operation, the maximum operating
junction temperature indicated under Recommended
Operating Conditions is 125C. The junction-to-ambient
thermal resistance, JA, is highly package dependent.
For a SOP-8 (Exposed Pad) package, the thermal
resistance, JA, is 29C/W on a standard JEDEC 51-7
high effective-thermal-conductivity four-layer test board.
The maximum power dissipation at TA = 25C can be
calculated as below :
PD(MAX) = (125C 25C) / (29C/W) = 3.44W for a
SOP-8 (Exposed Pad) package.
The maximum power dissipation depends on the
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3.2
2.8
2.4
2.0
1.6
1.2
0.8
0.4
0.0
0
25
50
75
100
125
Ambient Temperature (°C)
Thermal Considerations
The junction temperature should never exceed the
absolute maximum junction temperature TJ(MAX), listed
under Absolute Maximum Ratings, to avoid permanent
damage to the device. The maximum allowable power
dissipation depends on the thermal resistance of the IC
package, the PCB layout, the rate of surrounding
airflow, and the difference between the junction and
ambient temperatures. The maximum power
dissipation can be calculated using the following
formula :
Four-Layer PCB
3.6
Figure 7. Derating Curve of Maximum Power
Dissipation
Layout Considerations
PCB layout is very important for high-frequency switching
converter applications. The PCB traces can radiate
excessive noise and contribute to converter instability with
improper layout. It is good design to mount power
components and route the power traces on the same
layer. If the power trace, for example, VIN trace, must be
routed to another layer, there must be enough vias on the
power trace for passing current through with less power
loss. The width of power trace is decided by the
maximum current which may go through. With wide
traces and enough vias, resistance of the entire power
trace can be reduced to minimum to improve converter
performance. Below are some other layout guidelines,
which should be considered :
Place input decoupling capacitors close to the VIN pin.
Input capacitor can provide instant current to the
converter when high-side MOSFET is turned on. It is
better to connect the input capacitors to the VIN pin
directly with a trace on the same layer.
Place an inductor close to the SW pin and the trace
between them should be wide and short. It can gain
better efficiency with minimum resistance of the SW
trace since the output current will flow through the SW
trace. It is also a good design to keep the area of SW
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RT6204
trace as large as possible, without affecting other paths.
The area can help dissipate the heat in the internal
power stages. However, since a large voltage and
which is closest to the inductor. The feedback trace
should be also kept away from any dirty trace, for
example, a trace with high dv/dt, di/dt, or current rating,
current variation usually occur on the SW trace, any
sensitive trace should be kept away from this node.
The connection point of the feedback trace on the
VOUT side should be kept away from the current path
for the VOUT trace and be close to the output capacitor,
etc., and the total length should be kept as short as
possible to reduce the risk of noise coupling, and the
signal delay.
If possible, tie the grounds of the input capacitor and the
output capacitor together as the same reference ground.
CIN
GND
VIN
Route CBOOT
to another layer
CBOOT
CSS
BOOT
SS
VIN
EN
Compensator
GND
SW
COMP
SW
GND
FB
Resistive Voltage
Divider
L
DIODE
GND
VOUT
COUT
Figure 8. PCB Layout Guide
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RT6204
Outline Dimension
Dimensions In Millimeters
Symbol
Dimensions In Inches
Min
Max
Min
Max
A
4.801
5.004
0.189
0.197
B
3.810
4.000
0.150
0.157
C
1.346
1.753
0.053
0.069
D
0.330
0.510
0.013
0.020
F
1.194
1.346
0.047
0.053
H
0.170
0.254
0.007
0.010
I
0.000
0.152
0.000
0.006
J
5.791
6.200
0.228
0.244
M
0.406
1.270
0.016
0.050
X
2.000
2.300
0.079
0.091
Y
2.000
2.300
0.079
0.091
X
2.100
2.500
0.083
0.098
Y
3.000
3.500
0.118
0.138
Option 1
Option 2
8-Lead SOP (Exposed Pad) Plastic Package
Richtek Technology Corporation
14F, No. 8, Tai Yuen 1st Street, Chupei City
Hsinchu, Taiwan, R.O.C.
Tel: (8863)5526789
Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should
obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot assume
responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be accurate and
reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may
result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.
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