RT6212CHGJ6F

RT6212C/D 2A, 18V, 800kHz, ACOTTM Step-Down Converter General Description Features The RT6212C/D is a high-efficiency, monolithic synchronous step-down DC-DC converter that can deliver up to 2A output current from a 4.5V to 18V input supply. The RT6212C/D adopts ACOT architecture to allow the transient response to be improved and keep in constant frequency. Cycle-by-cycle current limit provides protection against shorted outputs and soft-start eliminates input current surge during start-up. Fault conditions also include output under-voltage  protection, output over-current protection, and thermal shutdown. Integrated 163m/86m MOSFETs 4.5V to 18V Supply Voltage Range 800kHz Switching Frequency ACOT Control    Feedback Reference Voltage 0.8V ± 1.5% Internal Start-Up into Pre-Biased Outputs Compact Package : TSOT-23-6 Pin Input Under-Voltage Lockout Over-Current Protection and Hiccup      Applications Set-Top Boxes Portable TVs  Ordering Information  RT6212C/D  Package Type J6F : TSOT-23-6 (FC) Lead Plating System G : Green (Halogen Free and Pb Free) UVP Option H : Hiccup  Access Point Routers DSL Modems  LCD TVs Marking Information RT6212CHGJ6F 30=DNN PSM/PWM C : PSM/PWM D : Force-PWM 30= : Product Code DNN : Date Code RT6212DHGJ6F Note : 2Z=DNN 2Z= : Product Code DNN : Date Code Richtek products are :  RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020.  Suitable for use in SnPb or Pb-free soldering processes. Simplified Application Circuit RT6212C/D BOOT VIN VIN CIN CBOOT L VOUT LX Enable EN R1 GND Copyright © 2017 Richtek Technology Corporation. All rights reserved. DS6212C/D-00 April 2017 CFF COUT FB R2 is a registered trademark of Richtek Technology Corporation. www.richtek.com 1 RT6212C/D Pin Configuration (TOP VIEW) BOOT EN 6 FB 5 4 2 3 GND LX VIN TSOT-23-6 (FC) Functional Pin Description Pin No. Pin Name Pin Function 1 GND System ground. Provides the ground return path for the control circuitry and low-side power MOSFET. 2 LX Switch node. LX is the switching node that supplies power to the output and connect the output LC filter from LX to the output load. 3 VIN Power input. Supplies the power switches of the device. 4 FB Feedback voltage input. This pin is used to set the desired output voltage via an external resistive divider. The feedback voltage is 0.8V typically. 5 EN Enable control input. Floating this pin or connecting this pin to GND can disable the device and connecting this pin to logic high can enable the device. 6 BOOT Bootstrap supply for high-side gate driver. Connect a 100nF or greater capacitor from LX to BOOT to power the high-side switch. Functional Block Diagram BOOT VIN VIN PVCC Reg VCC Minoff PVCC VIBIAS VREF UGATE OC Control LX Driver LGATE UV &OV GND GND LX VCC LX Ripple Gen. EN + + - Comparator EN VIN On-Time LX FB Copyright © 2017 Richtek Technology Corporation. All rights reserved. www.richtek.com 2 is a registered trademark of Richtek Technology Corporation. DS6212C/D-00 April 2017 RT6212C/D Operation The RT6212C/D is a synchronous step-down converter with advanced constant on-time control mode. Using the ACOTTM control mode can reduce the output capacitance and provide fast transient response. It can minimize the component size without additional external compensation network. UVLO Protection To protect the chip from operating at insufficient supply voltage, the UVLO is needed. When the input voltage of VIN is lower than the UVLO falling threshold voltage, the device will be lockout. Thermal Shutdown Current Protection The inductor current is monitored via the internal switches cycle-by-cycle. Once the output voltage drops under UV threshold, the RT6212C/D will enter hiccup mode. Copyright © 2017 Richtek Technology Corporation. All rights reserved. DS6212C/D-00 April 2017 When the junction temperature exceeds the OTP threshold value, the IC will shut down the switching operation. Once the junction temperature cools down and is lower than the OTP lower threshold, the converter will autocratically resume switching. is a registered trademark of Richtek Technology Corporation. www.richtek.com 3 RT6212C/D Absolute Maximum Ratings (Note 1)  Supply Input Voltage --------------------------------------------------------------------------------- 0.3V to 20V  Switch Node Voltage, LX ---------------------------------------------------------------------------- 0.3V to (VIN + 0.3V) < 10ns ---------------------------------------------------------------------------------------------------- 5V to 25V  BOOT Pin Voltage ------------------------------------------------------------------------------------ (VLX – 0.3V) to (VIN + 6.3V)  Other Pins ----------------------------------------------------------------------------------------------- 0.3V to 6V  Power Dissipation, PD @ TA = 25C TSOT-23-6 (FC) --------------------------------------------------------------------------------------- 1.667W  Package Thermal Resistance (Note 2) TSOT-23-6 (FC), JA --------------------------------------------------------------------------------- 60C/W TSOT-23-6 (FC), JC --------------------------------------------------------------------------------- 8C/W  Lead Temperature (Soldering, 10 sec.) ---------------------------------------------------------- 260C  Junction Temperature -------------------------------------------------------------------------------- 150C  Storage Temperature Range ----------------------------------------------------------------------- 65C to 150C  ESD Susceptibility (Note 3) HBM (Human Body Model) ------------------------------------------------------------------------- 2kV Recommended Operating Conditions (Note 4)  Supply Input Voltage --------------------------------------------------------------------------------- 4.5V to 18V  Ambient Temperature Range----------------------------------------------------------------------- 40C to 85C  Junction Temperature Range ---------------------------------------------------------------------- 40C to 125C Electrical Characteristics (VIN = 12V, TA = 25C, unless otherwise specified) Parameter Symbol Test Conditions Min Typ Max Unit Supply Voltage VIN Supply Input Operating Voltage VIN 4.5 -- 18 V Under-Voltage Lockout Threshold VUVLO 3.6 3.9 4.2 V Under-Voltage Lockout Threshold Hysteresis VUVLO -- 340 -- mV Shutdown Current ISHDN VEN = 0V -- -- 5 µA Quiescent Current IQ VEN = 2V, VFB = 0.85V -- 0.5 -- mA -- 1000 -- µs Soft-Start Soft-Start Time tSS Enable Voltage Enable Voltage Threshold VEN_R VEN rising 1.4 1.5 1.6 VEN_F VEN falling 1.18 1.28 1.38 Copyright © 2017 Richtek Technology Corporation. All rights reserved. www.richtek.com 4 V is a registered trademark of Richtek Technology Corporation. DS6212C/D-00 April 2017 RT6212C/D Parameter Symbol Test Conditions Min Typ Max Unit 0.788 0.8 0.812 V -- 163 -- -- 86 -- High-Side Switch Current Limit ILIM_H -- 5.8 -- Low-Side Switch Valley Current ILIM_L Limit 2.6 3.3 -- f SW 600 800 1000 Minimum On-Time tON_MIN 35 60 85 Minimum Off-Time tOFF_MIN 185 240 315 -- 125 -- % -- 10 -- µs UVP detect 45 50 55 Hysteresis -- 10 -- -- 5 -- Feedback Voltage VREF 4.5V ≤ VIN ≤ 18V High-Side On-Resistance RDS(ON)_H VBOOT − VLX = 4.8V Low-Side On-Resistance RDS(ON)_L Feedback Reference Voltage Internal MOSFET mΩ Current Limit A Switching Frequency Switching Frequency kHz On-Time Timer Control ns Output Under-Voltage and Over-Voltage Protections OVP Trip Threshold OVP detect OVP Propagation Delay UVP Trip Threshold UVP Propagation Delay % µs Thermal Shutdown Thermal Shutdown Threshold TSD -- 150 -- Thermal Shutdown Hysteresis TSD -- 20 -- °C Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may affect device reliability. Note 2. JA is measured under natural convection (still air) at TA = 25C with the component mounted on a high effective-thermal-conductivity four-layer test board on a JEDEC 51-7 thermal measurement standard. The first layer is filled with copper. JA is measured at the lead of the package. Note 3. Devices are ESD sensitive. Handling precaution recommended. Note 4. The device is not guaranteed to function outside its operating conditions. Copyright © 2017 Richtek Technology Corporation. All rights reserved. DS6212C/D-00 April 2017 is a registered trademark of Richtek Technology Corporation. www.richtek.com 5 RT6212C/D Typical Application Circuit RT6212C/D 3 VIN CIN 22μF Enable VIN BOOT LX 5 6 2 CBOOT 0.1μF EN GND 1 FB VOUT L 2.2μH R1 12k CFF Open COUT 22μF 4 R2 24k Table 1. Recommended Components Selection (RT6212C/DHGJ6F) VOUT (V) R1 (k) R2 (k) CFF (pF) L (H) COUT (F) 1 6 24 -- 1.2 22 1.2 12 24 -- 1.2 22 1.5 21 24 -- 2.2 22 1.8 30 24 10 to 100 2.2 22 2.5 51 24 10 to 68 2.2 22 3.3 75 24 10 to 68 2.2 22 5 126 24 10 to 68 3.3 22 Note : (1) All the input and output capacitors are the suggested values, referring to the effective capacitances, subject to any de-rating effect, like a DC bias. (2) For low output voltage application, it can optimize the load transient response of the device by adding feedforward capacitor (CFF). Copyright © 2017 Richtek Technology Corporation. All rights reserved. www.richtek.com 6 is a registered trademark of Richtek Technology Corporation. DS6212C/D-00 April 2017 RT6212C/D Typical Operating Characteristics Efficiency vs. Output Current 100 90 90 80 80 70 Efficiency (%) Efficiency (%) Efficiency vs. Output Current 100 VIN = 4.5V 60 VIN = 12V 50 VIN = 18V 40 30 70 VIN = 4.5V 60 VIN = 12V 50 VIN = 18V 40 30 20 20 10 10 RT6212C, VOUT = 1V RT6212D, VOUT = 1V 0 0 0 0.5 1 1.5 0 2 0.5 1.23 1.22 1.22 1.21 RT6212C 1.20 RT6212D 1.19 1.18 1.17 2 IOUT = 1A 1.20 1.19 IOUT = 2A 1.18 1.17 1.16 VIN = 12V, VOUT = 1.2V 1.15 1.16 0 0.5 1 1.5 4 2 6 8 10 12 14 16 18 Input Voltage (V) Output Current (A) Feedback Voltage vs. Input Voltage Feedback Voltage vs. Temperature 0.84 0.84 0.83 0.83 Feedback Voltage (V) Feedback Voltage (V) 1.5 Output Voltage vs. Input Voltage 1.23 Output Voltage (V) Output Voltage (V) Output Voltage vs. Output Current 1.24 1.21 1 Output Current (A) Output Current (A) 0.82 0.81 0.80 0.79 0.78 0.77 0.82 0.81 0.80 0.79 0.78 0.77 0.76 0.76 4 6 8 10 12 14 16 Input Voltage (V) Copyright © 2017 Richtek Technology Corporation. All rights reserved. DS6212C/D-00 April 2017 18 -50 -25 0 25 50 75 100 125 Temperature (°C) is a registered trademark of Richtek Technology Corporation. www.richtek.com 7 RT6212C/D Inductor Valley Current Limit vs. Input Voltage Inductor Valley Current Limit vs. Temperature Inductor Valley Current Limit (A)1 Inductor Valley Current Limit (A)1 4.5 4.0 3.5 3.0 2.5 2.0 Low-Side Switch 6 8 10 12 14 16 4.0 3.5 3.0 2.5 2.0 Low-Side Switch 1.5 1.5 4 4.5 -50 18 -25 0 Input Voltage (V) 75 18 3.5 15 3.0 2.5 2.0 1.5 9 6 3 VEN = 0V 0 6 8 10 12 125 12 VEN = 0V 1.0 4 100 Shutdown Current vs. Temperature 4.0 Shutdown Current (μA)1 Shutdown Current (μA)1 50 Temperature (°C) Shutdown Current vs. Input Voltage 14 16 18 -50 -25 0 25 50 75 100 125 Temperature (°C) Input Voltage (V) Quiescent Current vs. Input Voltage Quiescent Current vs. Temperature 1200 1200 1100 1100 Quiescent Current (μA) Quiescent Current (μA) 25 1000 900 800 700 600 1000 900 800 700 600 VEN = 2V, VFB = 0.85V 500 VEN = 2V, VFB = 0.85V 500 4 6 8 10 12 14 16 Input Voltage (V) Copyright © 2017 Richtek Technology Corporation. All rights reserved. www.richtek.com 8 18 -50 -25 0 25 50 75 100 125 Temperature (°C) is a registered trademark of Richtek Technology Corporation. DS6212C/D-00 April 2017 RT6212C/D Enable Threshold vs. Temperature 1.7 4.0 1.6 Enable Threshold (V) Input UVLO (V) Input UVLO vs. Temperature 4.1 Rising 3.9 3.8 3.7 Falling 3.6 3.5 1.5 Rising 1.4 1.3 Falling 1.2 1.1 3.4 1.0 -50 -25 0 25 50 75 100 125 -50 0 25 50 75 100 Temperature (°C) Power On from Input Voltage Power Off from Input Voltage VIN VIN (20V/Div) (20V/Div) VLX (10V/Div) VLX (10V/Div) VOUT (1V/Div) VOUT (1V/Div) IOUT (2A/Div) IOUT (2A/Div) VIN = 12V, VOUT = 1.2V, IOUT = 2A Time (25ms/Div) Power On from Enable Power Off from Enable VEN (5V/Div) VLX (10V/Div) VLX (10V/Div) VOUT (1V/Div) VOUT (1V/Div) IOUT (2A/Div) IOUT (2A/Div) VIN = 12V, VOUT = 1.2V, IOUT = 2A Time (2.5ms/Div) Copyright © 2017 Richtek Technology Corporation. All rights reserved. April 2017 125 VIN = 12V, VOUT = 1.2V, IOUT = 2A Time (2.5ms/Div) VEN (5V/Div) DS6212C/D-00 -25 Temperature (°C) VIN = 12V, VOUT = 1.2V, IOUT = 2A Time (2.5ms/Div) is a registered trademark of Richtek Technology Corporation. www.richtek.com 9 RT6212C/D Application Information Inductor Selection Selecting an inductor involves specifying its inductance and also its required peak current. The exact inductor value is generally flexible and is ultimately chosen to obtain the best mix of cost, physical size, and circuit efficiency. Lower inductor values benefit from reduced size and cost and they can improve the circuit's transient response, but they increase the inductor ripple current and output voltage ripple and reduce the efficiency due to the resulting higher peak currents. Conversely, higher inductor values increase efficiency, but the inductor will either be physically larger or have higher resistance since more turns of wire are required and transient response will be slower since more time is required to change current (up or down) in the inductor. A good compromise between size, efficiency, and transient response is to use a ripple current (IL) typically 40% of the desired full output load current. Calculate the approximate inductor value by selecting the input and output voltages, the switching frequency (f SW ), the maximum output current (IOUT(MAX)) and estimating a IL as some percentage of that current. L= VOUT   VIN  VOUT  VIN  fSW  IL meet the desired output current. If needed, reduce the inductor ripple current (IL) to increase the average inductor current (and the output current) while ensuring that IL(PEAK) does not exceed the upper current limit level. For best efficiency, choose an inductor with a low DC resistance that meets the cost and size requirements. For low inductor core losses some type of ferrite core is usually best and a shielded core type, although possibly larger or more expensive, will probably give fewer EMI and other noise problems. Considering the Typical Operating Circuit for 1.8V output at 2A and an input voltage of 12V, using an inductor ripple of 0.8A (40%), the calculated inductance value is : L 1.8  12  1.8   2.4μH 12  800kHz  0.8A The ripple current was selected at 0.8A and, as long as we use the calculated 2.4H inductance, that should be the actual ripple current amount. The ripple current and required peak current as below : IL = 1.8  12  1.8  = 0.8A 12  800kHz  2.4μH and IL(PEAK) = 2A  0.8A = 2.4A 2 Once an inductor value is chosen, the ripple current (IL) is calculated to determine the required peak inductor current. For the 2.4H value, the inductor's saturation and VOUT   VIN  VOUT  I and IL(PEAK) = IOUT(MAX)  L VIN  fSW  L 2 thermal rating should exceed at least 2.4A. For more conservative, the rating for inductor saturation current IL = must be equal to or greater than switch current limit of the device rather than the inductor peak current. To guarantee the required output current, the inductor needs a saturation current rating and a thermal rating that exceeds IL(PEAK). These are minimum requirements. To maintain control of inductor current in overload and short circuit conditions, some applications may desire current ratings up to the current limit value. However, the IC's output under-voltage shutdown feature make this unnecessary for most applications. IL(PEAK) should not exceed the minimum value of IC's upper current limit level or the IC may not be able to Copyright © 2017 Richtek Technology Corporation. All rights reserved. www.richtek.com 10 Input Capacitor Selection The input filter capacitors are needed to smooth out the switched current drawn from the input power source and to reduce voltage ripple on the input. The actual capacitance value is less important than the RMS current rating (and voltage rating, of course). The RMS input ripple current (IRMS) is a function of the input voltage, output voltage, and load current : IRMS = IOUT(MAX)  VOUT VIN VIN 1 VOUT is a registered trademark of Richtek Technology Corporation. DS6212C/D-00 April 2017 RT6212C/D Ceramic capacitors are most often used because of their low cost, small size, high RMS current ratings, and robust surge current capabilities. However, take care For the Typical Operating Circuit for 1.8V output and an inductor ripple of 0.8A, with 1 x 22F output capacitance each with about 5m ESR including PCB when these capacitors are used at the input of circuits supplied by a wall adapter or other supply connected through long, thin wires. Current surges through the inductive wires can induce ringing at the RT6212C/D input which could potentially cause large, damaging voltage spikes at VIN. If this phenomenon is observed, some bulk input capacitance may be required. Ceramic capacitors (to meet the RMS current requirement) can be placed in parallel with other types such as tantalum, electrolytic, or polymer (to reduce ringing and overshoot). trace resistance, the output voltage ripple components are : Choose capacitors rated at higher temperatures than required. Several ceramic capacitors may be paralleled to meet the RMS current, size, and height requirements of the application. The typical operating circuit uses two 10F and one 0.1F low ESR ceramic capacitors on VRIPPLE(ESR) = 0.8A  5m = 4mV 0.8A  5.68mV 8  22μF  800kHz VRIPPLE = 4mV  5.68mV  9.68mV VRIPPLE(C) = Output Transient Undershoot and Overshoot In addition to voltage ripple at the switching frequency, the output capacitor and its ESR also affect the voltage sag (undershoot) and soar (overshoot) when the load steps up and down abruptly. The ACOT transient response is very quick and output transients are usually small. However, the combination of small ceramic output the input. capacitors (with little capacitance), low output voltages Output Capacitor Selection (with little stored charge in the output capacitors), and The RT6212C/D are optimized for ceramic output capacitors and best performance will be obtained using them. The total output capacitance value is usually determined by the desired output voltage ripple level and transient response requirements for sag (undershoot on positive load steps) and soar (overshoot on negative load steps). Output Ripple Output ripple at the switching frequency is caused by the inductor current ripple and its effect on the output capacitor's ESR and stored charge. These two ripple components are called ESR ripple and capacitive ripple. Since ceramic capacitors have extremely low ESR and relatively little capacitance, both components are similar in amplitude and both should be considered if ripple is critical. VRIPPLE = VRIPPLE(ESR)  VRIPPLE(C) VRIPPLE(ESR) = IL  RESR VRIPPLE(C) = IL 8  COUT  fSW Copyright © 2017 Richtek Technology Corporation. All rights reserved. DS6212C/D-00 April 2017 low duty cycle applications (which require high inductance to get reasonable ripple currents with high input voltages) increases the size of voltage variations in response to very quick load changes. Typically, load changes occur slowly with respect to the IC's 800kHz switching frequency. But some modern digital loads can exhibit nearly instantaneous load changes and the following section shows how to calculate the worst-case voltage swings in response to very fast load steps. The output voltage transient undershoot and overshoot each have two components : the voltage steps caused by the output capacitor's ESR, and the voltage sag and soar due to the finite output capacitance and the inductor current slew rate. Use the following formulas to check if the ESR is low enough (typically not a problem with ceramic capacitors) and the output capacitance is large enough to prevent excessive sag and soar on very fast load step edges, with the chosen inductor value. is a registered trademark of Richtek Technology Corporation. www.richtek.com 11 RT6212C/D The amplitude of the ESR step up or down is a function of the load step and the ESR of the output capacitor : VESR _STEP = IOUT x RESR The amplitude of the capacitive sag is a function of the load step, the output capacitor value, the inductor value, the input-to-output voltage differential, and the maximum duty cycle. The maximum duty cycle during a fast transient is a function of the on-time and the minimum off-time since the ACOTTM control scheme will ramp the current using on-times spaced apart with minimum off-times, which is as fast as allowed. Calculate the approximate on-time (neglecting parasites) and maximum duty cycle for a given input and output voltage as : tON = VOUT tON and DMAX = VIN  fSW tON  tOFF_MIN The actual on-time will be slightly longer as the IC compensates for voltage drops in the circuit, but we can neglect both of these since the on-time increase compensates for the voltage losses. Calculate the output voltage sag as : L  (IOUT ) 2  COUT   VIN(MIN)  DMAX  VOUT  2 VSAG = The amplitude of the capacitive soar is a function of the load step, the output capacitor value, the inductor value and the output voltage : VSOAR = L  (IOUT )2 2  COUT  VOUT For the Typical Operating Circuit for 1.8V output, the circuit has an inductor 2.4H and 1 x 22F output capacitance with 5m ESR each. The ESR step is 2A x 5m = 10mV which is small, as expected. The output voltage sag and soar in response to full 0A-2A-0A instantaneous transients are : 1.8V = 187ns 12V  800kHz 187ns and DMAX = = 0.44 187ns  240ns tON = where 240ns is the minimum off time. VSAG  2.4μH  (2A)2  63mV 2  22μF  12V  0.44  1.8V  VSOAR  2.4μH  (2A)2  121mV 2  22μF  1.8V Copyright © 2017 Richtek Technology Corporation. All rights reserved. www.richtek.com 12 The sag is about 3.5% of the output voltage and the soar is a full 6.7% of the output voltage. The ESR step is negligible here but it does partially add to the soar, so keep that in mind whenever using higher-ESR output capacitors. The soar is typically much worse than the sag in high input, low-output step-down converters because the high input voltage demands a large inductor value which stores lots of energy that is all transferred into the output if the load stops drawing current. Also, for a given inductor, the soar for a low output voltage is a greater voltage change and an even greater percentage of the output voltage. Any sag is always short-lived, since the circuit quickly sources current to regain regulation in only a few switching cycles. With the RT6212D, any overshoot transient is typically also short-lived since the converter will sink current, reversing the inductor current sharply until the output reaches regulation again. The RT6212C discontinuous operation at light loads prevents sinking current so, for that IC, the output voltage will soar until load current or leakage brings the voltage down to normal. Most applications never experience instantaneous full load steps and the RT6212C/D high switching frequency and fast transient response can easily control voltage regulation at all times. Also, since the sag and soar both are proportional to the square of the load change, if load steps were reduced to 1A (from the 2A examples preceding) the voltage changes would be reduced by a factor of almost ten. For these reasons sag and soar are seldom an issue except in very low-voltage CPU core or DDR memory supply applications, particularly for devices with high clock frequencies and quick changes into and out of sleep modes. In such applications, simply increasing the amount of ceramic output capacitor (sag and soar are directly proportional to capacitance) or adding extra bulk capacitance can easily eliminate any excessive voltage transients. In any application with large quick transients, always calculate soar to make sure that over-voltage protection will not be triggered. Under-voltage is not likely since the threshold is very low (50%), that function has a long delay (5s), and the IC will quickly is a registered trademark of Richtek Technology Corporation. DS6212C/D-00 April 2017 RT6212C/D return the output to regulation. Over-voltage protection has a minimum threshold of 125% and short delay of 10s and can actually be triggered by incorrect component choices, particularly for the RT6212C which does not sink current. Feed-Forward Capacitor (CFF) The RT6212C/D are optimized for ceramic output capacitors and for low duty cycle applications. However for high-output voltages, with high feedback attenuation, the circuit's response becomes over-damped and transient response can be slowed. In high-output voltage circuits (VOUT > 3.3V) transient response is improved by adding a small “feed-forward” capacitor (CFF) across the upper FB divider resistor (Figure 1), to increase the circuit's Q and reduce damping to speed up the transient response without affecting the steady-state stability of the circuit. Choose a suitable capacitor value that following below step.  Get the BW the quickest method to do transient response form no load to full load. Confirm the damping frequency. The damping frequency is BW. Enable Operation (EN) For automatic start-up the low-voltage EN pin can be connected to VIN through a 100k resistor. Its large hysteresis band makes EN useful for simple delay and timing circuits. EN can be externally pulled to VIN by adding a resistor-capacitor delay (REN and CEN in Figure 2). Calculate the delay time using EN's internal threshold where switching operation begins (1.5V, typical). An external MOSFET can be added to implement digital control of EN when no system voltage above 2V is available (Figure 3). In this case, a 100k pull-up resistor, REN, is connected between VIN and the EN pin. MOSFET Q1 will be under logic control to pull down the EN pin. To prevent enabling circuit when VIN is smaller than the VOUT target value or some other desired voltage level, a resistive voltage divider can be placed between the input voltage and ground and connected to EN to create an additional input under voltage lockout threshold (Figure 4). EN VIN REN EN RT6212C/D CEN GND Figure 2. External Timing Control BW VIN REN 100k EN RT6212C/D Q1 Enable VOUT R1 GND CFF Figure 3. Digital Enable Control Circuit FB RT6212C/D R2 GND VIN REN1 EN REN2 RT6212C/D GND Figure 1. CFF Capacitor Setting  CFF can be calculated base on below equation : CFF  1 2  3.1412  R1 BW  0.8 Figure 4. Resistor Divider for Lockout Threshold Setting Output Voltage Setting Set the desired output voltage using a resistive divider from the output to ground with the midpoint connected Copyright © 2017 Richtek Technology Corporation. All rights reserved. DS6212C/D-00 April 2017 is a registered trademark of Richtek Technology Corporation. www.richtek.com 13 RT6212C/D to FB. The output voltage is set according to the following equation : VOUT = 0.8V x (1 + R1 / R2) enhancement due to undercharging the BOOT capacitor), use the external diode shown in Figure 6 to charge the BOOT capacitor and place the resistance between BOOT and the capacitor/diode connection. VOUT 5V R1 FB RT6212C/D BOOT R2 GND Figure 5. Output Voltage Setting Place the FB resistors within 5mm of the FB pin. Choose R2 between 10k and 100k to minimize power consumption without excessive noise pick-up and calculate R1 as follows : R1  R2  (VOUT  VREF ) VREF For output voltage accuracy, use divider resistors with 1% or better tolerance. External BOOT Bootstrap Diode When the input voltage is lower than 5.5V it is recommended to add an external bootstrap diode between VIN (or VINR) and the BOOT pin to improve enhancement of the internal MOSFET switch and improve efficiency. The bootstrap diode can be a low cost one such as 1N4148 or BAT54. External BOOT Capacitor Series Resistance The internal power MOSFET switch gate driver is optimized to turn the switch on fast enough for low power loss and good efficiency, but also slow enough to reduce EMI. Switch turn-on is when most EMI occurs since VLX rises rapidly. During switch turn-off, LX is discharged relatively slowly by the inductor current during the dead time between high-side and low-side switch on-times. In some cases it is desirable to reduce EMI further, at the expense of some additional power dissipation. The switch turn-on can be slowed by placing a small (