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RT6253BHGJ6F

RT6253BHGJ6F

  • 厂商:

    RICHTEK(台湾立锜)

  • 封装:

    TSOT-23-6

  • 描述:

    RT6253BHGJ6F

  • 数据手册
  • 价格&库存
RT6253BHGJ6F 数据手册
RT6253A/B 17V Input, 3A, ACOT® Buck Converter with Both FETs OC Protection General Description Features The RT6253A/B is a simple, easy-to-use, 3A synchronous step-down DC-DC converter with an input supply voltage range of 4.5V to 17V. The device ⚫ ⚫ Integrated 95m and 50m FETs Continuous Current  TSOT-23-6 possesses an accurate reference voltage and integrates low RDS(ON) power MOSFETs to achieve high efficiency.  (FC) : 3A SOT-563 (FC) : 2.5A Input Supply Voltage Range : 4.5V to 17V Output Voltage Range : 0.765V to 7V ® ⚫ Advanced Constant On-Time (ACOT ) Control  Ultrafast Transient Response  Optimized for Low-ESR Ceramic Output Capacitors ⚫ High Accuracy Feedback Reference Voltage : 1% ⚫ Optional for Operation Modes : ⚫ ⚫ The RT6253A/B adopts Advanced Constant On-Time (ACOT® ) control architecture to provide an ultrafast transient response with few external components and to operate in nearly constant switching frequency over the line, load, and output voltage range. The RT6253A operates in automatic PSM that maintains high efficiency during light load operation. The RT6253B operates in Forced PWM that helps meet tight voltage regulation accuracy requirements. ⚫ The RT6253A/B senses both FETs current for a robust over-current protection (OCP). The device features cycle-by-cycle current limit protection to prevent the device from the catastrophic damage in output short circuit, over-current or inductor saturation conditions. A built-in soft-start function prevents inrush current during start-up. The device also includes input under-voltage lockout, output under-voltage protection, and overtemperature protection (OTP) to provide safe and ⚫ ⚫ ⚫ ⚫  RT6253A : Power Saving Mode (PSM)  RT6253B : Forced PWM Mode Fixed Switching Frequency : 580kHz Enable Control and Fixed Soft-Start with typ. 1ms Safe Start-Up from Pre-biased Output Input Under-Voltage Lockout (UVLO) Protection Function  Output Under-Voltage Protection (UVP) with Hiccup Mode  High- / Low-side MOSFET OCP and OTP Function smooth operation in all operating conditions. ⚫ RoHS Compliant and Halogen Free Simplified Application Circuit RT6253A/B VIN VIN RBOOT (Optional) BOOT CBOOT CIN L VOUT SW Enable EN RFB1 GND CFF COUT FB RFB2 Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 October 2021 is a registered trademark of Richtek Technology Corporation. www.richtek.com 1 RT6253A/B Ordering Information Applications Set-Top Boxes ⚫ LCD TVs ⚫ RT6253A/B Package Type J6F : TSOT-23-6 (FC) H6F : SOT-563 (FC) Lead Plating System G : Green (Halogen Free and Pb Free) UVP Option H : Hiccup PWM Operation Mode A : Automatic PSM B : Forced PWM Home Networking Devices Surveillance ⚫ General Purpose ⚫ ⚫ Pin Configuration (TOP VIEW) BOOT EN FB 6 5 4 2 3 SW VIN Note : Richtek products are :  RoHS compliant and compatible with the current GND requirements of IPC/JEDEC J-STD-020.  Suitable for use in SnPb or Pb-free soldering processes. Marking Information TSOT-23-6 (FC) FB EN BOOT 6 5 4 1 2 3 RT6253AHGJ6F 36=DNN 36= : Product Code DNN : Date Code VIN SW GND SOT-563 (FC) RT6253BHGJ6F 35=DNN 35= : Product Code DNN : Date Code RT6253AHGH6F 05W 05 : Product Code W : Date Code RT6253BHGH6F 04W 04 : Product Code W : Date Code Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 2 is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B Functional Pin Description Pin No. Pin Name TSOT-23-6 (FC) SOT-563 (FC) Pin Function 1 3 GND Power ground. 2 2 SW Switch node between the internal switch and the synchronous rectifier. Connect this pin to the inductor and bootstrap capacitor. 3 1 VIN Power input. The input voltage range is from 4.5V to 17V. Connect input bypass capacitors directly to this pin and GND pins. The MLCC with capacitance higher than 20F is recommended. 4 6 FB Feedback voltage input. Connect this pin to the midpoint of the external feedback resistive divider to set the output voltage of the converter to the desired regulation level. The device regulates the FB voltage at feedback reference voltage. 5 5 EN Enable control input. Connect this pin to logic high enables the device and connect this pin to GND disables the device. 6 4 BOOT Bootstrap capacitor connection node to supply the high-side gate driver. Connect a 0.1F ceramic capacitor between this pin and the SW pin. Functional Block Diagram VIN SW EN + - VEN_TH UVLO Internal Regulator VCC PVCC OnTime REN_DN BOOT OC UV Protection 65% Control Soft-Start Ramp Gen. + + - FB Gate Driver & Dead-Time Control SW Comparator MIN OFF Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 October 2021 GND is a registered trademark of Richtek Technology Corporation. www.richtek.com 3 RT6253A/B Operation The RT6253A/B is a high-efficiency, synchronous stepdown DC-DC converter that can deliver up to 3A output current from a 4.5V to 17V input supply. Advanced Constant On-Time Control and PWM Operation The RT6253A/B adopts ACOT® control for its ultrafast transient response, low external component counts and stable with low ESR MLCC output capacitors. When the feedback voltage falls below the feedback reference voltage, the minimum off-time one-shot (200ns, typ.) has timed out and the inductor current is below the current limit threshold, then the internal on-time oneshot circuitry is triggered and the high-side switch is turn-on. Since the minimum off-time is short, the device exhibits ultrafast transient response and enables the use of smaller output capacitance. The on-time is inversely proportional to input voltage and directly proportional to output voltage to achieve pseudo-fixed frequency over the input voltage range. After the on-time one-shot timer expired, the high-side switch is turned off and the low-side switch is turned on until the on-time one-shot is triggered again. To achieve stable operation with low-ESR ceramic output capacitors, an internal ramp signal is added to the feedback reference voltage to simulate the output voltage ripple. held below a logic-low threshold voltage (VEN_L) of the enable input (EN), the converter will disable output voltage, that is, the converter is disabled and switching is inhibited even if the VIN voltage is above VIN undervoltage lockout threshold (VUVLO). During shutdown mode, the supply current can be reduced to ISHDN (10A or below). If the EN voltage rises above the logichigh threshold voltage (VEN_H) while the VIN voltage is higher than UVLO threshold, the device will be turned on, that is, switching being enabled and soft-start sequence being initiated. An internal resistor REN_DN from EN to GND allows EN float to shutdown the chip. Soft-Start (SS) The RT6253A/B provides an internal soft-start feature for inrush control, and the output voltage starts to rise in 0.3ms from EN rising edge. At power up, the internal capacitor is charged by an internal current source to generate a soft-start ramp voltage as a reference voltage to the PWM comparator. The device will initiate switching and the output voltage will smoothly ramp up to its targeted regulation voltage only after this ramp voltage is greater than the feedback voltage VFB to ensure the converter has a smooth start-up from prebiased output. VIN = 12V VIN VCC = 5V Power Saving Mode (RT6253A Only) The RT6253A automatically enters power saving mode (PSM) at light load to maintain high efficiency. As the load current decreases, the inductor current ripple valley eventually touches the zero current, which is the VCC EN 0.3ms tSS VOUT boundary between continuous conduction and discontinuous conduction modes. The low-side switch is turned off when the zero inductor current is detected. In this case, the output capacitor is only discharged by load current so that the switching frequency decreases. As the result, the light-load efficiency can be enhanced due to lower switching loss. Enable Control The RT6253A/B provides an EN pin, as an external chip enable control, to enable or disable the device. If VEN is Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 4 Input Under-Voltage Lockout In addition to the EN pin, the RT6253A/B also provides enable control through the VIN pin. It features an undervoltage lockout (UVLO) function that monitors the internal linear regulator (VCC). If VEN rises above VEN_H first, switching will still be inhibited until the VIN voltage rises above VUVLO. It is to ensure that the internal regulator is ready so that operation with not-fullyenhanced internal MOSFET switches can be prevented. is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B After the device is powered up, if the input voltage VIN goes below the UVLO falling threshold voltage (VUVLO − VUVLO), this switching will be inhibited; if VIN rises above the UVLO rising threshold (VUVLO), the device will resume normal operation with a complete soft-start. Output Under-Voltage Protection and Hiccup Mode The RT6253A/B includes output under-voltage protection (UVP) against over-load or short-circuited condition by constantly monitoring the feedback voltage VFB. If VFB drops below the under-voltage protection trip threshold (typically 65% of the internal feedback reference voltage), the UV comparator will go high to turn off both the internal high-side and low-side MOSFET switches. If the output under-voltage condition continues for a period of time, the RT6253A/B will enter output undervoltage protection with hiccup mode. During hiccup mode, the IC will shut down for tHICCUP_OFF (15ms), and then attempt to recover automatically for tHICCUP_ON (1.8ms). Upon completion of the soft-start sequence, if the fault condition is removed, the converter will resume normal operation; otherwise, such cycle for autorecovery will be repeated until the fault condition is cleared. The hiccup mode allows the circuit to operate safely with low input current and power dissipation, and then the converter resumes normal operation as soon as the over-load or short-circuit condition is removed. protection on both the high-side and low-side MOSFETs and prevents the device from the catastrophic damage in output short-circuit, over-current or inductor saturation conditions. The high-side MOSFET over-current protection is achieved by an internal current comparator that monitors the current in the high-side MOSFET during each on-time. The switch current is compared with the high-side switch peak-current limit (ILIM_H) after a certain amount of delay when the high-side switch being turned on each cycle. If an over-current condition occurs, the converter will immediately turns off the high-side switch and turns on the low-side switch to prevent the inductor current exceeding the high-side current limit. The low-side MOSFET over-current protection is achieved by measuring the inductor current through the synchronous rectifier (low-side switch) during the lowside on-time. Once the current rises above the low-side switch valley current limit (ILIM_L), the on-time one-shot will be inhibited until the inductor current ramps down to the current limit level (ILIM_L), that is, another on-time can only be triggered when the inductor current goes below the low-side current limit. If the output load current exceeds the available inductor current (clamped by the low-side current limit), the output capacitor needs to supply the extra current such that the output voltage will begin to drop. If it drops below the output undervoltage protection trip threshold, the IC will stop switching to avoid excessive heat. Output short VOUT, 2V/Div Negative Over-Current Limit Fault condition removed Resume normal operation The RT6253B is the part which is forced to PWM and allows negative current operation. ISW , 4A/Div In case of PWM operation, high negative current may be generated as an external power source is tied to VSW , 10/Div output terminal unexpectedly. As the risk described above, the internal circuit monitors negative current in each on-time interval of low-side MOSFET and compares it with NOC threshold. 10ms/Div Once the negative current exceeds the NOC threshold, the low-side MOSFET is turned off immediately, and The Over-Current Protection The RT6253A/B features cycle-by-cycle current-limit Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 October 2021 then the high-side MOSFET will be turned on to discharge the energy of output inductor. This behavior can keep the valley of negative current at NOC threshold to protect low-side MOSFET. However, the is a registered trademark of Richtek Technology Corporation. www.richtek.com 5 RT6253A/B negative current can’t be limited at NOC threshold anymore since minimum off-time is reached. Thermal Shutdown The RT6253A/B includes an over-temperature protection (OTP) circuitry to prevent overheating due to excessive power dissipation. The OTP will shut down switching operation when junction temperature exceeds a thermal shutdown threshold (TSD). Once the junction temperature cools down by a thermal shutdown hysteresis (TSD), the IC will resume normal operation with a complete soft-start. Note that the over temperature protection is intended to protect the device during momentary overload conditions. The protection is activated outside of the absolute maximum range of operation as a secondary fail-safe and therefore should not be relied upon operationally. Continuous operation above the specified absolute maximum operating junction temperature may impair the reliability of the device or permanently damage the device. Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 6 is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B Absolute Maximum Ratings (Note 1) ⚫ Supply Input Voltage, VIN ------------------------------------------------------------------------------ −0.3V to 20V ⚫ Enable Voltage, EN -------------------------------------------------------------------------------------- −0.3V to 20V ⚫ Switch Voltage, SW -------------------------------------------------------------------------------------- −0.3V to 20.3V < 100ns ----------------------------------------------------------------------------------------------------- −5V to 25V ⚫ BOOT Voltage , BOOT---------------------------------------------------------------------------------- – 0.3V to 26V ⚫ BOOT to SW, VBOOT − VSW --------------------------------------------------------------------------- −0.3V to 6V ⚫ Feedback Voltage, FB ---------------------------------------------------------------------------------- −0.3V to 6V ⚫ Lead Temperature (Soldering, 10 sec.) ------------------------------------------------------------- 260C ⚫ Junction Temperature ----------------------------------------------------------------------------------- 150C ⚫ Storage Temperature Range -------------------------------------------------------------------------- −65C to 150C ⚫ Power Dissipation, PD @ TA = 25C TSOT-23-6 (FC) ------------------------------------------------------------------------------------------ 1.64W SOT-563 (FC) --------------------------------------------------------------------------------------------- 1.25W ESD Ratings ⚫ ESD Susceptibility (Note 2) HBM (Human Body Model) ---------------------------------------------------------------------------- 2kV Recommended Operating Conditions (Note 3) ⚫ Supply Input Voltage ------------------------------------------------------------------------------------ 4.5V to 17V ⚫ Junction Temperature Range ------------------------------------------------------------------------- −40C to 125C Thermal Information (Note 4 and Note 5) Thermal Parameter TSOT-23-6 (FC) SOT-563 (FC) Unit JA Junction-to-ambient thermal resistance (JEDEC standard) 88.7 104.3 C/W JC(Top) Junction-to-case (top) thermal resistance 76.9 62.1 C/W JC(Bottom) Junction-to-case (bottom) thermal resistance 6 8.4 C/W JA(EVB) Junction-to-ambient thermal resistance (specific EVB) 61 80.6 C/W JC(Top) Junction-to-top characterization parameter 13.9 6.7 C/W JB Junction-to-board characterization parameter 31.53 45.17 C/W Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 October 2021 is a registered trademark of Richtek Technology Corporation. www.richtek.com 7 RT6253A/B Electrical Characteristics TSOT-23-6 (FC) (VIN = 12V, TA = 25C, unless otherwise specified ) Parameter Symbol Test Conditions Min Typ Max Unit Supply Voltage VIN Supply Input Operating Voltage VIN 4.5 -- 17 V Under-Voltage Lockout Threshold VUVLO 3.7 4 4.3 V Under-Voltage Lockout Threshold Hysteresis VUVLO -- 400 -- mV Shutdown Current ISHDN VEN = 0V -- -- 4 µA Quiescent Current IQ VEN = 2V, VFB = 0.8V -- 280 -- µA -- 1 -- ms Soft-Start Soft-Start Time tSS Enable Voltage Enable Voltage Threshold EN Pin Pull-Down Resistance VENH EN high-level input voltage 1.16 1.25 1.34 VENL EN low-level input voltage 1.01 1.1 1.19 REN_DN EN pin resistance to GND, VEN = 12V 225 450 900 k V Feedback Voltage and Discharge Resistance Feedback Threshold Voltage VFB VOUT = 1.05V 758 765 772 mV Feedback Input Current IFB VFB = 0.8V, TA = 25°C −0.1 0 0.1 A High-Side On-Resistance RDS(ON)_H VBOOT – VSW = 4.8V -- 95 -- Low-Side On-Resistance RDS(ON)_L -- 50 -- High-Side Switch Current Limit ILIM_H -- 5.6 -- Low-Side Switch Valley Current Limit ILIM_L Internal MOSFET mΩ Current Limit A 3.2 4.2 5.2 -- 580 -- kHz -- 60 -- ns VFB = 0.5V -- 200 260 ns Hiccup detect -- 65 -- % Switching Frequency Switching Frequency f SW VOUT = 1.05V, PWM mode On-Time Timer Control Minimum On-Time tON_MIN Minimum Off-Time tOFF_MIN Output Under-Voltage Protections UVP Trip Threshold VUVP Hiccup Power On-Time tHICCUP_ON -- 1.8 -- Hiccup Power Off-Time tHICCUP_OFF -- 15 -- Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 8 ms is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B Parameter Symbol Test Conditions Min Typ Max Unit Thermal Shutdown Thermal Shutdown Threshold TSD -- 155 -- Thermal Shutdown Hysteresis TSD -- 35 -- °C Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 October 2021 is a registered trademark of Richtek Technology Corporation. www.richtek.com 9 RT6253A/B SOT-563 (FC) (VIN = 12V, TA = 25C, unless otherwise specified) Parameter Symbol Test Conditions Min Typ Max Unit Supply Voltage VIN Supply Input Operating Voltage VIN 4.5 -- 17 V Under-Voltage Lockout Threshold VUVLO 3.9 4.2 4.5 V Under-Voltage Lockout Threshold Hysteresis VUVLO -- 420 -- mV Shutdown Current ISHDN VEN = 0V -- -- 4 µA Quiescent Current IQ VEN = 2V, VFB = 0.85V -- 295 -- µA -- 0.95 -- ms Soft-Start Soft-Start Time tSS Enable Voltage Enable Voltage Threshold EN Pin Pull-Down Resistance VENH EN high-level input voltage 1.24 1.31 1.38 VENL EN low-level input voltage 1.09 1.16 1.23 REN_DN EN pin resistance to GND, VEN = 12V 225 450 900 k V Feedback Voltage and Discharge Resistance Feedback Threshold Voltage VFB VOUT = 1.05V 799 807 815 mV Feedback Input Current IFB VFB = 0.85V, TA = 25°C −0.1 0 0.1 A High-Side On-Resistance RDS(ON)_H VBOOT – VSW = 4.8V -- 95 -- Low-Side On-Resistance RDS(ON)_L -- 50 -- High-Side Switch Current Limit ILIM_H -- 5.6 -- Low-Side Switch Valley Current Limit ILIM_L 3.45 4.4 5.35 -- 580 -- kHz -- 60 -- ns VFB = 0.5V -- 190 250 ns Hiccup detect -- 65 -- % Internal MOSFET mΩ Current Limit A Switching Frequency Switching Frequency f SW VOUT = 1.05V, PWM mode On-Time Timer Control Minimum On-Time tON_MIN Minimum Off-Time tOFF_MIN Output Under-Voltage Protections UVP Trip Threshold VUVP Hiccup Power On-Time tHICCUP_ON -- 1.8 -- Hiccup Power Off-Time tHICCUP_OFF -- 15 -- Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 10 ms is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B Parameter Symbol Test Conditions Min Typ Max Unit Thermal Shutdown Thermal Shutdown Threshold TSD -- 155 -- Thermal Shutdown Hysteresis TSD -- 35 -- °C Note 1. Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may affect device reliability. Note 2. Devices are ESD sensitive. Handling precaution recommended. Note 3. The device is not guaranteed to function outside its operating conditions. Note 4. θJA and θJC are measured or simulated at TA = 25C based on the JEDEC 51-7 standard. Note 5. θJA(EVB), ΨJC(TOP) and ΨJB are measured on a high effective-thermal-conductivity four-layer test board which is in size of 70mm x 50mm; furthermore, all layers with 1 oz. Cu. Thermal resistance/parameter values may vary depending on the PCB material, layout, and test environmental conditions. Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 October 2021 is a registered trademark of Richtek Technology Corporation. www.richtek.com 11 RT6253A/B Typical Application Circuit RT6253A/B VIN CIN 10μF x 2 CIN 0.1μF Enable VIN BOOT RBOOT (Optional) CBOOT 0.1μF L VOUT SW EN RFB1 GND CFF COUT FB RFB2 Table 1. Recommended Components Selection VOUT (V) RFB1 (k) RFB2 (k) CFF (pF) L (H) COUT (F) 52.3 10 10 to 100 3.3 to 4.7 20 to 68 33.2 30.9 10 10 to 100 2.2 to 4.7 20 to 68 2.5 22.6 21 10 10 to 100 2.2 to 4.7 20 to 68 1.8 13.7 12.4 10 10 to 100 1.5 to 4.7 20 to 68 1.5 9.53 8.66 10 -- 1.5 to 4.7 20 to 68 1.2 5.76 4.87 10 -- 1.5 to 4.7 20 to 68 1.0 3.09 2.4 10 -- 1.5 to 4.7 20 to 68 TSOT-23-6 (FC) SOT-563 (FC) 5.0 54.9 3.3 Note : (1) Please do not use a CFF higher than 100pF due to the noise coupling consideration. (2) Considering effective capacitance de-rating which is related to biased voltage level and size, the effective capacitance of COUT at target output level should meet the value in above table to make converter operated in stable and normal. Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 12 is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B Typical Operating Characteristics L : WE-74404054022 (DCR = 19m) for VOUT = 1V and 1.8V. L : WE-74404054047 (DCR = 30m) for VOUT = 3.3V and 5V. Efficiency vs. Output Current Efficiency vs. Output Current 100 100 90 90 80 70 VOUT = 3.3V 60 VOUT = 1.8V Efficiency (%) Efficiency (%) 80 VOUT = 1V 50 40 30 VOUT = 5V 60 VOUT = 3.3V 50 VOUT = 1.8V 40 VOUT = 1V 30 20 20 10 0 0.001 70 10 RT6253AHGJ6F, VIN = 5V 0.01 0.1 1 0 0.001 10 RT6253AHGJ6F, VIN = 12V 0.01 Efficiency vs. Output Current 100 90 90 80 70 VOUT = 3.3V 60 VOUT = 1.8V 50 VOUT = 1V Efficiency (%) Efficiency (%) 80 40 30 20 70 VOUT = 5V 60 VOUT = 3.3V 50 VOUT = 1.8V 40 VOUT = 1V 30 20 10 10 RT6253BHGJ6F, VIN = 5V 0.01 0.1 1 0 0.001 10 RT6253BHGJ6F, VIN = 12V 0.01 Efficiency vs. Output Current 1 10 Efficiency vs. Output Current 100 100 90 90 80 80 VOUT = 1.8V 70 Efficiency (%) Efficiency (%) 0.1 Output Current (A) Output Current (A) VOUT = 1V 60 50 40 30 20 70 VOUT = 5V 60 VOUT = 3.3V 50 VOUT = 1.8V 40 VOUT = 1V 30 20 10 10 RT6253AHGH6F, VIN = 5V 0.01 0.1 1 Output Current (A) Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 10 Efficiency vs. Output Current 100 0 0.001 1 Output Current (A) Output Current (A) 0 0.001 0.1 October 2021 10 0 0.001 RT6253AHGH6F, VIN = 12V 0.01 0.1 1 10 Output Current (A) is a registered trademark of Richtek Technology Corporation. www.richtek.com 13 RT6253A/B Efficiency vs. Output Current 100 90 90 80 80 70 VOUT = 1.8V 60 VOUT = 1V Efficiency (%) Efficiency (%) Efficiency vs. Output Current 100 50 40 70 VOUT = 1.8V 40 VOUT = 1V 30 20 20 10 RT6253BHGH6F, VIN = 5V 0 0.001 0.01 0.1 1 VOUT = 3.3V 50 30 10 VOUT = 5V 60 RT6253BHGH6F, VIN = 12V 0 0.001 10 0.01 0.1 1 10 Output Current (A) Output Current (A) Output Voltage vs. Output Current Output Voltage vs. Output Current 5.50 1.20 1.10 Output Voltage (V) Output Voltage (V) 1.15 1.05 1.00 RT6253AHGJ6F, VIN = 5V 0.95 RT6253BHGJ6F, VIN = 5V 0.90 RT6253AHGJ6F, VIN = 12V 5.25 5.00 RT6253AHGJ6F, VIN = 9V RT6253BHGJ6F, VIN = 9V RT6253AHGJ6F, VIN = 12V 4.75 RT6253BHGJ6F, VIN = 12V RT6253BHGJ6F, VIN = 12V 0.85 VOUT = 5V VOUT = 1V 4.50 0.80 0 0.5 1 1.5 2 2.5 0 3 0.5 1 1.5 2 2.5 3 Output Current (A) Output Current (A) Output Voltage vs. Output Current Output Voltage vs. Output Current 1.20 5.50 1.10 Output Voltage (V) Output Voltage (V) 1.15 1.05 1.00 RT6253AHGH6F, VIN = 5V 0.95 RT6253BHGH6F, VIN = 5V 0.90 RT6253AHGH6F, VIN = 12V 0.85 RT6253BHGH6F, VIN = 12V 5.25 5.00 RT6253AHGH6F, VIN = 9V RT6253BHGH6F, VIN = 9V 4.75 RT6253AHGH6F, VIN = 12V RT6253BHGH6F, VIN = 12V VOUT = 5V VOUT = 1V 4.50 0.80 0 0.5 1 1.5 2 2.5 Output Current (A) Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 14 3 0 0.5 1 1.5 2 2.5 3 Output Current (A) is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B Output Voltage vs. Input Voltage Output Voltage vs. Input Voltage 1.10 5.5 5.3 1.05 1.00 Output Voltage (V) Output Voltage (V) 5.4 RT6253AHGJ6F RT6253BHGJ6F 0.95 5.2 5.1 5.0 RT6253AHGJ6F RT6253BHGJ6F 4.9 4.8 4.7 4.6 VOUT = 1V VOUT = 5V 4.5 0.90 5 6 7 8 9 9 10 11 12 13 14 15 16 17 10 11 12 13 14 15 16 17 Input Voltage (V) Input Voltage (V) Output Voltage vs. Input Voltage Output Voltage vs. Input Voltage 5.5 1.10 5.4 1.00 Output Voltage (V) Output Voltage (V) 5.3 1.05 RT6253AHGH6F RT6253BHGH6F 0.95 5.2 5.1 5.0 RT6253AHGH6F RT6253BHGH6F 4.9 4.8 4.7 4.6 VOUT = 1V VOUT = 5V 4.5 0.90 5 6 7 8 9 9 10 11 12 13 14 15 16 17 10 11 12 13 14 Quiescent Current vs. Temperature 16 17 Shutdown Current vs. Temperature 340 1.0 RT6253AHGJ6F RT6253AHGJ6F VIN = 17V Shutdown Current (μA)1 320 Quiescent Current (μA) 15 Input Voltage (V) Input Voltage (V) 300 280 260 VIN = 17V 240 VIN = 12V VIN = 9V 220 VIN = 12V 0.5 VIN = 9V VIN = 5V 0.0 -0.5 VIN = 5V -1.0 200 -50 -25 0 25 50 75 100 Temperature (°C) Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 October 2021 125 -50 -25 0 25 50 75 100 125 Temperature (°C) is a registered trademark of Richtek Technology Corporation. www.richtek.com 15 RT6253A/B Quiescent Current vs. Temperature Shutdown Current vs. Temperature 340 1.0 RT6253AHGH6F RT6253AHGH6F VIN = 17V Shutdown Current (μA)1 Quiescent Current (μA) 320 300 280 VIN = 17V 260 VIN = 12V 240 VIN = 9V VIN = 5V 220 VIN = 12V 0.5 VIN = 9V VIN = 5V 0.0 -0.5 -1.0 200 -50 -25 0 25 50 75 100 -50 125 -25 0 0.80 0.84 0.79 0.83 0.78 0.77 0.76 0.75 0.74 0.73 -25 0 25 50 100 125 0.81 0.80 0.79 0.78 0.77 0.75 -50 75 0.82 0.76 RT6253AHGJ6F 0.71 75 100 -50 125 -25 0 25 50 RT6253AHGH6 F 75 100 125 Temperature (°C) Temperature (°C) Frequency vs. Input Voltage Frequency vs. Output Current 700 600 500 Frequency (kHz)1 650 Frequency (kHz)1 50 Reference Voltage vs. Temperature 0.85 Reference Voltage (V) Reference Voltage (V) Reference Voltage vs. Temperature 0.81 0.72 25 Temperature (°C) Temperature (°C) 600 550 500 400 300 200 100 450 5 6 7 8 9 10 11 12 13 14 15 16 17 Input Voltage (V) Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 16 0 0.001 0.01 0.1 1 10 Output Current (A) is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B Frequency vs. Temperature EN Threshold vs. Temperature 700 1.4 650 1.3 EN Threshold (V) Frequency (kHz)1 EN VIH 600 550 1.2 EN VIL 1.1 500 1.0 450 0.9 RT6253AHGJ6F -50 -25 0 25 50 75 100 -50 125 -25 0 25 EN Threshold vs. Temperature 100 125 4.1 4.0 EN VIH UVLO_H 3.9 1.3 Input Voltage (V) EN Threshold (V) 75 UVLO vs. Temperature 1.4 1.2 EN VIL 1.1 3.8 3.7 UVLO_L 3.6 3.5 3.4 3.3 3.2 RT6253HGH6F RT6253AHGJ6F 3.1 1.0 -50 -25 0 25 50 75 100 -50 125 UVLO vs. Temperature 0 25 50 75 100 125 Power On from EN 4.4 4.3 VOUT (5V/Div) UVLO_H 4.2 -25 Temperature (°C) Temperature (°C) Input Voltage (V) 50 Temperature (°C) Temperature (°C) 4.1 VEN (2V/Div) 4.0 3.9 VSW (10V/Div) UVLO_L 3.8 3.7 3.6 3.5 RT6253AHGH6F 3.4 -50 -25 0 25 50 75 100 Temperature (°C) Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 October 2021 125 IL (2A/Div) VIN = 12V, VOUT = 5V IOUT = 3A, L = 4.7H Time (1ms/Div) is a registered trademark of Richtek Technology Corporation. www.richtek.com 17 RT6253A/B Power Off from EN VOUT (5V/Div) VOUT (5V/Div) VEN (2V/Div) VIN (10V/Div) VSW (10V/Div) IL (2A/Div) Power On from VIN VSW (10V/Div) VIN = 12V, VOUT = 5V IOUT = 3A, L = 4.7H IL (2A/Div) Time (5ms/Div) Time (200s/Div) Output Ripple as IOUT = 10mA Power Off from VIN VOUT (5V/Div) VOUT (50mV/Div) VIN (10V/Div) VSW (10V/Div) IL (2A/Div) VSW (5V/Div) VIN = 12V, VOUT = 5V IOUT = 3A, L = 4.7H IL (2A/Div) VIN = 12V, VOUT = 5V, IOUT = 10mA, L = 4.7H Time (5ms/Div) Time (10s/Div) Output Ripple as IOUT = 3A Load Transient (No Load to Full Load) VOUT (20mV/Div) VOUT (100mV/Div) VIN = 12V, VOUT = 5V IOUT = 0A to 3A, L = 4.7H VSW (5V/Div) IL (2A/Div) VIN = 12V, VOUT = 5V, IOUT = 3A, L = 4.7H Time (1s/Div) Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 18 VIN = 12V, VOUT = 5V IOUT = 3A, L = 4.7H IOUT (1A/Div) Time (400s/Div) is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B Over-Current Protection and UVP Load Transient (Half Load to Full Load) VOUT (2V/Div) VOUT (100mV/Div) VIN = 12V, VOUT = 5V IOUT = 1A to 3A, L = 4.7H IOUT (1A/Div) VSW (10V/Div) IL (2A/Div) Time (400s/Div) Time (20s/Div) Short Circuit Protection Short Circuit before Power On VOUT (2V/Div) VIN = 12V, VOUT = 5V, L = 4.7H VOUT (2V/Div) VIN = 12V, VOUT = 5V, L = 4.7H VIN (10V/Div) VSW (10V/Div) VSW (10V/Div) IL (2A/Div) IL (2A/Div) Time (10ms/Div) Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 VIN = 12V, VOUT = 5V L = 4.7H October 2021 Time (10ms/Div) is a registered trademark of Richtek Technology Corporation. www.richtek.com 19 RT6253A/B Application Information The output stage of a synchronous buck converter is composed of an inductor and capacitor, which stores and delivers energy to the load, and forms a secondorder low-pass filter to smooth out the switch node voltage to maintain a regulated output voltage. the calculated inductance value is : L= 1.2  (12 − 1.2) = 2.48μH 12  580kHz  0.75A For the typical application, a standard inductance value of 2.2H can be selected. Inductor Selection 1.2  (12 − 1.2) = 0.85A (28.3% of the IC rated current) 12  580kHz  2.2μH The inductor selection trade-offs among size, cost, efficiency, and transient response requirements. IL = Generally, three key inductor parameters are specified for operation with the device: inductance value (L), and IL(PEAK) = 3A + 0.85A = 3.425A 2 inductor saturation current (ISAT), and DC resistance For the 2.2H value, the inductor's saturation and thermal rating should exceed at least 3.425A. For more conservative, the rating for inductor saturation current must be equal to or greater than switch current limit of the device rather than the inductor peak current. (DCR). A good compromise between size and loss is to choose the peak-to-peak ripple current equals to 20% to 50% of the IC rated current. The switching frequency, input voltage, output voltage, and selected inductor ripple current determines the inductor value as follows : L= VOUT  ( VIN − VOUT ) VIN  fSW  IL Once an inductor value is chosen, the ripple current (IL) is calculated to determine the required peak inductor current. IL = VOUT  ( VIN − VOUT ) I and IL(PEAK) = IOUT(MAX) + L VIN  fSW  L 2 IL(PEAK) should not exceed the minimum value of IC's upper current limit level. Besides, the current flowing through the inductor is the inductor ripple current plus the output current. During power up, faults, or transient load conditions, the inductor current can increase above the calculated peak inductor current level calculated above. For EMI sensitive application, choosing shielding type inductor is preferred. Input Capacitor Selection Input capacitance, CIN, is needed to filter the pulsating current at the drain of the high-side power MOSFET. CIN should be sized to do this without causing a large variation in input voltage. The waveform of CIN ripple voltage and ripple current are shown in Figure 1. The peak-to-peak voltage ripple on input capacitor can be estimated as the equation below : VCIN = D  IOUT   1− D  + IOUT  ESR  CIN  fSW  where D= VOUT VIN  In transient conditions, the inductor current can increase For ceramic capacitors, the equivalent series resistance up to the switch current limit of the device. For this reason, (ESR) is very low, the ripple which is caused by ESR can be ignored, and the minimum input capacitance can the most conservative approach is to specify an inductor with a saturation current rating which is equal to or greater than the switch current limit rather than the peak inductor current. Considering the Typical Application Circuit for 1.2V output at 3A and an input voltage of 12V, using an inductor ripple of 0.75A (25% of the IC rated current), Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 20 be estimated as the equation below : CIN_MIN = IOUT_MAX  D (1− D ) VCIN_MAX  fSW where VCIN_MAX  200mV is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B capacitor should be 0402 or 0603 in size. VCIN CIN Ripple Voltage VESR = IOUT x ESR (1-D) x IOUT CIN Ripple Current D x IOUT D x tSW (1-D) x tSW Figure 1. CIN Ripple Voltage and Ripple Current In addition, the input capacitor needs to have a very low ESR and must be rated to handle the worst-case RMS input current of : IRMS  IOUT_MAX  VOUT  VIN VIN −1 VOUT It is common to use the worse IRMS  IOUT/2 at VIN = 2VOUT for design. Note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life which makes it advisable to further de-rate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size, Output Capacitor Selection The RT6253A/B are optimized for ceramic output capacitors and best performance will be obtained using them. The total output capacitance value is usually determined by the desired output voltage ripple level and transient response requirements for sag (undershoot on load apply) and soar (overshoot on load release). Output Ripple The output voltage ripple at the switching frequency is a function of the inductor current ripple going through the output capacitor’s impedance. To derive the output voltage ripple, the output capacitor with Capacitance (COUT) and its equivalent series resistance (RESR) must be taken into consideration. The output peak-to-peak ripple voltage (VRIPPLE) caused by the inductor current ripple (IL) is characterized by two components, which are ESR ripple (VRIPPLE(ESR)) and capacitive ripple (VRIPPLE(C)) and can be expressed as below : VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C) VRIPPLE(ESR) = IL  RESR IL 8  COUT  fSW height and thermal requirements in the design. For low input voltage applications, sufficient bulk input VRIPPLE(C) = capacitance is needed to minimize transient effects during output load changes. As ceramic capacitors are used, both parameters should be estimated due to the extremely low ESR and relatively small capacitance. Refer to the RT6253A/B's typical application circuit of 1.2V application, the actual inductor current ripple (IL) is 0.85A, and the output Ceramic capacitors are ideal for switching regulator applications because of its small size, robustness, and very low ESR. However, care must be taken when these capacitors are used at the input. A ceramic input capacitor combined with trace or cable inductance forms a high quality (under damped) tank circuit. If the RT6253A/B circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the device’s rating. This situation is easily avoided by placing the low ESR ceramic input capacitor in parallel with a bulk capacitor with higher ESR to damp the voltage ringing. The input capacitor should be placed as close as possible to the VIN pins, with a low inductance connection to the GND of the IC. In addition to a larger bulk capacitor, a small ceramic capacitors of 0.1F should be placed close to the VIN and GND pin. This Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 October 2021 capacitors are 2 x 22F (Murata ceramic capacitor : GRM219R60J226ME47), VRIPPLE can be obtained as below. The ripple caused by ESR (2m) can be calculated as : VRIPPLE(ESR) = 0.85A  2m = 1.7mV Considering the capacitance derating, the effective capacitance is approximately 18F as the output voltage is 1.2V, and another parameter is : 0.85A = 5.1mV 8  2  18μF  580kHz VRIPPLE = 1.7mV + 5.1mV = 6.8mV VRIPPLE( C) = Output Transient Undershoot and Overshoot is a registered trademark of Richtek Technology Corporation. www.richtek.com 21 RT6253A/B In addition to voltage ripple at the switching frequency, the output capacitor and its ESR also affect the voltage sag (undershoot) and soar (overshoot) when the load steps up and down abruptly. The ACOT® transient response is very quick and output transients are usually small. The following section shows how to calculate the worst-case voltage swings in response to very fast load steps. and the output voltage : Both undershoot voltage and overshoot voltage consist of two factors : the voltage steps caused by the output capacitor's ESR, and the voltage sag and soar due to the finite output capacitance and the inductor current slew rate. Use the following formulas to check if the ESR is low enough (typically not a problem with ceramic capacitors) and the output capacitance is large enough to prevent excessive sag and soar on very fast load step edges, with the chosen inductor value. Output Voltage Setting VSOAR = L  (IOUT )2 2  COUT  VOUT Because some modern digital loads can exhibit nearly instantaneous load changes, the amplitude of the ESR should be taken into consideration while calculating the VSAG & VSOAR. Set the desired output voltage using a resistive divider from the output to ground with the midpoint connected to FB, as shown in Figure 2. The output voltage is set according to the following equation : VOUT = VREF x (1 + RFB1 / RFB2) VOUT RFB1 The amplitude of the ESR step up or down is a function of the load step and the ESR of the output capacitor : FB RT6253A/B VESR _STEP = IOUT x RESR The amplitude of the capacitive sag is a function of the load step, the output capacitor value, the inductor value, the input-to-output voltage differential, and the maximum duty cycle. The maximum duty cycle during a fast transient is a function of the on-time and the minimum off-time since the ACOT® control scheme will ramp the current using on-times spaced apart with minimum off-times, which is as fast as allowed. Calculate the approximate on-time (neglecting parasites) and maximum duty cycle for a given input and output voltage as : tON = VOUT tON and DMAX = VIN  fSW tON + tOFF_MIN The real on-time will slightly extend due to the voltage drop which is related to output current; however, this ontime compensation can be neglected. Besides, the minimum on-time is 60ns, typ. If the calculated on-time is smaller than minimum on-time, it and VOUT will both be clamped. Calculate the output voltage sag as : VSAG = L  (IOUT )2 2  COUT  ( VIN(MIN)  DMAX − VOUT ) The amplitude of the capacitive soar is a function of the load step, the output capacitor value, the inductor value Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 22 RFB2 GND Figure 2. Output Voltage Setting Place the FB resistors within 5mm of the FB pin. Choose RFB2 between 10k and 100k to minimize power consumption without excessive noise pick-up and calculate RFB1 as follows : RFB1 = RFB2  (VOUT − VREF ) VREF For output voltage accuracy, use divider resistors with 1% or better tolerance. Feed-Forward Capacitor Selection (CFF) The RT6253A/B is optimized for low duty-cycle applications and the control loop is stable with low ESR ceramic output capacitors. In higher duty-cycle applications (higher output voltages or lower input voltages), the internal ripple signal will increase in amplitude. Before the ACOT® control loop can react to an output voltage fluctuation, the voltage change on the feedback signal must exceed the internal ripple amplitude. Because of the large internal ripple in this condition, the response may become slower and underdamped. This situation will result in ringing waveform at output terminal. In case of high output voltage is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B application, the phenomenon described above is more visible because of large attenuation in feedback network. As shown in Figure 3, adding a feedforward capacitor (CFF) across the upper feedback resistor is recommended. This increases the damping of the control system. calculated CFF, please decrease feedforward capacitor CFF. the value of L VOUT SW RT6253A/B RFB1 CFF COUT FB GND RFB2 Figure 3. Feedback Loop with Feedforward Capacitor Loop stability can be checked by viewing the load transient response. A load step with a speed that exceeds the converter bandwidth must be applied. For ACOT® , loop bandwidth can be in the order of 100 to 200kHz, so a load step with 500ns maximum rising time (dI/dt ≈ 2A/s) ensures the excitation frequency is sufficient. It is important that the converter operates in PWM mode, outside the light load efficiency range, and below any current limit threshold. A load transient from 30% to 60% of maximum load is reasonable which is shown in Figure 4. Figure 5. Load Transient Response With and Without Feedforward Capacitor Enable Operation The RT6253A/B is enabled when the VIN pin voltage rises above VUVLO while the EN pin voltage exceeds VEN_H. The RT6253A/B is disabled when the VIN pin voltage falls below VUVLO − VUVLO or when the EN pin voltage is below VEN_L. An internal pull-down resistor REN_DN, which is connected form EN to GND, ensures that the chip still stays in shutdown even if EN pin is floated. For automatic start-up, the EN pin, with high-voltage rating, can be connected to the input supply VIN directly as shown in Figure 6. fCO 60% Load 30% Load Figure 4. Example of Measuring the Converter BW by Fast Load Transient CFF can be calculated basing on below equation : CFF = 1 2  BW 1  1 + 1  RFB1  RFB1 RFB2  The built-in hysteresis band makes the EN pin useful for simple delay and timing circuits. The EN pin can be externally connected to VIN by adding a resistor REN and a capacitor CEN, as shown in Figure 7, to have an additional delay. The time delay can be calculated by the equation below with the EN's internal threshold, at which switching operation begins. CEN = t Rth  ln Figure 5. shows the transient performance with and without feedfoward capacitor. Note that, after defining the CFF please also check the load regulation, because feedforward capacitor might inject an offset voltage into VOUT to cause VOUT inaccuracy. If the output voltage is over spec caused by Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 October 2021 Vth Vth − VEN_H , where Rth = REN // REN_DN Vth = VIN  REN_DN REN_DN + REN is a registered trademark of Richtek Technology Corporation. www.richtek.com 23 RT6253A/B An external MOSFET can be used for logic control which is shown in Figure 8. In this case, REN is connected between VIN and the EN pin. The MOSFET RT6253A/B VIN REN1 Q1 will be under logic control to pull down the EN pin. If the device is desired to be shut down by EN pin before VIN falls below the UVLO threshold, a resistive divider (REN1 and REN2) can be used to externally set the input under-voltage lockout threshold as shown in Figure 9. For a given REN1, REN2 can be found by the equation below for the desired VIN stop voltage. VIN_STOP  REN2 //REN_DN < VEN_L REN1 + REN2 //REN_DN After REN1 and REN2 are defined, the input voltage VIN_START is obtained from VEN_H  REN1 + REN2 //REN_DN = VIN_START REN2 //REN_DN EN REN2 RENL_DN Figure 9. Resistor Divider for Lockout Threshold Setting If VIN shuts down faster than VOUT and VOUT is larger than 3.7V, buck converter becomes boost converter and generates negative current. To prevent these condition, EN should be shut down before VIN falls below VOUT. Therefore, the resistor divider for lockout threshold is recommended. Bootstrap Driver Supply VIN RT6253A/B EN REN_DN Figure 6. Automatic Start-Up Setting VIN The bootstrap capacitor (CBOOT) between the BOOT pin and the SW pin is used to create a voltage rail above the applied input voltage, VIN. Specifically, the bootstrap capacitor is charged through an internal diode to a voltage equal to approximately PVCC each time the low-side switch is turned on. The charge on this capacitor is then used to supply the required current during the remainder of the switching cycle. For most applications, a 0.1F, 0603 ceramic capacitor with X5R is recommended and the capacitor should have a 6.3 V or higher voltage rating. RT6253A/B REN External Bootstrap Diode (Optional) EN CEN REN_DN Figure 7. External Timing Control VIN RT6253A/B REN EN Enable Q1 A bootstrap capacitor of 0.1F low-ESR ceramic capacitor is connected between the BOOT and SW pins to supply the high-side gate driver. It is recommended to add an external bootstrap diode between an external 5V voltage supply and the BOOT pin as shown in Figure 10 to improve efficiency when the input voltage is below 5.5V. The bootstrap diode can be a low-cost one, such as 1N4148 or BAT54. The external 5V can be a fixed 5V voltage supply from the system, or a 5V output voltage generated by the RT6253A/B. Note that the BOOT voltage VBOOT must be lower than 5.5V. REN_DN Figure 8. Digital Enable Control Circuit Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 24 is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B 5V 5V DBOOT DBOOT RBOOT BOOT BOOT CBOOT 0.1μF RT6253A/B SW SW Figure 10. External Bootstrap Diode Figure12. External Bootstrap Diode and Resistor at the BOOT Pin External Bootstrap Resistor (Optional) The gate driver of an internal power MOSFET, utilized as a high-side switch, is optimized for turning on the switch. The gate driver is not only fast enough for reducing switching power loss, but also slow enough for minimizing EMI. The EMI issue is worse when the switch is turned on rapidly due to the induced high di/dt noises. When the high-side switch is turned off, the discharging time on SW node is relatively slow because there’s the presence of dead time, both high-side and low-side MOSFETs are turned off in this interval. In some cases, it is desirable to reduce EMI further, even at the expense of some additional power dissipation. The turn-on rate of the high-side switch can be slowed by placing a small bootstrap resistor RBOOT between the BOOT pin and the external bootstrap capacitor as shown in Figure 11. The recommended range for the RBOOT is several ohms to 47 ohms, and it could be 0402 or 0603 in size. This will slow down the rates of the high-side switch turn on and the rise of VSW . In order to improve EMI performance and enhancement of the internal MOSFET switch, the recommended application circuit is shown in Figure 12, which includes an external bootstrap diode for charging the bootstrap capacitor and a bootstrap resistor RBOOT placed between the BOOT pin and the capacitor/diode connection. SW Figure 11. External Bootstrap Resistor at the BOOT Pin Copyright © 2021 Richtek Technology Corporation. All rights reserved. October 2021 In many applications, the RT6253A/B does not generate much heat due to its high efficiency and low thermal resistance of its TSOT-23-6 (FC) package. However, in applications which the RT6253A/B runs at a high ambient temperature and high input voltage, the generated heat may exceed the maximum junction temperature of the part. The RT6253A/B includes an over-temperature protection (OTP) circuitry to prevent overheating due to excessive power dissipation. If the junction temperature reaches approximately 155°C, the RT6253A/B stops switching the power MOSFETs until the temperature is cooled down by 35°C. Note that the over temperature protection is intended to protect the device during momentary overload conditions. The protection is activated outside of the absolute maximum range of operation as a secondary fail-safe and therefore should not be relied upon operationally. Continuous operation above the specified absolute maximum operating junction temperature may impair device reliability or permanently damage the device. The maximum power dissipation can be calculated by the following formula : ( CBOOT 0.1μF RT6253A/B Thermal Considerations ) PD(MAX) = TJ(MAX) − TA / θJA(EFFECTIVE) RBOOT BOOT DS6253A/B-03 CBOOT 0.1μF RT6253A/B where TJ(MAX) is the maximum allowed junction temperature of the die. For recommended operating condition specifications, the maximum junction temperature is 125°C. TA is the ambient operating temperature, and θJA(EFFECTIVE) is the system-level junction to ambient thermal resistance. It can be estimated from thermal modeling or measurements in is a registered trademark of Richtek Technology Corporation. www.richtek.com 25 RT6253A/B the system. As a result, the new power dissipation is 1.182W due to The thermal resistance of the device strongly depends on the surrounding PCB layout and can be improved by providing a heat sink of surrounding copper ground. The addition of backside copper with thermal vias, stiffeners, and other enhancements can also help reduce thermal resistance. the variation of RDS(ON). Therefore, the estimated new As an example, considering the RT6253A is used in application where VIN = 12V, IOUT = 3A, f SW = 580kHz, VOUT = 5V. The efficiency at 5V, 3A is 90.63% by using WE-74404054047 (4.7H, 30m DCR) as the inductor and measured at room temperature. The core loss can be obtained from its website, and it's 131mW. In this case, the power dissipation of the RT6253A is temperature of 125°C, care should be taken to reduce ( ) 1− η 2  POUT − IO  DCR + PCORE = 1.15W η Considering the system-level θJA(EFFECTIVE) is 67°C/W (other heat sources are also considered), the junction temperature of the regulator operating in a 25°C ambient temperature is approximately : TJ = 1.08W  67C/W + 25C = 102C Figure 13 shows the RT6253A/B RDS(ON) versus different junction temperatures. If the application requires a higher ambient temperature, the device power dissipation and the junction temperature of the device need to be recalculated based on a higher RDS(ON) since it increases with temperature. TJ' = 1.182W  67C/W + 40C = 119.2C If the application requires a higher ambient temperature and may exceed the recommended maximum junction the temperature rise of the part by using a heat sink or air flow. Resistance vs. Temperature 250 200 Resistance (mΩ) PD, RT = junction temperature is 150 RDS(ON)_H 100 50 RDS(ON)_L 0 -50 -25 0 25 50 75 100 125 Temperature (°C) Figure 13. RT6253A/B RDS(ON) vs. Temperature Using 40°C ambient temperature as an example. Due to the variation of junction temperature is dominated by the ambient temperature, the T'J at 40°C ambient temperature can be pre-estimated as TJ' = 102C + ( 40C − 25C) = 117C According to Figure 13, the increasing RDS(ON) can be found as RDS( ON) _H = 130.4m (at 117C) − 125m (102C ) = 5.4m RDS( ON) _L = 65.5m (at 117C) − 63.2m (102C ) = 2.3m The external power dissipation caused by the increasing RDS(ON) at higher temperature can be calculated as PD,RDS( ON) = ( 3A )  2 5 5  2   5.4m + ( 3A )   1 −  12  12  2.3m = 0.032W Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 26 is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B Layout Considerations  Follow the PCB layout guidelines below for optimal performance of the device.   Keep the high-current paths short, especially at the ground terminals. This practice is essential for stable and jitter-free operation. The high current path comprising of input capacitor, high-side FET, inductor, and the output capacitor should be as short as possible. This practice is essential for high efficiency. Place the input MLCC capacitors as close to the VIN SW node is with high frequency voltage swing and should be kept at small area. Keep analog components away from the SW node to prevent stray capacitive noise pickup.  Connect feedback network behind the output capacitors. Place the feedback components next to the FB pin.  For better thermal performance, design a wide and thick plane for GND pin or add a lot of vias to GND plane. An example of PCB layout guide is shown in Figure 14. and GND pins as possible. The major MLCC capacitors should be placed on the same layer as the RT6253A/B. Add extra vias for thermal dissipation Keep the SW node at small area and keep analog components away from the SW node to prevent stray capacitive noise pickup. GND GND CBOOT RBOOT SW SW 1 REN VIN The VIN trace should have enough width, and use several vias to shunt the high input current. L CIN3 CIN2 Place the input MLCC capacitors as close to the VIN and GND pins as possible. COUT1 Connect feedback network behind the output Place the feedback components next to the FB pin. RFB1 RFB2 CIN1 VOUT COUT2 CFF VOUT GND GND Figure 14. Layout Guide Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 October 2021 is a registered trademark of Richtek Technology Corporation. www.richtek.com 27 RT6253A/B Outline Dimension Dimensions In Millimeters Symbol Dimensions In Inches Min. Max. Min. Max. A 0.700 1.000 0.028 0.039 A1 0.000 0.100 0.000 0.004 B 1.397 1.803 0.055 0.071 b 0.300 0.559 0.012 0.022 C 2.591 3.000 0.102 0.118 D 2.692 3.099 0.106 0.122 e 0.950 0.037 H 0.080 0.254 0.003 0.010 L 0.300 0.610 0.012 0.024 TSOT-23-6 (FC) Surface Mount Package Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 28 is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B Symbol Dimensions In Millimeters Dimensions In Inches Min Max Min Max A 0.500 0.600 0.020 0.024 A1 0.000 0.050 0.000 0.002 A3 0.080 0.180 0.003 0.007 b 0.150 0.300 0.006 0.012 D 1.500 1.700 0.059 0.067 E 1.500 1.700 0.059 0.067 E1 1.100 1.300 0.043 0.051 e 0.500 0.020 L 0.100 0.300 0.004 0.012 L1 0.200 0.400 0.008 0.016 SOT-563 (FC) Surface Mount Package Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 October 2021 is a registered trademark of Richtek Technology Corporation. www.richtek.com 29 RT6253A/B Footprint Information Package TSOT-26/TSOT-26(FC)/SOT-26/SOT26(COL) Copyright © 2021 Richtek Technology Corporation. All rights reserved. www.richtek.com 30 Footprint Dimension (mm) Number of Pin P1 A B C D M 6 0.95 3.60 1.60 1.00 0.70 2.60 Tolerance ±0.10 is a registered trademark of Richtek Technology Corporation. DS6253A/B-03 October 2021 RT6253A/B Package Footprint Dimension (mm) Number of SOT-563(FC) Tolerance Pin P1 A B C D M 6 0.50 2.42 1.02 0.70 0.30 1.30 ±0.10 Richtek Technology Corporation 14F, No. 8, Tai Yuen 1st Street, Chupei City Hsinchu, Taiwan, R.O.C. Tel: (8863)5526789 Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries. Copyright © 2021 Richtek Technology Corporation. All rights reserved. DS6253A/B-03 October 2021 is a registered trademark of Richtek Technology Corporation. www.richtek.com 31
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RT6253BHGJ6F
  •  国内价格
  • 1+1.22100
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  • 1500+0.81510
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RT6253BHGJ6F
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    RT6253BHGJ6F
    •  国内价格
    • 1+1.52630
    • 1000+1.52630
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    库存:1