RT6253A/B
17V Input, 3A, ACOT® Buck Converter with Both FETs OC
Protection
General Description
Features
The RT6253A/B is a simple, easy-to-use, 3A
synchronous step-down DC-DC converter with an input
supply voltage range of 4.5V to 17V. The device
⚫
⚫
Integrated 95m and 50m FETs
Continuous Current
TSOT-23-6
possesses an accurate reference voltage and
integrates low RDS(ON) power MOSFETs to achieve
high efficiency.
(FC) : 3A
SOT-563 (FC) : 2.5A
Input Supply Voltage Range : 4.5V to 17V
Output Voltage Range : 0.765V to 7V
®
⚫ Advanced Constant On-Time (ACOT ) Control
Ultrafast Transient Response
Optimized
for Low-ESR Ceramic Output
Capacitors
⚫ High Accuracy Feedback Reference Voltage : 1%
⚫ Optional for Operation Modes :
⚫
⚫
The RT6253A/B adopts Advanced Constant On-Time
(ACOT® ) control architecture to provide an ultrafast
transient response with few external components and to
operate in nearly constant switching frequency over the
line, load, and output voltage range. The RT6253A
operates in automatic PSM that maintains high
efficiency during light load operation. The RT6253B
operates in Forced PWM that helps meet tight voltage
regulation accuracy requirements.
⚫
The RT6253A/B senses both FETs current for a robust
over-current protection (OCP). The device features
cycle-by-cycle current limit protection to prevent the
device from the catastrophic damage in output short
circuit, over-current or inductor saturation conditions. A
built-in soft-start function prevents inrush current during
start-up. The device also includes input under-voltage
lockout, output under-voltage protection, and overtemperature protection (OTP) to provide safe and
⚫
⚫
⚫
⚫
RT6253A : Power Saving Mode (PSM)
RT6253B : Forced PWM Mode
Fixed Switching Frequency : 580kHz
Enable Control and Fixed Soft-Start with typ. 1ms
Safe Start-Up from Pre-biased Output
Input Under-Voltage Lockout (UVLO)
Protection Function
Output Under-Voltage Protection (UVP) with
Hiccup Mode
High- / Low-side MOSFET OCP and OTP
Function
smooth operation in all operating conditions.
⚫
RoHS Compliant and Halogen Free
Simplified Application Circuit
RT6253A/B
VIN
VIN
RBOOT
(Optional)
BOOT
CBOOT
CIN
L
VOUT
SW
Enable
EN
RFB1
GND
CFF
COUT
FB
RFB2
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DS6253A/B-03
October
2021
is a registered trademark of Richtek Technology Corporation.
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1
RT6253A/B
Ordering Information
Applications
Set-Top Boxes
⚫ LCD TVs
⚫
RT6253A/B
Package Type
J6F : TSOT-23-6 (FC)
H6F : SOT-563 (FC)
Lead Plating System
G : Green (Halogen Free and Pb Free)
UVP Option
H : Hiccup
PWM Operation Mode
A : Automatic PSM
B : Forced PWM
Home Networking Devices
Surveillance
⚫ General Purpose
⚫
⚫
Pin Configuration
(TOP VIEW)
BOOT
EN
FB
6
5
4
2
3
SW
VIN
Note :
Richtek products are :
RoHS compliant and compatible with the current
GND
requirements of IPC/JEDEC J-STD-020.
Suitable for use in SnPb or Pb-free soldering processes.
Marking Information
TSOT-23-6 (FC)
FB EN BOOT
6
5
4
1
2
3
RT6253AHGJ6F
36=DNN
36= : Product Code
DNN : Date Code
VIN SW GND
SOT-563 (FC)
RT6253BHGJ6F
35=DNN
35= : Product Code
DNN : Date Code
RT6253AHGH6F
05W
05 : Product Code
W : Date Code
RT6253BHGH6F
04W
04 : Product Code
W : Date Code
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DS6253A/B-03
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2021
RT6253A/B
Functional Pin Description
Pin No.
Pin
Name
TSOT-23-6 (FC) SOT-563 (FC)
Pin Function
1
3
GND
Power ground.
2
2
SW
Switch node between the internal switch and the synchronous
rectifier. Connect this pin to the inductor and bootstrap capacitor.
3
1
VIN
Power input. The input voltage range is from 4.5V to 17V. Connect
input bypass capacitors directly to this pin and GND pins. The MLCC
with capacitance higher than 20F is recommended.
4
6
FB
Feedback voltage input. Connect this pin to the midpoint of the
external feedback resistive divider to set the output voltage of the
converter to the desired regulation level. The device regulates the
FB voltage at feedback reference voltage.
5
5
EN
Enable control input. Connect this pin to logic high enables the
device and connect this pin to GND disables the device.
6
4
BOOT
Bootstrap capacitor connection node to supply the high-side gate
driver. Connect a 0.1F ceramic capacitor between this pin and the
SW pin.
Functional Block Diagram
VIN
SW
EN
+
-
VEN_TH
UVLO
Internal
Regulator
VCC
PVCC
OnTime
REN_DN
BOOT
OC
UV
Protection
65%
Control
Soft-Start
Ramp
Gen.
+
+
-
FB
Gate Driver &
Dead-Time
Control
SW
Comparator
MIN OFF
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DS6253A/B-03
October
2021
GND
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RT6253A/B
Operation
The RT6253A/B is a high-efficiency, synchronous stepdown DC-DC converter that can deliver up to 3A output
current from a 4.5V to 17V input supply.
Advanced Constant On-Time Control and PWM
Operation
The RT6253A/B adopts ACOT® control for its ultrafast
transient response, low external component counts and
stable with low ESR MLCC output capacitors. When the
feedback voltage falls below the feedback reference
voltage, the minimum off-time one-shot (200ns, typ.)
has timed out and the inductor current is below the
current limit threshold, then the internal on-time oneshot circuitry is triggered and the high-side switch is
turn-on. Since the minimum off-time is short, the device
exhibits ultrafast transient response and enables the
use of smaller output capacitance.
The on-time is inversely proportional to input voltage
and directly proportional to output voltage to achieve
pseudo-fixed frequency over the input voltage range.
After the on-time one-shot timer expired, the high-side
switch is turned off and the low-side switch is turned on
until the on-time one-shot is triggered again. To achieve
stable operation with low-ESR ceramic output
capacitors, an internal ramp signal is added to the
feedback reference voltage to simulate the output
voltage ripple.
held below a logic-low threshold voltage (VEN_L) of the
enable input (EN), the converter will disable output
voltage, that is, the converter is disabled and switching
is inhibited even if the VIN voltage is above VIN undervoltage lockout threshold (VUVLO). During shutdown
mode, the supply current can be reduced to ISHDN
(10A or below). If the EN voltage rises above the logichigh threshold voltage (VEN_H) while the VIN voltage is
higher than UVLO threshold, the device will be turned
on, that is, switching being enabled and soft-start
sequence being initiated. An internal resistor REN_DN
from EN to GND allows EN float to shutdown the chip.
Soft-Start (SS)
The RT6253A/B provides an internal soft-start feature
for inrush control, and the output voltage starts to rise in
0.3ms from EN rising edge. At power up, the internal
capacitor is charged by an internal current source to
generate a soft-start ramp voltage as a reference
voltage to the PWM comparator. The device will initiate
switching and the output voltage will smoothly ramp up
to its targeted regulation voltage only after this ramp
voltage is greater than the feedback voltage VFB to
ensure the converter has a smooth start-up from prebiased output.
VIN = 12V
VIN
VCC = 5V
Power Saving Mode (RT6253A Only)
The RT6253A automatically enters power saving mode
(PSM) at light load to maintain high efficiency. As the
load current decreases, the inductor current ripple
valley eventually touches the zero current, which is the
VCC
EN
0.3ms
tSS
VOUT
boundary between continuous conduction and
discontinuous conduction modes. The low-side switch
is turned off when the zero inductor current is detected.
In this case, the output capacitor is only discharged by
load current so that the switching frequency decreases.
As the result, the light-load efficiency can be enhanced
due to lower switching loss.
Enable Control
The RT6253A/B provides an EN pin, as an external chip
enable control, to enable or disable the device. If VEN is
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Input Under-Voltage Lockout
In addition to the EN pin, the RT6253A/B also provides
enable control through the VIN pin. It features an undervoltage lockout (UVLO) function that monitors the
internal linear regulator (VCC). If VEN rises above VEN_H
first, switching will still be inhibited until the VIN voltage
rises above VUVLO. It is to ensure that the internal
regulator is ready so that operation with not-fullyenhanced internal MOSFET switches can be prevented.
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2021
RT6253A/B
After the device is powered up, if the input voltage VIN
goes below the UVLO falling threshold voltage (VUVLO
− VUVLO), this switching will be inhibited; if VIN rises
above the UVLO rising threshold (VUVLO), the device
will resume normal operation with a complete soft-start.
Output Under-Voltage Protection and Hiccup Mode
The RT6253A/B includes output under-voltage
protection (UVP) against over-load or short-circuited
condition by constantly monitoring the feedback voltage
VFB. If VFB drops below the under-voltage protection trip
threshold (typically 65% of the internal feedback
reference voltage), the UV comparator will go high to
turn off both the internal high-side and low-side
MOSFET switches.
If the output under-voltage condition continues for a
period of time, the RT6253A/B will enter output undervoltage protection with hiccup mode. During hiccup
mode, the IC will shut down for tHICCUP_OFF (15ms), and
then attempt to recover automatically for tHICCUP_ON
(1.8ms). Upon completion of the soft-start sequence, if
the fault condition is removed, the converter will resume
normal operation; otherwise, such cycle for autorecovery will be repeated until the fault condition is
cleared. The hiccup mode allows the circuit to operate
safely with low input current and power dissipation, and
then the converter resumes normal operation as soon
as the over-load or short-circuit condition is removed.
protection on both the high-side and low-side MOSFETs
and prevents the device from the catastrophic damage
in output short-circuit, over-current or inductor
saturation conditions.
The high-side MOSFET over-current protection is
achieved by an internal current comparator that
monitors the current in the high-side MOSFET during
each on-time. The switch current is compared with the
high-side switch peak-current limit (ILIM_H) after a
certain amount of delay when the high-side switch being
turned on each cycle. If an over-current condition occurs,
the converter will immediately turns off the high-side
switch and turns on the low-side switch to prevent the
inductor current exceeding the high-side current limit.
The low-side MOSFET over-current protection is
achieved by measuring the inductor current through the
synchronous rectifier (low-side switch) during the lowside on-time. Once the current rises above the low-side
switch valley current limit (ILIM_L), the on-time one-shot
will be inhibited until the inductor current ramps down to
the current limit level (ILIM_L), that is, another on-time
can only be triggered when the inductor current goes
below the low-side current limit. If the output load
current exceeds the available inductor current (clamped
by the low-side current limit), the output capacitor needs
to supply the extra current such that the output voltage
will begin to drop. If it drops below the output undervoltage protection trip threshold, the IC will stop
switching to avoid excessive heat.
Output short
VOUT, 2V/Div
Negative Over-Current Limit
Fault condition removed
Resume normal operation
The RT6253B is the part which is forced to PWM and
allows negative current operation.
ISW , 4A/Div
In case of PWM operation, high negative current may
be generated as an external power source is tied to
VSW , 10/Div
output terminal unexpectedly. As the risk described
above, the internal circuit monitors negative current in
each on-time interval of low-side MOSFET and
compares it with NOC threshold.
10ms/Div
Once the negative current exceeds the NOC threshold,
the low-side MOSFET is turned off immediately, and
The Over-Current Protection
The RT6253A/B features cycle-by-cycle current-limit
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DS6253A/B-03
October
2021
then the high-side MOSFET will be turned on to
discharge the energy of output inductor. This behavior
can keep the valley of negative current at NOC
threshold to protect low-side MOSFET. However, the
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RT6253A/B
negative current can’t be limited at NOC threshold
anymore since minimum off-time is reached.
Thermal Shutdown
The RT6253A/B includes an over-temperature
protection (OTP) circuitry to prevent overheating due to
excessive power dissipation. The OTP will shut down
switching operation when junction temperature exceeds
a thermal shutdown threshold (TSD). Once the junction
temperature cools down by a thermal shutdown
hysteresis (TSD), the IC will resume normal operation
with a complete soft-start.
Note that the over temperature protection is intended to
protect the device during momentary overload
conditions. The protection is activated outside of the
absolute maximum range of operation as a secondary
fail-safe and therefore should not be relied upon
operationally. Continuous operation above the specified
absolute maximum operating junction temperature may
impair the reliability of the device or permanently
damage the device.
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DS6253A/B-03
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RT6253A/B
Absolute Maximum Ratings
(Note 1)
⚫
Supply Input Voltage, VIN ------------------------------------------------------------------------------ −0.3V to 20V
⚫
Enable Voltage, EN -------------------------------------------------------------------------------------- −0.3V to 20V
⚫
Switch Voltage, SW -------------------------------------------------------------------------------------- −0.3V to 20.3V
< 100ns ----------------------------------------------------------------------------------------------------- −5V to 25V
⚫
BOOT Voltage , BOOT---------------------------------------------------------------------------------- – 0.3V to 26V
⚫
BOOT to SW, VBOOT − VSW --------------------------------------------------------------------------- −0.3V to 6V
⚫
Feedback Voltage, FB ---------------------------------------------------------------------------------- −0.3V to 6V
⚫
Lead Temperature (Soldering, 10 sec.) ------------------------------------------------------------- 260C
⚫
Junction Temperature ----------------------------------------------------------------------------------- 150C
⚫
Storage Temperature Range -------------------------------------------------------------------------- −65C to 150C
⚫
Power Dissipation, PD @ TA = 25C
TSOT-23-6 (FC) ------------------------------------------------------------------------------------------ 1.64W
SOT-563 (FC) --------------------------------------------------------------------------------------------- 1.25W
ESD Ratings
⚫
ESD Susceptibility
(Note 2)
HBM (Human Body Model) ---------------------------------------------------------------------------- 2kV
Recommended Operating Conditions
(Note 3)
⚫
Supply Input Voltage ------------------------------------------------------------------------------------ 4.5V to 17V
⚫
Junction Temperature Range ------------------------------------------------------------------------- −40C to 125C
Thermal Information
(Note 4 and Note 5)
Thermal Parameter
TSOT-23-6 (FC)
SOT-563 (FC)
Unit
JA
Junction-to-ambient thermal resistance
(JEDEC standard)
88.7
104.3
C/W
JC(Top)
Junction-to-case (top) thermal resistance
76.9
62.1
C/W
JC(Bottom)
Junction-to-case (bottom) thermal resistance
6
8.4
C/W
JA(EVB)
Junction-to-ambient thermal resistance
(specific EVB)
61
80.6
C/W
JC(Top)
Junction-to-top characterization parameter
13.9
6.7
C/W
JB
Junction-to-board characterization
parameter
31.53
45.17
C/W
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October
2021
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RT6253A/B
Electrical Characteristics
TSOT-23-6 (FC)
(VIN = 12V, TA = 25C, unless otherwise specified )
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Supply Voltage
VIN Supply Input Operating
Voltage
VIN
4.5
--
17
V
Under-Voltage Lockout
Threshold
VUVLO
3.7
4
4.3
V
Under-Voltage Lockout
Threshold Hysteresis
VUVLO
--
400
--
mV
Shutdown Current
ISHDN
VEN = 0V
--
--
4
µA
Quiescent Current
IQ
VEN = 2V, VFB = 0.8V
--
280
--
µA
--
1
--
ms
Soft-Start
Soft-Start Time
tSS
Enable Voltage
Enable Voltage Threshold
EN Pin Pull-Down
Resistance
VENH
EN high-level input voltage
1.16
1.25
1.34
VENL
EN low-level input voltage
1.01
1.1
1.19
REN_DN
EN pin resistance to GND, VEN = 12V
225
450
900
k
V
Feedback Voltage and Discharge Resistance
Feedback Threshold Voltage VFB
VOUT = 1.05V
758
765
772
mV
Feedback Input Current
IFB
VFB = 0.8V, TA = 25°C
−0.1
0
0.1
A
High-Side On-Resistance
RDS(ON)_H
VBOOT – VSW = 4.8V
--
95
--
Low-Side On-Resistance
RDS(ON)_L
--
50
--
High-Side Switch Current
Limit
ILIM_H
--
5.6
--
Low-Side Switch Valley
Current Limit
ILIM_L
Internal MOSFET
mΩ
Current Limit
A
3.2
4.2
5.2
--
580
--
kHz
--
60
--
ns
VFB = 0.5V
--
200
260
ns
Hiccup detect
--
65
--
%
Switching Frequency
Switching Frequency
f SW
VOUT = 1.05V, PWM mode
On-Time Timer Control
Minimum On-Time
tON_MIN
Minimum Off-Time
tOFF_MIN
Output Under-Voltage Protections
UVP Trip Threshold
VUVP
Hiccup Power On-Time
tHICCUP_ON
--
1.8
--
Hiccup Power Off-Time
tHICCUP_OFF
--
15
--
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2021
RT6253A/B
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Thermal Shutdown
Thermal Shutdown
Threshold
TSD
--
155
--
Thermal Shutdown
Hysteresis
TSD
--
35
--
°C
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October
2021
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9
RT6253A/B
SOT-563 (FC)
(VIN = 12V, TA = 25C, unless otherwise specified)
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Supply Voltage
VIN Supply Input Operating
Voltage
VIN
4.5
--
17
V
Under-Voltage Lockout
Threshold
VUVLO
3.9
4.2
4.5
V
Under-Voltage Lockout
Threshold Hysteresis
VUVLO
--
420
--
mV
Shutdown Current
ISHDN
VEN = 0V
--
--
4
µA
Quiescent Current
IQ
VEN = 2V, VFB = 0.85V
--
295
--
µA
--
0.95
--
ms
Soft-Start
Soft-Start Time
tSS
Enable Voltage
Enable Voltage Threshold
EN Pin Pull-Down
Resistance
VENH
EN high-level input voltage
1.24
1.31
1.38
VENL
EN low-level input voltage
1.09
1.16
1.23
REN_DN
EN pin resistance to GND, VEN = 12V
225
450
900
k
V
Feedback Voltage and Discharge Resistance
Feedback Threshold Voltage VFB
VOUT = 1.05V
799
807
815
mV
Feedback Input Current
IFB
VFB = 0.85V, TA = 25°C
−0.1
0
0.1
A
High-Side On-Resistance
RDS(ON)_H
VBOOT – VSW = 4.8V
--
95
--
Low-Side On-Resistance
RDS(ON)_L
--
50
--
High-Side Switch Current
Limit
ILIM_H
--
5.6
--
Low-Side Switch Valley
Current Limit
ILIM_L
3.45
4.4
5.35
--
580
--
kHz
--
60
--
ns
VFB = 0.5V
--
190
250
ns
Hiccup detect
--
65
--
%
Internal MOSFET
mΩ
Current Limit
A
Switching Frequency
Switching Frequency
f SW
VOUT = 1.05V, PWM mode
On-Time Timer Control
Minimum On-Time
tON_MIN
Minimum Off-Time
tOFF_MIN
Output Under-Voltage Protections
UVP Trip Threshold
VUVP
Hiccup Power On-Time
tHICCUP_ON
--
1.8
--
Hiccup Power Off-Time
tHICCUP_OFF
--
15
--
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DS6253A/B-03
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RT6253A/B
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
Thermal Shutdown
Thermal Shutdown
Threshold
TSD
--
155
--
Thermal Shutdown
Hysteresis
TSD
--
35
--
°C
Note 1. Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are
stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the
operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may affect device
reliability.
Note 2. Devices are ESD sensitive. Handling precaution recommended.
Note 3. The device is not guaranteed to function outside its operating conditions.
Note 4. θJA and θJC are measured or simulated at TA = 25C based on the JEDEC 51-7 standard.
Note 5. θJA(EVB), ΨJC(TOP) and ΨJB are measured on a high effective-thermal-conductivity four-layer test board which is in size of
70mm x 50mm; furthermore, all layers with 1 oz. Cu. Thermal resistance/parameter values may vary depending on the
PCB material, layout, and test environmental conditions.
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
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October
2021
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RT6253A/B
Typical Application Circuit
RT6253A/B
VIN
CIN
10μF x 2
CIN
0.1μF
Enable
VIN
BOOT
RBOOT
(Optional)
CBOOT
0.1μF
L
VOUT
SW
EN
RFB1
GND
CFF
COUT
FB
RFB2
Table 1. Recommended Components Selection
VOUT (V)
RFB1 (k)
RFB2 (k)
CFF (pF)
L (H)
COUT (F)
52.3
10
10 to 100
3.3 to 4.7
20 to 68
33.2
30.9
10
10 to 100
2.2 to 4.7
20 to 68
2.5
22.6
21
10
10 to 100
2.2 to 4.7
20 to 68
1.8
13.7
12.4
10
10 to 100
1.5 to 4.7
20 to 68
1.5
9.53
8.66
10
--
1.5 to 4.7
20 to 68
1.2
5.76
4.87
10
--
1.5 to 4.7
20 to 68
1.0
3.09
2.4
10
--
1.5 to 4.7
20 to 68
TSOT-23-6 (FC)
SOT-563 (FC)
5.0
54.9
3.3
Note :
(1) Please do not use a CFF higher than 100pF due to the noise coupling consideration.
(2) Considering effective capacitance de-rating which is related to biased voltage level and size, the effective
capacitance of COUT at target output level should meet the value in above table to make converter operated in
stable and normal.
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RT6253A/B
Typical Operating Characteristics
L : WE-74404054022 (DCR = 19m) for VOUT = 1V and 1.8V.
L : WE-74404054047 (DCR = 30m) for VOUT = 3.3V and 5V.
Efficiency vs. Output Current
Efficiency vs. Output Current
100
100
90
90
80
70
VOUT = 3.3V
60
VOUT = 1.8V
Efficiency (%)
Efficiency (%)
80
VOUT = 1V
50
40
30
VOUT = 5V
60
VOUT = 3.3V
50
VOUT = 1.8V
40
VOUT = 1V
30
20
20
10
0
0.001
70
10
RT6253AHGJ6F, VIN = 5V
0.01
0.1
1
0
0.001
10
RT6253AHGJ6F, VIN = 12V
0.01
Efficiency vs. Output Current
100
90
90
80
70
VOUT = 3.3V
60
VOUT = 1.8V
50
VOUT = 1V
Efficiency (%)
Efficiency (%)
80
40
30
20
70
VOUT = 5V
60
VOUT = 3.3V
50
VOUT = 1.8V
40
VOUT = 1V
30
20
10
10
RT6253BHGJ6F, VIN = 5V
0.01
0.1
1
0
0.001
10
RT6253BHGJ6F, VIN = 12V
0.01
Efficiency vs. Output Current
1
10
Efficiency vs. Output Current
100
100
90
90
80
80
VOUT = 1.8V
70
Efficiency (%)
Efficiency (%)
0.1
Output Current (A)
Output Current (A)
VOUT = 1V
60
50
40
30
20
70
VOUT = 5V
60
VOUT = 3.3V
50
VOUT = 1.8V
40
VOUT = 1V
30
20
10
10
RT6253AHGH6F, VIN = 5V
0.01
0.1
1
Output Current (A)
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DS6253A/B-03
10
Efficiency vs. Output Current
100
0
0.001
1
Output Current (A)
Output Current (A)
0
0.001
0.1
October
2021
10
0
0.001
RT6253AHGH6F, VIN = 12V
0.01
0.1
1
10
Output Current (A)
is a registered trademark of Richtek Technology Corporation.
www.richtek.com
13
RT6253A/B
Efficiency vs. Output Current
100
90
90
80
80
70
VOUT = 1.8V
60
VOUT = 1V
Efficiency (%)
Efficiency (%)
Efficiency vs. Output Current
100
50
40
70
VOUT = 1.8V
40
VOUT = 1V
30
20
20
10
RT6253BHGH6F, VIN = 5V
0
0.001
0.01
0.1
1
VOUT = 3.3V
50
30
10
VOUT = 5V
60
RT6253BHGH6F, VIN = 12V
0
0.001
10
0.01
0.1
1
10
Output Current (A)
Output Current (A)
Output Voltage vs. Output Current
Output Voltage vs. Output Current
5.50
1.20
1.10
Output Voltage (V)
Output Voltage (V)
1.15
1.05
1.00
RT6253AHGJ6F, VIN = 5V
0.95
RT6253BHGJ6F, VIN = 5V
0.90
RT6253AHGJ6F, VIN = 12V
5.25
5.00
RT6253AHGJ6F, VIN = 9V
RT6253BHGJ6F, VIN = 9V
RT6253AHGJ6F, VIN = 12V
4.75
RT6253BHGJ6F, VIN = 12V
RT6253BHGJ6F, VIN = 12V
0.85
VOUT = 5V
VOUT = 1V
4.50
0.80
0
0.5
1
1.5
2
2.5
0
3
0.5
1
1.5
2
2.5
3
Output Current (A)
Output Current (A)
Output Voltage vs. Output Current
Output Voltage vs. Output Current
1.20
5.50
1.10
Output Voltage (V)
Output Voltage (V)
1.15
1.05
1.00
RT6253AHGH6F, VIN = 5V
0.95
RT6253BHGH6F, VIN = 5V
0.90
RT6253AHGH6F, VIN = 12V
0.85
RT6253BHGH6F, VIN = 12V
5.25
5.00
RT6253AHGH6F, VIN = 9V
RT6253BHGH6F, VIN = 9V
4.75
RT6253AHGH6F, VIN = 12V
RT6253BHGH6F, VIN = 12V
VOUT = 5V
VOUT = 1V
4.50
0.80
0
0.5
1
1.5
2
2.5
Output Current (A)
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14
3
0
0.5
1
1.5
2
2.5
3
Output Current (A)
is a registered trademark of Richtek Technology Corporation.
DS6253A/B-03
October
2021
RT6253A/B
Output Voltage vs. Input Voltage
Output Voltage vs. Input Voltage
1.10
5.5
5.3
1.05
1.00
Output Voltage (V)
Output Voltage (V)
5.4
RT6253AHGJ6F
RT6253BHGJ6F
0.95
5.2
5.1
5.0
RT6253AHGJ6F
RT6253BHGJ6F
4.9
4.8
4.7
4.6
VOUT = 1V
VOUT = 5V
4.5
0.90
5
6
7
8
9
9
10 11 12 13 14 15 16 17
10
11
12
13
14
15
16
17
Input Voltage (V)
Input Voltage (V)
Output Voltage vs. Input Voltage
Output Voltage vs. Input Voltage
5.5
1.10
5.4
1.00
Output Voltage (V)
Output Voltage (V)
5.3
1.05
RT6253AHGH6F
RT6253BHGH6F
0.95
5.2
5.1
5.0
RT6253AHGH6F
RT6253BHGH6F
4.9
4.8
4.7
4.6
VOUT = 1V
VOUT = 5V
4.5
0.90
5
6
7
8
9
9
10 11 12 13 14 15 16 17
10
11
12
13
14
Quiescent Current vs. Temperature
16
17
Shutdown Current vs. Temperature
340
1.0
RT6253AHGJ6F
RT6253AHGJ6F
VIN = 17V
Shutdown Current (μA)1
320
Quiescent Current (μA)
15
Input Voltage (V)
Input Voltage (V)
300
280
260
VIN = 17V
240
VIN = 12V
VIN = 9V
220
VIN = 12V
0.5
VIN = 9V
VIN = 5V
0.0
-0.5
VIN = 5V
-1.0
200
-50
-25
0
25
50
75
100
Temperature (°C)
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DS6253A/B-03
October
2021
125
-50
-25
0
25
50
75
100
125
Temperature (°C)
is a registered trademark of Richtek Technology Corporation.
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15
RT6253A/B
Quiescent Current vs. Temperature
Shutdown Current vs. Temperature
340
1.0
RT6253AHGH6F
RT6253AHGH6F
VIN = 17V
Shutdown Current (μA)1
Quiescent Current (μA)
320
300
280
VIN = 17V
260
VIN = 12V
240
VIN = 9V
VIN = 5V
220
VIN = 12V
0.5
VIN = 9V
VIN = 5V
0.0
-0.5
-1.0
200
-50
-25
0
25
50
75
100
-50
125
-25
0
0.80
0.84
0.79
0.83
0.78
0.77
0.76
0.75
0.74
0.73
-25
0
25
50
100
125
0.81
0.80
0.79
0.78
0.77
0.75
-50
75
0.82
0.76
RT6253AHGJ6F
0.71
75
100
-50
125
-25
0
25
50
RT6253AHGH6
F
75
100
125
Temperature (°C)
Temperature (°C)
Frequency vs. Input Voltage
Frequency vs. Output Current
700
600
500
Frequency (kHz)1
650
Frequency (kHz)1
50
Reference Voltage vs. Temperature
0.85
Reference Voltage (V)
Reference Voltage (V)
Reference Voltage vs. Temperature
0.81
0.72
25
Temperature (°C)
Temperature (°C)
600
550
500
400
300
200
100
450
5
6
7
8
9
10 11 12 13 14 15 16 17
Input Voltage (V)
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16
0
0.001
0.01
0.1
1
10
Output Current (A)
is a registered trademark of Richtek Technology Corporation.
DS6253A/B-03
October
2021
RT6253A/B
Frequency vs. Temperature
EN Threshold vs. Temperature
700
1.4
650
1.3
EN Threshold (V)
Frequency (kHz)1
EN VIH
600
550
1.2
EN VIL
1.1
500
1.0
450
0.9
RT6253AHGJ6F
-50
-25
0
25
50
75
100
-50
125
-25
0
25
EN Threshold vs. Temperature
100
125
4.1
4.0
EN VIH
UVLO_H
3.9
1.3
Input Voltage (V)
EN Threshold (V)
75
UVLO vs. Temperature
1.4
1.2
EN VIL
1.1
3.8
3.7
UVLO_L
3.6
3.5
3.4
3.3
3.2
RT6253HGH6F
RT6253AHGJ6F
3.1
1.0
-50
-25
0
25
50
75
100
-50
125
UVLO vs. Temperature
0
25
50
75
100
125
Power On from EN
4.4
4.3
VOUT
(5V/Div)
UVLO_H
4.2
-25
Temperature (°C)
Temperature (°C)
Input Voltage (V)
50
Temperature (°C)
Temperature (°C)
4.1
VEN
(2V/Div)
4.0
3.9
VSW
(10V/Div)
UVLO_L
3.8
3.7
3.6
3.5
RT6253AHGH6F
3.4
-50
-25
0
25
50
75
100
Temperature (°C)
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DS6253A/B-03
October
2021
125
IL
(2A/Div)
VIN = 12V, VOUT = 5V
IOUT = 3A, L = 4.7H
Time (1ms/Div)
is a registered trademark of Richtek Technology Corporation.
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RT6253A/B
Power Off from EN
VOUT
(5V/Div)
VOUT
(5V/Div)
VEN
(2V/Div)
VIN
(10V/Div)
VSW
(10V/Div)
IL
(2A/Div)
Power On from VIN
VSW
(10V/Div)
VIN = 12V, VOUT = 5V
IOUT = 3A, L = 4.7H
IL
(2A/Div)
Time (5ms/Div)
Time (200s/Div)
Output Ripple as IOUT = 10mA
Power Off from VIN
VOUT
(5V/Div)
VOUT
(50mV/Div)
VIN
(10V/Div)
VSW
(10V/Div)
IL
(2A/Div)
VSW
(5V/Div)
VIN = 12V, VOUT = 5V
IOUT = 3A, L = 4.7H
IL
(2A/Div)
VIN = 12V, VOUT = 5V, IOUT = 10mA, L = 4.7H
Time (5ms/Div)
Time (10s/Div)
Output Ripple as IOUT = 3A
Load Transient (No Load to Full Load)
VOUT
(20mV/Div)
VOUT
(100mV/Div)
VIN = 12V, VOUT = 5V
IOUT = 0A to 3A, L = 4.7H
VSW
(5V/Div)
IL
(2A/Div)
VIN = 12V, VOUT = 5V, IOUT = 3A, L = 4.7H
Time (1s/Div)
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18
VIN = 12V, VOUT = 5V
IOUT = 3A, L = 4.7H
IOUT
(1A/Div)
Time (400s/Div)
is a registered trademark of Richtek Technology Corporation.
DS6253A/B-03
October
2021
RT6253A/B
Over-Current Protection and UVP
Load Transient (Half Load to Full Load)
VOUT
(2V/Div)
VOUT
(100mV/Div)
VIN = 12V, VOUT = 5V
IOUT = 1A to 3A, L = 4.7H
IOUT
(1A/Div)
VSW
(10V/Div)
IL
(2A/Div)
Time (400s/Div)
Time (20s/Div)
Short Circuit Protection
Short Circuit before Power On
VOUT
(2V/Div)
VIN = 12V, VOUT = 5V, L = 4.7H
VOUT
(2V/Div)
VIN = 12V, VOUT = 5V, L = 4.7H
VIN
(10V/Div)
VSW
(10V/Div)
VSW
(10V/Div)
IL
(2A/Div)
IL
(2A/Div)
Time (10ms/Div)
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DS6253A/B-03
VIN = 12V, VOUT = 5V
L = 4.7H
October
2021
Time (10ms/Div)
is a registered trademark of Richtek Technology Corporation.
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19
RT6253A/B
Application Information
The output stage of a synchronous buck converter is
composed of an inductor and capacitor, which stores
and delivers energy to the load, and forms a secondorder low-pass filter to smooth out the switch node
voltage to maintain a regulated output voltage.
the calculated inductance value is :
L=
1.2 (12 − 1.2)
= 2.48μH
12 580kHz 0.75A
For the typical application, a standard inductance value
of 2.2H can be selected.
Inductor Selection
1.2 (12 − 1.2)
= 0.85A (28.3% of the IC rated current)
12 580kHz 2.2μH
The inductor selection trade-offs among size, cost,
efficiency, and transient response requirements.
IL =
Generally, three key inductor parameters are specified
for operation with the device: inductance value (L),
and IL(PEAK) = 3A + 0.85A = 3.425A
2
inductor saturation current (ISAT), and DC resistance
For the 2.2H value, the inductor's saturation and
thermal rating should exceed at least 3.425A. For more
conservative, the rating for inductor saturation current
must be equal to or greater than switch current limit of
the device rather than the inductor peak current.
(DCR).
A good compromise between size and loss is to choose
the peak-to-peak ripple current equals to 20% to 50% of
the IC rated current. The switching frequency, input
voltage, output voltage, and selected inductor ripple
current determines the inductor value as follows :
L=
VOUT ( VIN − VOUT )
VIN fSW IL
Once an inductor value is chosen, the ripple current (IL)
is calculated to determine the required peak inductor
current.
IL =
VOUT ( VIN − VOUT )
I
and IL(PEAK) = IOUT(MAX) + L
VIN fSW L
2
IL(PEAK) should not exceed the minimum value of IC's
upper current limit level. Besides, the current flowing
through the inductor is the inductor ripple current plus the
output current. During power up, faults, or transient load
conditions, the inductor current can increase above the
calculated peak inductor current level calculated above.
For EMI sensitive application, choosing shielding type
inductor is preferred.
Input Capacitor Selection
Input capacitance, CIN, is needed to filter the pulsating
current at the drain of the high-side power MOSFET.
CIN should be sized to do this without causing a large
variation in input voltage. The waveform of CIN ripple
voltage and ripple current are shown in Figure 1. The
peak-to-peak voltage ripple on input capacitor can be
estimated as the equation below :
VCIN = D IOUT 1− D + IOUT ESR
CIN fSW
where
D=
VOUT
VIN
In transient conditions, the inductor current can increase
For ceramic capacitors, the equivalent series resistance
up to the switch current limit of the device. For this reason,
(ESR) is very low, the ripple which is caused by ESR
can be ignored, and the minimum input capacitance can
the most conservative approach is to specify an inductor
with a saturation current rating which is equal to or greater
than the switch current limit rather than the peak inductor
current.
Considering the Typical Application Circuit for 1.2V
output at 3A and an input voltage of 12V, using an
inductor ripple of 0.75A (25% of the IC rated current),
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
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20
be estimated as the equation below :
CIN_MIN = IOUT_MAX
D (1− D )
VCIN_MAX fSW
where VCIN_MAX 200mV
is a registered trademark of Richtek Technology Corporation.
DS6253A/B-03
October
2021
RT6253A/B
capacitor should be 0402 or 0603 in size.
VCIN
CIN Ripple Voltage
VESR = IOUT x ESR
(1-D) x IOUT
CIN Ripple Current
D x IOUT
D x tSW (1-D) x tSW
Figure 1. CIN Ripple Voltage and Ripple Current
In addition, the input capacitor needs to have a very low
ESR and must be rated to handle the worst-case RMS
input current of :
IRMS IOUT_MAX
VOUT
VIN
VIN
−1
VOUT
It is common to use the worse IRMS IOUT/2 at VIN =
2VOUT for design. Note that ripple current ratings from
capacitor manufacturers are often based on only 2000
hours of life which makes it advisable to further de-rate
the capacitor, or choose a capacitor rated at a higher
temperature than required.
Several capacitors may also be paralleled to meet size,
Output Capacitor Selection
The RT6253A/B are optimized for ceramic output
capacitors and best performance will be obtained using
them. The total output capacitance value is usually
determined by the desired output voltage ripple level and
transient response requirements for sag (undershoot on
load apply) and soar (overshoot on load release).
Output Ripple
The output voltage ripple at the switching frequency is
a function of the inductor current ripple going through
the output capacitor’s impedance. To derive the output
voltage ripple, the output capacitor with Capacitance
(COUT) and its equivalent series resistance (RESR) must
be taken into consideration. The output peak-to-peak
ripple voltage (VRIPPLE) caused by the inductor current
ripple (IL) is characterized by two components, which
are ESR ripple (VRIPPLE(ESR)) and capacitive ripple
(VRIPPLE(C)) and can be expressed as below :
VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C)
VRIPPLE(ESR) = IL RESR
IL
8 COUT fSW
height and thermal requirements in the design. For low
input voltage applications, sufficient bulk input
VRIPPLE(C) =
capacitance is needed to minimize transient effects
during output load changes.
As ceramic capacitors are used, both parameters
should be estimated due to the extremely low ESR and
relatively small capacitance. Refer to the RT6253A/B's
typical application circuit of 1.2V application, the actual
inductor current ripple (IL) is 0.85A, and the output
Ceramic capacitors are ideal for switching regulator
applications because of its small size, robustness, and
very low ESR. However, care must be taken when these
capacitors are used at the input. A ceramic input
capacitor combined with trace or cable inductance
forms a high quality (under damped) tank circuit. If the
RT6253A/B circuit is plugged into a live supply, the input
voltage can ring to twice its nominal value, possibly
exceeding the device’s rating. This situation is easily
avoided by placing the low ESR ceramic input capacitor
in parallel with a bulk capacitor with higher ESR to damp
the voltage ringing.
The input capacitor should be placed as close as
possible to the VIN pins, with a low inductance
connection to the GND of the IC. In addition to a larger
bulk capacitor, a small ceramic capacitors of 0.1F
should be placed close to the VIN and GND pin. This
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DS6253A/B-03
October
2021
capacitors are 2 x 22F (Murata ceramic capacitor :
GRM219R60J226ME47), VRIPPLE can be obtained as
below.
The ripple caused by ESR (2m) can be calculated as :
VRIPPLE(ESR) = 0.85A 2m = 1.7mV
Considering the capacitance derating, the effective
capacitance is approximately 18F as the output
voltage is 1.2V, and another parameter is :
0.85A
= 5.1mV
8 2 18μF 580kHz
VRIPPLE = 1.7mV + 5.1mV = 6.8mV
VRIPPLE( C) =
Output Transient Undershoot and Overshoot
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21
RT6253A/B
In addition to voltage ripple at the switching frequency,
the output capacitor and its ESR also affect the voltage
sag (undershoot) and soar (overshoot) when the load
steps up and down abruptly. The ACOT® transient
response is very quick and output transients are usually
small. The following section shows how to calculate the
worst-case voltage swings in response to very fast load
steps.
and the output voltage :
Both undershoot voltage and overshoot voltage consist
of two factors : the voltage steps caused by the output
capacitor's ESR, and the voltage sag and soar due to
the finite output capacitance and the inductor current
slew rate. Use the following formulas to check if the ESR
is low enough (typically not a problem with ceramic
capacitors) and the output capacitance is large enough
to prevent excessive sag and soar on very fast load step
edges, with the chosen inductor value.
Output Voltage Setting
VSOAR =
L (IOUT )2
2 COUT VOUT
Because some modern digital loads can exhibit nearly
instantaneous load changes, the amplitude of the ESR
should be taken into consideration while calculating the
VSAG & VSOAR.
Set the desired output voltage using a resistive divider
from the output to ground with the midpoint connected
to FB, as shown in Figure 2. The output voltage is set
according to the following equation :
VOUT = VREF x (1 + RFB1 / RFB2)
VOUT
RFB1
The amplitude of the ESR step up or down is a function
of the load step and the ESR of the output capacitor :
FB
RT6253A/B
VESR _STEP = IOUT x RESR
The amplitude of the capacitive sag is a function of the
load step, the output capacitor value, the inductor value,
the input-to-output voltage differential, and the
maximum duty cycle. The maximum duty cycle during a
fast transient is a function of the on-time and the
minimum off-time since the ACOT® control scheme will
ramp the current using on-times spaced apart with
minimum off-times, which is as fast as allowed.
Calculate the approximate on-time (neglecting
parasites) and maximum duty cycle for a given input
and output voltage as :
tON =
VOUT
tON
and DMAX =
VIN fSW
tON + tOFF_MIN
The real on-time will slightly extend due to the voltage
drop which is related to output current; however, this ontime compensation can be neglected. Besides, the
minimum on-time is 60ns, typ. If the calculated on-time
is smaller than minimum on-time, it and VOUT will both
be clamped. Calculate the output voltage sag as :
VSAG =
L (IOUT )2
2 COUT ( VIN(MIN) DMAX − VOUT )
The amplitude of the capacitive soar is a function of the
load step, the output capacitor value, the inductor value
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22
RFB2
GND
Figure 2. Output Voltage Setting
Place the FB resistors within 5mm of the FB pin. Choose
RFB2 between 10k and 100k to minimize power
consumption without excessive noise pick-up and
calculate RFB1 as follows :
RFB1 =
RFB2 (VOUT − VREF )
VREF
For output voltage accuracy, use divider resistors with
1% or better tolerance.
Feed-Forward Capacitor Selection (CFF)
The RT6253A/B is optimized for low duty-cycle
applications and the control loop is stable with low ESR
ceramic output capacitors. In higher duty-cycle
applications (higher output voltages or lower input
voltages), the internal ripple signal will increase in
amplitude. Before the ACOT® control loop can react to
an output voltage fluctuation, the voltage change on the
feedback signal must exceed the internal ripple
amplitude. Because of the large internal ripple in this
condition, the response may become slower and underdamped. This situation will result in ringing waveform at
output terminal. In case of high output voltage
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DS6253A/B-03
October
2021
RT6253A/B
application, the phenomenon described above is more
visible because of large attenuation in feedback network.
As shown in Figure 3, adding a feedforward capacitor
(CFF) across the upper feedback resistor is
recommended. This increases the damping of the
control system.
calculated CFF, please decrease
feedforward capacitor CFF.
the
value
of
L
VOUT
SW
RT6253A/B
RFB1
CFF
COUT
FB
GND
RFB2
Figure 3. Feedback Loop with Feedforward Capacitor
Loop stability can be checked by viewing the load
transient response. A load step with a speed that
exceeds the converter bandwidth must be applied. For
ACOT® , loop bandwidth can be in the order of 100 to
200kHz, so a load step with 500ns maximum rising time
(dI/dt ≈ 2A/s) ensures the excitation frequency is
sufficient. It is important that the converter operates in
PWM mode, outside the light load efficiency range, and
below any current limit threshold. A load transient from
30% to 60% of maximum load is reasonable which is
shown in Figure 4.
Figure 5. Load Transient Response With and Without
Feedforward Capacitor
Enable Operation
The RT6253A/B is enabled when the VIN pin voltage
rises above VUVLO while the EN pin voltage exceeds
VEN_H. The RT6253A/B is disabled when the VIN pin
voltage falls below VUVLO − VUVLO or when the EN pin
voltage is below VEN_L. An internal pull-down resistor
REN_DN, which is connected form EN to GND, ensures
that the chip still stays in shutdown even if EN pin is
floated.
For automatic start-up, the EN pin, with high-voltage
rating, can be connected to the input supply VIN directly
as shown in Figure 6.
fCO
60% Load
30% Load
Figure 4. Example of Measuring the Converter BW by
Fast Load Transient
CFF can be calculated basing on below equation :
CFF =
1
2 BW
1 1 + 1
RFB1 RFB1 RFB2
The built-in hysteresis band makes the EN pin useful for
simple delay and timing circuits. The EN pin can be
externally connected to VIN by adding a resistor REN
and a capacitor CEN, as shown in Figure 7, to have an
additional delay. The time delay can be calculated by
the equation below with the EN's internal threshold, at
which switching operation begins.
CEN =
t
Rth ln
Figure 5. shows the transient performance with and
without feedfoward capacitor.
Note that, after defining the CFF please also check the
load regulation, because feedforward capacitor might
inject an offset voltage into VOUT to cause VOUT
inaccuracy. If the output voltage is over spec caused by
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DS6253A/B-03
October
2021
Vth
Vth − VEN_H
, where
Rth = REN // REN_DN
Vth = VIN
REN_DN
REN_DN + REN
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23
RT6253A/B
An external MOSFET can be used for logic control
which is shown in Figure 8. In this case, REN is
connected between VIN and the EN pin. The MOSFET
RT6253A/B
VIN
REN1
Q1 will be under logic control to pull down the EN pin.
If the device is desired to be shut down by EN pin before
VIN falls below the UVLO threshold, a resistive divider
(REN1 and REN2) can be used to externally set the input
under-voltage lockout threshold as shown in Figure 9.
For a given REN1, REN2 can be found by the equation
below for the desired VIN stop voltage.
VIN_STOP
REN2 //REN_DN
< VEN_L
REN1 + REN2 //REN_DN
After REN1 and REN2 are defined, the input voltage
VIN_START is obtained from
VEN_H
REN1 + REN2 //REN_DN
= VIN_START
REN2 //REN_DN
EN
REN2
RENL_DN
Figure 9. Resistor Divider for Lockout Threshold
Setting
If VIN shuts down faster than VOUT and VOUT is larger
than 3.7V, buck converter becomes boost converter
and generates negative current. To prevent these
condition, EN should be shut down before VIN falls
below VOUT. Therefore, the resistor divider for lockout
threshold is recommended.
Bootstrap Driver Supply
VIN
RT6253A/B
EN
REN_DN
Figure 6. Automatic Start-Up Setting
VIN
The bootstrap capacitor (CBOOT) between the BOOT
pin and the SW pin is used to create a voltage rail above
the applied input voltage, VIN. Specifically, the
bootstrap capacitor is charged through an internal diode
to a voltage equal to approximately PVCC each time the
low-side switch is turned on. The charge on this
capacitor is then used to supply the required current
during the remainder of the switching cycle. For most
applications, a 0.1F, 0603 ceramic capacitor with X5R
is recommended and the capacitor should have a 6.3 V
or higher voltage rating.
RT6253A/B
REN
External Bootstrap Diode (Optional)
EN
CEN
REN_DN
Figure 7. External Timing Control
VIN
RT6253A/B
REN
EN
Enable
Q1
A bootstrap capacitor of 0.1F low-ESR ceramic capacitor
is connected between the BOOT and SW pins to supply
the high-side gate driver. It is recommended to add an
external bootstrap diode between an external 5V voltage
supply and the BOOT pin as shown in Figure 10 to
improve efficiency when the input voltage is below 5.5V.
The bootstrap diode can be a low-cost one, such as
1N4148 or BAT54. The external 5V can be a fixed 5V
voltage supply from the system, or a 5V output voltage
generated by the RT6253A/B. Note that the BOOT voltage
VBOOT must be lower than 5.5V.
REN_DN
Figure 8. Digital Enable Control Circuit
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24
is a registered trademark of Richtek Technology Corporation.
DS6253A/B-03
October
2021
RT6253A/B
5V
5V
DBOOT
DBOOT
RBOOT
BOOT
BOOT
CBOOT
0.1μF
RT6253A/B
SW
SW
Figure 10. External Bootstrap Diode
Figure12. External Bootstrap Diode and Resistor at the
BOOT Pin
External Bootstrap Resistor (Optional)
The gate driver of an internal power MOSFET, utilized
as a high-side switch, is optimized for turning on the
switch. The gate driver is not only fast enough for
reducing switching power loss, but also slow enough for
minimizing EMI. The EMI issue is worse when the
switch is turned on rapidly due to the induced high di/dt
noises. When the high-side switch is turned off, the
discharging time on SW node is relatively slow because
there’s the presence of dead time, both high-side and
low-side MOSFETs are turned off in this interval. In
some cases, it is desirable to reduce EMI further, even
at the expense of some additional power dissipation.
The turn-on rate of the high-side switch can be slowed
by placing a small bootstrap resistor RBOOT between
the BOOT pin and the external bootstrap capacitor as
shown in Figure 11. The recommended range for the
RBOOT is several ohms to 47 ohms, and it could be 0402
or 0603 in size.
This will slow down the rates of the high-side switch turn
on and the rise of VSW . In order to improve EMI
performance and enhancement of the internal MOSFET
switch, the recommended application circuit is shown in
Figure 12, which includes an external bootstrap diode
for charging the bootstrap capacitor and a bootstrap
resistor RBOOT placed between the BOOT pin and the
capacitor/diode connection.
SW
Figure 11. External Bootstrap Resistor at the BOOT
Pin
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
October
2021
In many applications, the RT6253A/B does not generate
much heat due to its high efficiency and low thermal
resistance of its TSOT-23-6 (FC) package. However, in
applications which the RT6253A/B runs at a high
ambient temperature and high input voltage, the
generated heat may exceed the maximum junction
temperature of the part.
The RT6253A/B includes an over-temperature protection
(OTP) circuitry to prevent overheating due to excessive
power dissipation. If the junction temperature reaches
approximately 155°C, the RT6253A/B stops switching
the power MOSFETs until the temperature is cooled
down by 35°C.
Note that the over temperature protection is intended to
protect the device during momentary overload
conditions. The protection is activated outside of the
absolute maximum range of operation as a secondary
fail-safe and therefore should not be relied upon
operationally. Continuous operation above the specified
absolute maximum operating junction temperature may
impair device reliability or permanently damage the
device.
The maximum power dissipation can be calculated by
the following formula :
(
CBOOT
0.1μF
RT6253A/B
Thermal Considerations
)
PD(MAX) = TJ(MAX) − TA / θJA(EFFECTIVE)
RBOOT
BOOT
DS6253A/B-03
CBOOT
0.1μF
RT6253A/B
where TJ(MAX) is the maximum allowed junction
temperature of the die. For recommended operating
condition specifications, the maximum junction
temperature is 125°C. TA is the ambient operating
temperature, and θJA(EFFECTIVE) is the system-level
junction to ambient thermal resistance. It can be
estimated from thermal modeling or measurements in
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25
RT6253A/B
the system.
As a result, the new power dissipation is 1.182W due to
The thermal resistance of the device strongly depends
on the surrounding PCB layout and can be improved by
providing a heat sink of surrounding copper ground. The
addition of backside copper with thermal vias, stiffeners,
and other enhancements can also help reduce thermal
resistance.
the variation of RDS(ON). Therefore, the estimated new
As an example, considering the RT6253A is used in
application where VIN = 12V, IOUT = 3A, f SW = 580kHz,
VOUT = 5V. The efficiency at 5V, 3A is 90.63% by using
WE-74404054047 (4.7H, 30m DCR) as the inductor
and measured at room temperature. The core loss can
be obtained from its website, and it's 131mW. In this
case, the power dissipation of the RT6253A is
temperature of 125°C, care should be taken to reduce
(
)
1− η
2
POUT − IO
DCR + PCORE = 1.15W
η
Considering the system-level θJA(EFFECTIVE) is 67°C/W
(other heat sources are also considered), the junction
temperature of the regulator operating in a 25°C ambient
temperature is approximately :
TJ = 1.08W 67C/W + 25C = 102C
Figure 13 shows the RT6253A/B RDS(ON) versus
different junction temperatures. If the application
requires a higher ambient temperature, the device
power dissipation and the junction temperature of the
device need to be recalculated based on a higher
RDS(ON) since it increases with temperature.
TJ' = 1.182W 67C/W + 40C = 119.2C
If the application requires a higher ambient temperature
and may exceed the recommended maximum junction
the temperature rise of the part by using a heat sink or
air flow.
Resistance vs. Temperature
250
200
Resistance (mΩ)
PD, RT =
junction temperature is
150
RDS(ON)_H
100
50
RDS(ON)_L
0
-50
-25
0
25
50
75
100
125
Temperature (°C)
Figure 13. RT6253A/B RDS(ON) vs. Temperature
Using 40°C ambient temperature as an example. Due
to the variation of junction temperature is dominated by
the ambient temperature, the T'J at 40°C ambient
temperature can be pre-estimated as
TJ' = 102C + ( 40C − 25C) = 117C
According to Figure 13, the increasing RDS(ON) can be
found as
RDS( ON) _H = 130.4m (at 117C) − 125m (102C ) = 5.4m
RDS( ON) _L = 65.5m (at 117C) − 63.2m (102C ) = 2.3m
The external
power dissipation caused by the
increasing RDS(ON) at higher temperature can be
calculated as
PD,RDS( ON) = ( 3A )
2
5
5
2
5.4m + ( 3A ) 1 −
12
12
2.3m = 0.032W
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is a registered trademark of Richtek Technology Corporation.
DS6253A/B-03
October
2021
RT6253A/B
Layout Considerations
Follow the PCB layout guidelines below for optimal
performance of the device.
Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable
and jitter-free operation. The high current path
comprising of input capacitor, high-side FET,
inductor, and the output capacitor should be as short
as possible. This practice is essential for high
efficiency.
Place the input MLCC capacitors as close to the VIN
SW node is with high frequency voltage swing and
should be kept at small area. Keep analog
components away from the SW node to prevent stray
capacitive noise pickup.
Connect feedback network behind the output
capacitors. Place the feedback components next to
the FB pin.
For better thermal performance, design a wide and
thick plane for GND pin or add a lot of vias to GND
plane.
An example of PCB layout guide is shown in Figure 14.
and GND pins as possible. The major MLCC
capacitors should be placed on the same layer as the
RT6253A/B.
Add extra vias for thermal dissipation
Keep the SW node at small
area and keep analog
components away from the
SW node to prevent stray
capacitive noise pickup.
GND
GND
CBOOT RBOOT
SW
SW
1
REN
VIN
The VIN trace should
have enough width,
and use several vias to
shunt the high input
current.
L
CIN3
CIN2
Place the input MLCC capacitors as
close to the VIN and GND pins as
possible.
COUT1
Connect feedback network
behind the output
Place the feedback
components next to the FB pin.
RFB1
RFB2
CIN1
VOUT
COUT2
CFF
VOUT
GND
GND
Figure 14. Layout Guide
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October
2021
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27
RT6253A/B
Outline Dimension
Dimensions In Millimeters
Symbol
Dimensions In Inches
Min.
Max.
Min.
Max.
A
0.700
1.000
0.028
0.039
A1
0.000
0.100
0.000
0.004
B
1.397
1.803
0.055
0.071
b
0.300
0.559
0.012
0.022
C
2.591
3.000
0.102
0.118
D
2.692
3.099
0.106
0.122
e
0.950
0.037
H
0.080
0.254
0.003
0.010
L
0.300
0.610
0.012
0.024
TSOT-23-6 (FC) Surface Mount Package
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is a registered trademark of Richtek Technology Corporation.
DS6253A/B-03
October
2021
RT6253A/B
Symbol
Dimensions In Millimeters
Dimensions In Inches
Min
Max
Min
Max
A
0.500
0.600
0.020
0.024
A1
0.000
0.050
0.000
0.002
A3
0.080
0.180
0.003
0.007
b
0.150
0.300
0.006
0.012
D
1.500
1.700
0.059
0.067
E
1.500
1.700
0.059
0.067
E1
1.100
1.300
0.043
0.051
e
0.500
0.020
L
0.100
0.300
0.004
0.012
L1
0.200
0.400
0.008
0.016
SOT-563 (FC) Surface Mount Package
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October
2021
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29
RT6253A/B
Footprint Information
Package
TSOT-26/TSOT-26(FC)/SOT-26/SOT26(COL)
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30
Footprint Dimension (mm)
Number of
Pin
P1
A
B
C
D
M
6
0.95
3.60
1.60
1.00
0.70
2.60
Tolerance
±0.10
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DS6253A/B-03
October
2021
RT6253A/B
Package
Footprint Dimension (mm)
Number of
SOT-563(FC)
Tolerance
Pin
P1
A
B
C
D
M
6
0.50
2.42
1.02
0.70
0.30
1.30
±0.10
Richtek Technology Corporation
14F, No. 8, Tai Yuen 1st Street, Chupei City
Hsinchu, Taiwan, R.O.C.
Tel: (8863)5526789
Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should
obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot assume
responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be accurate and reliable.
However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DS6253A/B-03
October
2021
is a registered trademark of Richtek Technology Corporation.
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31