RT6276BHGQUF

® RT6276A/B 23V, 6A, 500kHz ACOT® Synchronous Step-Down Converter General Description Features The RT6276A/B is a simple, easy-to-use, 6A synchronous step-down DC-DC converter with an input supply voltage range of 4.5V to 23V. The device build-in an accurate 0.6V reference voltage and integrates low RDS(ON) power  MOSFETs to achieve high efficiency in a UQFN-12HL 3x3 package. The RT6276A/B adopts Advanced Constant OnTime (ACOT®) control architecture to provide an ultrafast transient response with few external components and to operate in nearly constant switching frequency over the line, load, and output voltage range. The RT6276A operates in automatic PSM that maintains high efficiency during light load operation. The RT6276B operates in Forced PWM that helps meet tight voltage regulation accuracy requirements. The RT6276A/B senses both FETs current for a robust over-current protection. It features cycle-bycycle current limit protection and prevent the device from the catastrophic damage in output short circuit, over current or inductor saturation. A built-in soft-start function prevents inrush current during start-up. The device also includes input under-voltage lockout, output under-voltage protection, over-voltage protection, and over-temperature protection (thermal shutdown) to provide safe and smooth operation in all operating conditions. The RT6276A/B is offered in a UQFN-12HL 3x3(FC) package.             Ω/15mΩ Ω Low RDS(ON) 6A Converter With Built-In 30mΩ Power FETs Input Supply Voltage Range : 4.5V to 23V Output Voltage Range : 0.6V to 6V Advanced Constant On-Time (ACOT®) Control  Ultrafast Transient Response  No Needs For External Compensations  Optimized for Low-ESR Ceramic Output Capacitors 0.6V ± 1% High-Accuracy Feedback Reference Voltage Optional for Operation Modes :  Power Saving Mode (PSM) (RT6276A)  Forced PWM Mode (RT6276B) Fixed Switching Frequency : 500kHz Built-In Internally Fixed Soft-Start (Typ. 0.4ms) Input Under-Voltage Lockout (UVLO) Output Under-Voltage Protection (UVP) with Hiccup Mode Output Over-Voltage Protection (OVP) with AutoRecovery Mode Ultrasonic Mode (USM) for RT6276A Available In UQFN-12HL 3x3 (FC) (U-Type) Package Ordering Information RT6276A/B Applications       Package Type QUF : UQFN-12HL 3x3 (FC) (U-Type) Industrial and Commercial Low Power Systems Computer Peripherals LCD Monitors and TVs Green Electronics/Appliances Point of Load Regulation for High-Performance DSPs, FPGAs, and ASICs Networking Communication Lead Plating System G : Green (Halogen Free and Pb Free) UVP Option H : Hiccup PWM Operation Mode A : Automatic PSM B: Forced PWM Note : Richtek products are :  RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020.  Copyright © 2020 Richtek Technology Corporation. All rights reserved. DS6276A/B-02 June 2020 Suitable for use in SnPb or Pb-free soldering processes. is a registered trademark of Richtek Technology Corporation. www.richtek.com 1 RT6276A/B Marking Information Pin Configuration RT6276AHGQUF BOOT AGND PGOOD 12 11 10 VCC FB BYP ILMT (TOP VIEW) 9 8 1 LX 2 NC 3 ME= : Product Code ME=YM DNN YMDNN : Date Code 7 5 EN VIN 4 PGND 6 RT6276BHGQUF MD= : Product Code MD=YM DNN UQFN-12HL 3x3 YMDNN : Date Code Functional Pin Description Pin No. Pin Name Pin Function 1 BOOT Boot-strap pin. Supply high side gate driver. Decouple this pin to LX pin with 0.1F ceramic cap. 2 LX Inductor pin. Connect this pin to the switching node of inductor. 3 NC No internal connection. 4 PGND Power ground. 5 VIN Input pin. Decouple this pin to GND pin with at least 10F ceramic cap. 6 EN Enable control. Pull this pin high to turn on the buck. Do not leave this pin floating. EN pin will also be used to set USM mode for RT6276A, when EN pin voltage is between 2.3V and 23V, it will enter USM mode, if EN pin voltage is between 0.8V and 1.7V, then it is normal mode. 7 PGOOD Power good Indicator. Open drain output when the output voltage is within 90% to 120% of regulation point. 8 AGND Analog ground. Connect AGND to a clean inner GND point with separate trace. 9 VCC 5V linear regulator output for internal control circuit. A capacitor (typical 1F) should be connected to PGND. Don’t connect to external Load. 10 FB Output feedback pin. Connect this pin to the center point of the output resistor divider to program the output voltage. 11 BYP Bypass input for the internal LDO. BYP is externally connected to the output of switching regulator. When the BYP voltage rises above the bypass switch turn-on threshold, the LDO regulator shuts down and the VCC pin is connected to the BYP pin through an internal switch. But when BYP pin is not used, it should be connected to ground. 12 ILMT Current limit setting pin. The current limit is set to 4A, 6A or 8A when this pin is pull low, floating or pull high respectively. Copyright © 2020 Richtek Technology Corporation. All rights reserved. www.richtek.com 2 is a registered trademark of Richtek Technology Corporation. DS6276A/B-02 June 2020 RT6276A/B Functional Block Diagram VIN BOOT Input UVLO 4.2V LX ILMT EN - Internal SST Thermal Protection 0.6V FB PWM Control & Protect Logic + Current Sense PGND PGOOD AGND + EN 4.7V + - VIN 5V LDO VCC BYP Copyright © 2020 Richtek Technology Corporation. All rights reserved. DS6276A/B-02 June 2020 is a registered trademark of Richtek Technology Corporation. www.richtek.com 3 RT6276A/B Operation The RT6276A/B is a high-efficiency, synchronous stepdown DC-DC converter that can deliver up to 6A output current from a 4.5V to 23V input supply. The RT6276A/B adopts ACOT®control mode, which can reduce the output capacitance and provide ultrafast transient responses, and allow minimal component sizes without any additional external compensation network. Input Under-Voltage Lockout In addition to the EN pin, the RT6276A/B also provides enable control through the VIN pin. It features an undervoltage lockout (UVLO) function that monitors the internal linear regulator (VCC). If VEN rises above VENH first, switching will still be inhibited until the VIN voltage rises above VUVLO. It is to ensure that the internal regulator is ready so that operation with not-fully-enhanced internal MOSFET switches can be prevented. After the device is powered up, if the input voltage VIN goes below the UVLO falling threshold voltage (VUVLO − ΔVUVLO), this switching will be inhibited; if VIN rises above the UVLO rising threshold (VUVLO), the device will resume switching. Soft-Start The soft-start function is used to prevent large inrush currents while the converter is being powered up. The RT6276A/B provides an internal soft-start feature for inrush control. During the start-up sequence, the internal capacitor is charged by an internal current source ISS to generate a soft-start ramp voltage as a reference voltage to the PWM comparator. The device will initiate switching and the output voltage will smoothly ramp up to its targeted regulation voltage only after this ramp voltage is greater than the feedback voltage VFB to ensure the converters have a smooth start-up. The typical soft-start time is 0.4ms. Output Over-Voltage Protection (OVP) and UnderVoltage Protection (UVP) The RT6276A/B includes output under-voltage protection (UVP) against over-load or short-circuited condition by constantly monitoring the feedback voltage VFB. If VFB drops below the under-voltage protection trip threshold (typically 60% of the internal reference voltage), the UV comparator will go high to turn off both the internal highside and low-side MOSFET switches. Power Good The PGOOD pin is an open-drain output and is connected to an external pull-up resistor. It is controlled by a comparator, which the feedback signal VFB is fed to. If VFB is above 90% of the internal reference voltage, the PGOOD pin will be in high impedance and VPGOOD will be held high. Otherwise, the PGOOD output will be pulled low. Over-Temperature Protection (Thermal Shutdown) The RT6276A/B includes an over-temperature protection (OTP) circuitry to prevent overheating due to excessive power dissipation. The OTP will shut down switching operation when junction temperature exceeds a thermal shutdown threshold TSD. Once the junction temperature cools down by a thermal shutdown hysteresis (ΔTSD), the IC will resume normal operation with a complete soft-start. Over-Current Limit The RT6276A/B current limit is (4A, 6A, 8A) it is a cycleby-cycle “valley” type, measuring the inductor current through the synchronous rectifier during the off-time while the inductor current ramps down. If output voltage drops below the output under-voltage protection level, the RT6276A/B will stop switching to avoid excessive heat. Copyright © 2020 Richtek Technology Corporation. All rights reserved. www.richtek.com 4 is a registered trademark of Richtek Technology Corporation. DS6276A/B-02 June 2020 RT6276A/B Absolute Maximum Ratings (Note 1) Supply Voltage, VIN -------------------------------------------------------------------------------------------Enable Pin Voltage, EN -------------------------------------------------------------------------------------- FB Pin Voltage, FB ------------------------------------------------------------------------------------------- Switch Voltage, LX --------------------------------------------------------------------------------------------33kHz). Once the internal oscillator is triggered, the controller will Copyright © 2020 Richtek Technology Corporation. All rights reserved. www.richtek.com 12 When PGOOD is pulled high and BYP pin of the RT6276A/ B voltage is above 4.7V, an internal 3Ω P-MOSFET switch connects VCC to the BYP pin while the VCC linear regulator is simultaneously turned off. Because the VCC is power source for internal logic device and gate driver. If bypass function was enabled, it’ s recommended to put a RC filter to BYP pin to enhance power quality for IC internal power. But when BYP pin is not used, it should be connected to ground. is a registered trademark of Richtek Technology Corporation. DS6276A/B-02 June 2020 RT6276A/B Current Limit The RT6276A/B current limit is adjustable (4A, 6A, 8A) by ILMT pin and it is a cycle-by-cycle “valley” type, measuring the inductor current through the synchronous rectifier during the off-time while the inductor current ramps down. The current is determined by measuring the voltage between source and drain of the synchronous rectifier, adding temperature compensation for greater accuracy. If the current exceeds the current limit, the on-time oneshot is inhibited until the inductor current ramps down below the current limit. Thus, only when the inductor current is well below the current limit, another on-time is permitted. If the output current exceeds the available inductor current (controlled by the current limit mechanism), the output voltage will drop. If it drops below the output under-voltage protection level (see next section) the IC will stop switching to avoid excessive heat. Output Over-Voltage Protection and Under-Voltage Protection The RT6276A/B include output over-voltage protection (OVP). If the output voltage rises above the regulation level, the high-side switch naturally remains off and the synchronous rectifier will turn on until the inductor current reaches the zero or next on-time one-shot is triggered If the output voltage exceeds the OVP threshold for longer than 20μs (typical), the IC's OVP is triggered. The RT6276A/B also include output under-voltage protection (UVP). If the output voltage drops below the UVP trip threshold for longer than 20μs (typical) the IC's UVP is triggered. The RT6276A/B use auto-recovery mode in OVP and Hiccup Mode in UVP. When the UVP function is triggered remains for a period, the RT6276A/B will retry automatically. When the UVP condition is removed, the converter will resume operation. the UVP is disabled during soft-start period. If the OVP function is triggered, the IC will stops switching. When the OVP condition is removed, the converter will resume operation. Input Under-Voltage Lockout In addition to the enable function, the RT6276A/B provide an Under-Voltage Lockout (UVLO) function that monitors the input voltage. To prevent operation without fullyCopyright © 2020 Richtek Technology Corporation. All rights reserved. DS6276A/B-02 June 2020 enhanced internal MOSFET switches, this function inhibits switching when input voltage drops below the UVLO-falling threshold. The IC resumes switching when input voltage exceeds the UVLO-rising threshold. Enable and Disable The RT6276A/B's EN is used to control converter, the enable voltage (EN) has a logic-low level of 0.4V. When VEN is below this level the IC enters shutdown mode When VEN exceeds its logic-high level of 0.8V the converter is fully operational. Soft-Start The RT6276A/B provide an internal soft-start function to prevent large inrush current and output voltage overshoot when the converter starts up. The soft-start (SS) automatically begins once the chip is enabled. During softstart, it clamps the ramping of internal reference voltage which is compared with FB signal. The typical soft-start duration is 0.4ms. A unique PWM duty limit control that prevents output over-voltage during soft-start period is designed specifically for FB floating. Power Good Output (PGOOD) The power good output is an open drain output that requires a pull-up resistor. When the output voltage is 15% (typical) below its set voltage, PGOOD will be pulled low. It is held low until the output voltage returns to 90% of its set voltage once more. During soft-start, PGOOD is actively held low and only allowed to be pulled high after soft-start is over and the output reaches 90% of its set voltage. There is a 10μs delay built into PGOOD circuitry to prevent false transition. External Bootstrap Capacitor (CBOOT) Connect a 0.1μF low ESR ceramic capacitor between BOOT pin and LX pin. This bootstrap capacitor provides the gate driver supply voltage for the high side N-channel MOSFET switch. The internal power MOSFET switch gate driver is optimized to turn the switch on fast enough for low power loss and good efficiency, but also slow enough to reduce EMI. Switch turn-on is when most EMI occurs since VLX rises rapidly. In some cases, it is desirable to reduce EMI is a registered trademark of Richtek Technology Corporation. www.richtek.com 13 RT6276A/B further, by the expense of some additional power dissipation. The switch turn-on can be slowed by placing a small (10Ω ≤ RBOOT ≤ 30Ω) resistance between BOOT and the external bootstrap capacitor. This will slow the high-side switch turn-on and VLX's rise. In order to improve EMI performance and enhancement of the internal MOSFET switch. Output Voltage Setting Set the desired output voltage using a resistive divider from the output to ground with the midpoint connected to FB. The output voltage is set according to the following equation: VOUT,VALLEY =  1+ R1   0.6V  R2  VOUT R1 FB RT6276A/B R2 GND Figure 2. Output Voltage Setting Place the FB resistors within 5mm of the FB pin. Choose R2 between 10kΩ and 100kΩ to minimize power consumption without excessive noise pick-up and calculate R1 as follows : R1 = R2 x (VOUT  0.6V) 0.6V For output voltage accuracy, use divider resistors with 1% or better tolerance. Inductor Selection Selecting an inductor involves specifying its inductance and also its required peak current. The exact inductor value is generally flexible and is ultimately chosen to obtain the best mix of cost, physical size, and circuit efficiency. Lower inductor values benefit from reduced size and cost and they can improve the circuit’s transient response, but they increase the inductor ripple current and output voltage ripple and reduce the efficiency due to the resulting higher peak currents. Conversely, higher inductor values increase efficiency, but the inductor will either be physically larger or have higher resistance since more turns of wire are required and transient response will be slower since Copyright © 2020 Richtek Technology Corporation. All rights reserved. www.richtek.com 14 more time is required to change current (up or down) in the inductor. A good compromise between size, efficiency, and transient response is to use a ripple current (ΔIL) about 20-50% of the desired full output load current. Calculate the approximate inductor value by selecting the input and output voltages, the switching frequency (f SW), the maximum output current (IOUT(MAX)) and estimating a ΔIL as some percentage of that current. V  (VIN  VOUT ) L  OUT VIN  fSW  IL Once an inductor value is chosen, the ripple current (ΔIL) is calculated to determine the required peak inductor current. VOUT  (VIN  VOUT ) and VIN  fSW  L I IL(PEAK)  IOUT(MAX)  L 2 To guarantee the required output current, the inductor needs a saturation current rating and a thermal rating that exceeds IL(PEAK). These are minimum requirements. To maintain control of inductor current in overload and shortcircuit conditions, some applications may desire current ratings up to the current limit value. However, the IC's output under-voltage shutdown feature make this unnecessary for most applications. IL  For best efficiency, choose an inductor with a low DC resistance that meets the cost and size requirements. For low inductor core losses some type of ferrite core is usually best and a shielded core type, although possibly larger or more expensive, will probably give fewer EMI and other noise problems. Input Capacitor Selection High quality ceramic input decoupling capacitor, such as X5R or X7R, with values greater than 20μF are recommended for the input capacitor. The X5R and X7R ceramic capacitors are usually selected for power regulator capacitors because the dielectric material has less capacitance variation and more temperature stability. Voltage rating and current rating are the key parameters when selecting an input capacitor. Generally, selecting an input capacitor with voltage rating 1.5 times greater than the maximum input voltage is a conservatively safe design. The input capacitor is used to supply the input RMS is a registered trademark of Richtek Technology Corporation. DS6276A/B-02 June 2020 RT6276A/B current, which can be calculated using the following equation : IRMS  VOUT  V I 2   (1  OUT )  IOUT 2  L  VIN VIN 12   The next step is to select a proper capacitor for RMS current rating. One good design uses more than one capacitor with low Equivalent Series Resistance (ESR) in parallel to form a capacitor bank. The input capacitance value determines the input ripple voltage of the regulator. The input voltage ripple can be approximately calculated using the following equation : VIN  IOUT  VIN V  (1  OUT ) CIN  fSW  VOUT VIN The typical operating circuit is recommended to use two 10μF low ESR ceramic capacitors on the input. Output Capacitor Selection The IC is optimized for ceramic output capacitors and best performance will be obtained by using them. The total output capacitance value is usually determined by the desired output voltage ripple level and transient response requirements for sag (undershoot on positive load steps) and soar (overshoot on negative load steps). Output ripple at the switching frequency is caused by the inductor current ripple and its effect on the output capacitor's ESR and stored charge. These two ripple components are called ESR ripple and capacitive ripple. Since ceramic capacitors have extremely low ESR and relatively little capacitance, both components are similar in amplitude and both should be considered if ripple is critical. VRIPPLE  VRIPPLE(ESR)  VRIPPLE(C) VRIPPLE(ESR)  IL  RESR VRIPPLE(C)  IL 8  COUT  fSW In addition to voltage ripple at the switching frequency, the output capacitor and its ESR also affect the voltage sag (undershoot) and soar (overshoot) when the load steps up and down abruptly. The ACOT® transient response is very quick and output transients are usually small. However, the combination of small ceramic output capacitors (with little capacitance), low output voltages (with little stored charge in the output capacitors), and Copyright © 2020 Richtek Technology Corporation. All rights reserved. DS6276A/B-02 June 2020 low duty cycle applications (which require high inductance to get reasonable ripple currents with high input voltages) increases the size of voltage variations in response to very quick load changes. Typically, load changes occur slowly with respect to the IC's switching frequency. However, some modern digital loads can exhibit nearly instantaneous load changes and the following section shows how to calculate the worst-case voltage swings in response to very fast load steps. The amplitude of the ESR step up or down is a function of the load step and the ESR of the output capacitor : VESR_STEP  IOUT  RESR The amplitude of the capacitive sag is a function of the load step, the output capacitor value, the inductor value, the input-to-output voltage differential, and the maximum duty cycle. The maximum duty cycle during a fast transient is a function of the on-time and the minimum off-time since the ACOT® control scheme will ramp the current using on-times spaced apart with minimum off-times, which is as fast as allowed. Calculate the approximate on-time (neglecting parasitics) and maximum duty cycle for a given input and output voltage as : VOUT tON tON  and DMAX  VIN  fSW tON + tOFF_MIN The actual on-time will be slightly longer as the IC compensates for voltage drops in the circuit, but we can neglect both of these since the on-time increases compensations for the voltage losses. Calculate the output voltage SAG as : VSAG  L  (IOUT )2 2  COUT  ( VIN(MIN)  DMAX  VOUT ) The amplitude of the capacitive SOAR is a function of the load step, the output capacitor value, the inductor value and the output voltage : VSOAR  L  ( IOUT )2 2  COUT  VOUT Most applications never experience instantaneous full load steps and the IC's high switching frequency and fast transient response can easily control voltage regulation at all times. Therefore, sag and soar are seldom an issue except in very low-voltage CPU core or DDR memory supply applications, particularly for devices with high clock frequencies and quick changes into and out of sleep is a registered trademark of Richtek Technology Corporation. www.richtek.com 15 RT6276A/B In any application with large quick transients, it should calculate soar and sag to make sure that over-voltage protection and under-voltage protection will not be triggered. Thermal Considerations The junction temperature should never exceed the absolute maximum junction temperature TJ(MAX), listed under Absolute Maximum Ratings, to avoid permanent damage to the device. The maximum allowable power dissipation depends on the thermal resistance of the IC package, the PCB layout, the rate of surrounding airflow, and the difference between the junction and ambient temperatures. The maximum power dissipation can be calculated using the following formula : 4.0 Maximum Power Dissipation (W)1 modes. In such applications, simply increasing the amount of ceramic output capacitor (sag and soar are directly proportional to capacitance) or adding extra bulk capacitance can easily eliminate any excessive voltage transients. Four-Layer PCB 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 0 25 50 75 100 125 Ambient Temperature (°C) Figure 3. Derating Curve of Maximum Power Dissipation PD(MAX) = (TJ(MAX) − TA) / θJA where TJ(MAX) is the maximum junction temperature, TA is the ambient temperature, and θJA is the junction-to-ambient thermal resistance. For continuous operation, the maximum operating junction temperature indicated under Recommended Operating Conditions is 125°C. The junction-to-ambient thermal resistance, θJA, is highly package dependent. For a UQFN12HL 3x3 (FC) package, the thermal resistance, θJA, is 38.4°C/W on a standard JEDEC 51-7 high effective-thermalconductivity four-layer test board. The maximum power dissipation at TA = 25°C can be calculated as below : PD(MAX) = (125°C − 25°C) / (38.4°C/W) = 2.6W for a UQFN-12HL 3x3 (FC) package. The maximum power dissipation depends on the operating ambient temperature for the fixed TJ(MAX) and the thermal resistance, θJA. The derating curves in Figure 3 allows the designer to see the effect of rising ambient temperature on the maximum power dissipation. Copyright © 2020 Richtek Technology Corporation. All rights reserved. www.richtek.com 16 is a registered trademark of Richtek Technology Corporation. DS6276A/B-02 June 2020 RT6276A/B Layout Considerations AGND VOUT   Put the input capacitor as close as possible to the device pins (VIN and GND). The LX node encounters high frequency voltage swings so it should be kept in a small area. Keep sensitive components away from the LX node to prevent stray as possible.  The GND pin should be connected to a strong ground plane for heat sinking and noise protection.  Avoid using vias in the power path connections that have switched currents (from CIN to GND and CIN to VIN) and the switching node (LX). L COUT LX NC FB BYP 12 11 10 AGND PGOOD Make traces of the high current paths as short and wide as possible. LX should be connected to inductor by wide and short trace. Keep sensitive components away from this trace. BOOT VCC  R2 The optional compensation Compensation components must be connected as close CFF (opt.) GND to the IC as possible. CBYS CVCC GND VVCC ILMT Layout is very important in high frequency switching converter design. The PCB can radiate excessive noise and contribute to converter instability with improper layout. Certain points must be considered before starting a layout using the IC. R1 9 8 1 7 The input capacitor must be placed as close to the IC as possible 5 EN VIN 4 PGND 6 VIN 2 CIN 3 The output capacitor must be placed near the IC GND Figure 4. PCB Layout Guide An example of PCB layout guide is shown in Figure 4 for reference. Copyright © 2020 Richtek Technology Corporation. All rights reserved. DS6276A/B-02 June 2020 is a registered trademark of Richtek Technology Corporation. www.richtek.com 17 RT6276A/B Outline Dimension Symbol Dimensions In Millimeters Dimensions In Inches Min Max Min Max A 0.500 0.600 0.020 0.024 A1 0.000 0.050 0.000 0.002 A3 0.100 0.200 0.004 0.008 D 2.900 3.100 0.114 0.122 E 2.900 3.100 0.114 0.122 b 0.100 0.200 0.004 0.008 b1 0.180 0.280 0.007 0.011 L 0.800 1.000 0.031 0.039 L1 1.730 1.930 0.068 0.076 L2 0.250 0.450 0.010 0.018 e 0.720 0.028 e1 0.900 0.035 e2 0.450 0.018 K 0.500 0.020 K1 0.950 0.037 U-Type 12HL QFN 3x3 (FC) Package Copyright © 2020 Richtek Technology Corporation. All rights reserved. www.richtek.com 18 is a registered trademark of Richtek Technology Corporation. DS6276A/B-02 June 2020 RT6276A/B Footprint Information Package Number of Pin UQFN3*312H(FC) 12 Footprint Dimension (mm) P P1 P2 Ax C*8 C1*2 C2*2 D*12 K K1 K2 K3 0.450 0.720 0.900 3.800 0.800 1.350 2.280 0.230 0.619 0.169 1.100 1.900 Tolerance ±0.050 Richtek Technology Corporation 14F, No. 8, Tai Yuen 1st Street, Chupei City Hsinchu, Taiwan, R.O.C. Tel: (8863)5526789 Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries. DS6276A/B-02 June 2020 www.richtek.com 19