®
RTQ2965-QA
60VIN, 5A, Asynchronous Step-Down Converter with Low
Quiescent Current
General Description
Features
The RTQ2965 is a 5A, high-efficiency, peak current mode
control asynchronous step-down converter which is
optimized for automotive applications. The device operates
with input voltages from 4V to 60V and is protected from
load dump transients up to 65V, eases input surge
protection design. The device can program the output
voltage between 0.8V to VIN. The low quiescent current
design with the integrated low R DS(ON) of high-side
MOSFET achieves high efficiency over the wide load range.
The peak current mode control with simple external
compensation allows the use of small inductors and
results in fast transient response and good loop stability.
The wide switching frequency of 100kHz to 2500kHz allows
for efficiency and size optimization when selecting the
output filter components. The ultra-low minimum on-time
enables constant-frequency operation even at very high
step-down ratios. For switching noise sensitive
applications, it can be externally synchronized from
300kHz to 2200kHz. The build-in spread-spectrum
frequency modulation further helping systems designers
with better EMC management.
The RTQ2965 provides complete protection functions such
as input under-voltage lockout, output under-voltage
protection, output over-voltage protection, over-current
protection, and thermal shutdown. Cycle-by-cycle current
limit provides protection against shorted outputs, and softstart eliminates input current surge during start-up. The
RTQ2965 is available in WDFN-10L 4x4 and SOP-8
(Exposed pad) packages.
AEC-Q100 Grade 1 Qualified
FMEA Compliant Pinout
Wide Input Voltage Range
4.5V to 60V
4V to 60V (Soft-start is finished)
Wide Output Voltage Range : 0.8V to VIN
0.8V ±1% Reference Accuracy
Peak Current Mode Control
Integrated 70mΩ
Ω High-Side MOSFET
Low Quiescent Current : 100μ
μA
Low Shutdown Current : 2.25μ
μA
Adjustable Switching : 100kHz to 2.5MHz
Synchronizable Switching : 300kHz to 2.2MHz
Power Saving Mode (PSM) at Light Load
Built-In Spread-Spectrum Frequency Modulation
Low Dropout at Light Loads with Integrated Boot
Recharge FET
Externally Adjustable Soft-Start by Part Number
Option
Power Good Indication by Part Number Option
Enable Control
Adjustable UVLO Voltage and Hysteresis
Adjacent Pin-Short Protection
Built-In UVLO, UVP, OVP, OCP, OTP
Applications
Automotive, Communications and Industrial 12V, 24V
and 48V Power Systems
Industrial Automation and Motor Control
Vehicle Accessories : GPS, Entertainment
Simplified Application Circuit
RTQ2965GQW
VIN
VIN
CBOOT
BOOT
CIN
SW
Enable Signal
PGOOD
D1
RFB1
RFB2
RT/SYNC
L1
SW
VOUT
D1
EN
Enable Signal
RRT
CBOOT
BOOT
CIN
COUT
FB
PGOOD
VIN
VOUT
EN
CCOMP1
RTQ2965GSP
VIN
L1
CCOMP1
RFB1
COUT
FB
RFB2
RCOMP
COMP
RCOMP
COMP
SS/TR
CCOMP2
CSS
GND PAD
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CCOMP2
RT/SYNC
RRT
GND PAD
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RTQ2965-QA
Ordering Information
RTQ2965
Pin Configuration
-QA
(TOP VIEW)
Grade
QA : AEC-Q100 Qualified and
Screened by High Temperature
Pin 1 Orientation***
(2) : Quadrant 2, Follow EIA-481-D
(WDFN-10L 4x4 only)
Package Type
SP : SOP-8 (Exposed Pad-Option 2)
QW : WDFN-10L 4x4 (W-Type)
Lead Plating System
G : Green (Halogen Free and Pb Free)
Note :
BOOT
VIN
2
EN
3
RT/SYNC
4
8
SW
7
GND
6
COMP
5
FB
PAD
9
SOP-8 (Exposed pad)
BOOT
1
10
PGOOD
VIN
2
9
SW
EN
3
8
GND
SS/TR
4
7
COMP
RT/SYNC
5
6
FB
PAD
11
***Empty means Pin1 orientation is Quadrant 1
WDFN-10L 4x4
Richtek products are :
RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020.
Suitable for use in SnPb or Pb-free soldering processes.
Marking Information
RTQ2965GSP-QA
RTQ2965GSP-QA : Product Number
RTQ2965
GSP-QAYMDNN
YMDNN : Date Code
RTQ2965GQW-QA
8U= : Product Code
8U=YM
DNN
YMDNN : Date Code
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RTQ2965-QA
Functional Pin Description
Pin No.
Pin Name
SOP-8
(Exposed Pad)
WDFN-10L 4x4
1
1
Pin Function
BOOT
Bootstrap capacitor connection node to supply the high-side
gate driver. Connect a 0.1F ceramic capacitor between this
pin and the SW pin.
2
2
VIN
Power input. The input voltage range is from 4V to 60V.
Connect a suitable input capacitor between this pin and GND,
usually four 2.2F or larger ceramic capacitors with two
typical capacitance 4.7F.
3
3
EN
Enable control pin with internal pull-up current source. Float
or provide a logic-high ( 1.2V) enables the converter; a
logic-low forces the device into shutdown mode.
SS/TR
Soft-start and tracking control input. Connect a capacitor from
SS to GND to set the soft-start period. ”Do Not” leave this pin
floating to avoid inrush current during power up. It also can
be used to track and sequence because the SS/TR pin
voltage can override the internal reference voltage.
--
4
4
5
RT/SYNC
Frequency setting and external synchronous signal input.
Connect a resistor from this pin to GND to set the switching
frequency. Tie to a clock source for synchronization to an
external frequency.
5
6
FB
Output voltage sense. Sense the output voltage at the FB pin
through a resistive divider. The feedback reference voltage is
0.8V typically.
6
7
COMP
Compensation node. Connect external compensation
elements to this pin to stabilize the control loop.
7
8
GND
Ground. Provide the ground return path for the control
circuitry.
8
9
SW
Switch node. SW is the switching node that supplies power
to the output. Connect the output LC filter from SW to the
output load.
--
10
PGOOD
Open-drain power-good indication output. Once being
started-up, PGOOD will be pulled low to GND if any internal
protection is triggered.
9 (Exposed Pad)
11 (Exposed Pad) PAD
Exposed pad. The exposed pad is internally unconnected
and must be soldered to a large PCB copper area for
maximum power dissipation.
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RTQ2965-QA
Functional Block Diagram
WDFN-10L 4x4 package
PGOOD
VIN
EN
Shutdown
+
Thermal
Shutdown
UV
IEN
-
Voltage
Reference
+
IEN_Hys
OV
Logic
Shutdown
Logic
+
-
Enable
Threshold
PGOOD
Enable
Comparator
SS/TR
VCC
Regulator
Current
Sense
Pulse-Skip
Minimum Clamp
ISS
0.8V
Shutdown
-
+
FB
UVLO
+ EA
+
-
High-Side
MOSFET
Logic
+
SW
Shutdown
+
COMP
BOOT
BOOT
UVLO
PWM
Comparator
Slope
Compensation
OC
Clamp
Frequency
Foldback
GND
BOOT
recharge
MOSFET
Oscillator
RT/SYNC
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RTQ2965-QA
SOP-8 (Exposed pad) package
VIN
EN
Shutdown
+
Thermal
Shutdown
UV
IEN
UVLO
IEN_Hys
-
Voltage
Reference
+
OV
Logic
Shutdown
Logic
+
Enable
Threshold
-
Shutdown
-
Enable
Comparator
VCC
Regulator
Current
Sense
Pulse-Skip
Minimum Clamp
FB
0.8V
SS
+ EA
+
-
High-Side
MOSFET
Logic
+
SW
Shutdown
+
COMP
BOOT
BOOT
UVLO
PWM
Comparator
Slope
Compensation
OC
Clamp
Frequency
Foldback
GND
BOOT
recharge
MOSFET
Oscillator
RT/SYNC
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RTQ2965-QA
Operation
Control Loop
The RTQ2965 is a high efficiency asynchronous step-down
converter utilizes the peak current mode control. An
internal oscillator initiates the turn-on of the high-side
MOSFET. At the beginning of each clock cycle, the internal
high-side MOSFET turns on, allowing current to ramp-up
in the inductor. The inductor current is internally monitored
during each switching cycle. The output voltage is sensed
on the FB pin via the resistor divider, R1 and R2, and
compared with the internal reference voltage (VREF) to
generate a compensation signal (VCOMP) on the COMP
pin. A control signal derived from the inductor current is
compared to the voltage at the COMP pin, derived from
the feedback voltage. When the inductor current reaches
its threshold, the high-side MOSFET is turned off and
inductor current ramps down. While the high-side MOSFET
is off, the inductor current is supplied through the external
low-side diode, freewheel diode, connected between the
SW pin and GND. This cycle repeats at the next clock
cycle. In this way, duty-cycle and output voltage are
controlled by regulating inductor current.
Light Load Operation
The RTQ2965 operates in power saving mode (PSM) at
light load to improve light load efficiency. IC starts to switch
when VFB is lower than PSM threshold ( VREF x 1.005,
typically) and stops switching when VFB is high enough.
During PSM, the peak inductor current (I L_PEAK) is
controlled by the minimum on-time of high-side MOSFET
to ensure the low output voltage ripple. During nonswitching period, most of the internal circuit is shut down,
and the supply current drops to quiescent current (100μA,
typically) to reduce the quiescent power consumption.
With lower output loading, the non-switching period is
longer, so the effective switching frequency becomes lower
to reduce the switching loss and switch driving loss.
Switching Frequency Selection and Synchronization
The RTQ2965 provides an RT/SYNC pin for switching
frequency selection. The switching frequency can be set
by using external resistor RRT/SYNC and the switching
frequency range is from 100kHz to 2.5MHz. The RTQ2965
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can also be synchronized with an external clock ranging
from 300kHz to 2.2MHz by RT/SYNC pin. The switching
frequency of synchronization should be equal to or higher
than the frequency set by the RT resistor. For example, if
the switching frequency of synchronization is 500kHz or
higher, the RRT/SYNC should be selected for 500kHz.
The RTQ2965 implements a frequency foldback function
to protect the device at over-load or short-circuited
condition, especially higher switching frequencies and input
voltages. The switching frequency is divided by 1, 2, 4,
and 8 as the FB pin voltage falls from 0.8 V to 0 V for
switching frequency control by RT resistor setting mode
and the synchronization mode both. The frequency foldback
function increases the switching cycle period and provides
more time for the inductor current to ramp down.
Maximum Duty Cycle Operation
The RTQ2965 is designed to operate in dropout at the
high duty cycle approaching 100%. If the operational duty
cycle is large and the required off-time becomes smaller
than minimum off-time, the RTQ2965 starts to enable skip
off-time function and keeps high-side MOSFET on
continuously.
The RTQ2965 implements skip off-time function to achieve
high duty approaching 100% and the maximum output
voltage is near the minimum input supply voltage of the
application for input voltage momentarily falls down to the
normal output voltage requirement. The input voltage at
which the devices enter dropout changes depending on
the input voltage, output voltage, switching frequency, load
current, and the efficiency of the design.
For normal operation, the minimum input voltage can be
calculated from below equation :
VOUT + IOUT_MAX RL + VD
VIN_MIN =
1 fSW tOFF_MIN
+ IOUT_MAX RDS(ON)_H VD
where VIN_MIN is the minimum normal operating input
voltage; RDS(ON)_H is the on-resistance of the high-side
MOSFET; VD is the forward conduction voltage of the
freewheel diode; RL is the DC resistance of inductor.
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BOOT UVLO
The BOOT UVLO circuit is implemented to ensure a
sufficient voltage of BOOT capacitor for turning on the highside MOSFET at any conditions. The BOOT UVLO usually
actives at extremely high conversion ratio or the higher
VOUT application operates at very light load. With such
conditions when the BOOT to SW voltage falls below
VBOOT_UVLO_L (2.7V, typically), the device turns on the
internal BOOT recharge FET (150ns, typically) to charge
the BOOT capacitor. The BOOT UVLO is sustained until
the BOOT to SW voltage is higher than VBOOT_UVLO_H
(2.8V, typically).
Enable Control
The RTQ2965 provides an EN pin, as an external chip
enable control, to enable or disable the device. If VEN is
held below the enable threshold voltage, switching is
inhibited even if the VIN voltage is above VIN under-voltage
lockout threshold (VUVLOH). If VEN is held below 0.4V, the
converter will enter into shutdown mode, that is, the
converter is disabled. During shutdown mode, the supply
current can be reduced to ISHDN (2.25μA, typically). If the
EN voltage rises above the enable threshold voltage while
the VIN voltage is higher than VUVLOH, the device will be
turned on, that is, switching being enabled and soft-start
sequence being initiated. The EN pin has an internal pullup current source IEN (1.2μA, typically) that enables
operation of the RTQ2965 when the EN pin floats. The EN
pin can be used to adjust the under-voltage lockout (UVLO)
threshold and hysteresis by using two external resistors.
The RTQ2965 implements additional hysteresis current
source IEN_Hys (3.4μA, typically) to adjust the UVLO. The
IEN_Hys is sourced out of the EN pin when VEN is larger
than enable threshold voltage. When the VEN falls below
enable threshold voltage, the IEN_Hys will be stopped
sourcing out of the EN pin.
by selecting the value of the external soft-start capacitor
CSS/TR connected from the SS/TR pin to ground or
controlled by external ramp voltage to SS/TR pin. During
the start-up sequence, the soft-start capacitor is charged
by an internal current source ISS (1.7μA, typically) to
generate a soft-start ramp voltage as a reference voltage
to the PWM comparator. The high-side MOSFET will start
switching if the voltage difference between SS/TR pin and
FB pin is equal to 42mV ( i.e. VSS/TR − VFB = 42mV,
typically) during power-up period. If the output is prebiased to a certain voltage during start-up for some reason,
the device will not start switching until the voltage difference
between SS/TR pin and FB pin is equal to 42mV
(typically). Only when this ramp voltage is higher than the
feedback voltage VFB, the switching will be resumed. The
FB voltage will track the SS/TR pin ramp voltage with a
SS/TR to FB offset voltage (42mV, typically) during softstart interval. The output voltage can then ramp up smoothly
to its targeted regulation voltage, and the converter can
have a monotonic smooth start-up. For soft-start control,
the SS pin should never be left unconnected. After the FB
pin voltage rises above 94% of VREF (typically), the
PGOOD pin will be in high impedance and the VPGOOD will
be held high. The typical start-up waveform shown in
Figure 1 indicates the sequence and timing between the
output voltage and related voltage.
VIN = 4.5V to 60V
VIN
EN
tSS
540µs
SS
42mV
VOUT
10% x
VOUT
90% x VOUT
94% x VOUT
PGOOD
Soft-Start and Tracking Control
The soft-start function is used to prevent large inrush
currents while the converter is being powered up. The
RTQ2965GSP provides internal soft-start and the
RTQ2965GQW provides external soft-start function for
inrush currents control. The RTQ2965GQW provides an
SS/TR pin so that the soft-start time can be programmed
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DSQ2965-QA-01 March 2021
Figure 1 Start-Up Sequence for RTQ2965GQW
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RTQ2965-QA
Power Good Indication
Spread-Spectrum Operation
The RTQ2965GQW features an open-drain power-good
output (PGOOD) to monitor the output voltage status. The
output delay of comparator prevents false flag operation
for short excursions in the output voltage, such as during
line and load transients. Pull-up PGOOD with a resistor
to an external voltage source and it is recommended to
use pull-up resistance between the values of 1 and 10kΩ
to reduce the switching noise coupling to PGOOD pin.
The PGOOD assertion requires input voltage above 2V.
The power-good function is controlled by a comparator
connected to the feedback signal VFB. If VFB rises above
the power-good high threshold (VTH_PGLH1) (94% of the
reference voltage, typically), the PGOOD pin will be in
high impedance and VPGOOD will be held high after a certain
delay elapsed. When VFB falls below power-good low
threshold (VTH_PGHL2) (92% of the reference voltage,
typically) or exceeds VTH_PGHL1 (109% of the reference
voltage, typically), the PGOOD pin will be pulled low. For
VFB higher than VTH_PGHL1, VPGOOD can be pulled high
again if VFB drops back by a power-good high threshold
(VTH_PGLH2) (106% of the reference voltage, typically).
Once being started-up, if any internal protection is
triggered, PGOOD will be pulled low to GND. The internal
open-drain pull-down device (45Ω, typically) will pull the
PGOOD pin low. The power good indication profile is shown
in Figure 2.
Due to the periodicity of the switching signals, the energy
concentrates in one particular frequency and also in its
VTH_PGHL1
VTH_PGLH2
VTH_PGLH1
VTH_PGHL2
VFB
VPGOOD
Figure 2. The Logic of PGOOD for RTQ2965GQW
odds harmonics. These levels or energy is radiated and
therefore this is where a potential EMI issue arises. The
RTQ2965 have built-in spread-spectrum function and it
can be enable to overcome EMI issue. The switching
frequency varies by +6% relative to the switching frequency
setting. By varying the frequency +6% only in the positive
direction, the RTQ2965 still guarantees that the 2.1MHz
switching frequency setting does not drop into the AM
band limit of 1.8MHz.
Input Under-Voltage Lockout
In addition to the EN pin, the RTQ2965 also provides enable
control through the VIN pin. If VEN rises above VTH_EN first,
the switching will be inhibited until the VIN voltage rises
above VUVLOH. It is to ensure that the internal regulator is
ready so that operation with not-fully-enhanced internal
high-side MOSFET can be prevented. After the device is
powered up, if the input voltage VIN goes below the UVLO
falling threshold voltage (VUVLOL), this switching will be
inhibited; if VIN rises above the UVLO rising threshold
(VUVLOH), the device will resume switching. Note that VIN
= 4V is only design for cold crank requirement, normal
input voltage should be larger than the VUVLOH.
High-Side MOSFET Peak Current Limit Protection
The RTQ2965 includes a cycle-by-cycle high-side
MOSFET peak current-limit protection against the
condition that the inductor current increasing abnormally,
even over the inductor saturation current rating. The
inductor current through the high-side MOSFET will be
measured after a certain amount of delay when the highside MOSFET being turned on. If an over-current condition
occurs, the converter will immediately turn off the highside MOSFET to prevent the inductor current exceeding
the high-side MOSFET peak current limit (ILIM).
Output Under-Voltage Protection
The RTQ2965 includes output under-voltage protection
(UVP) against over-load or short-circuited condition by
constantly monitoring the feedback voltage VFB. If VFB
drops below the under-voltage protection trip threshold
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RTQ2965-QA
(50% of the internal reference voltage, typically), the UV
comparator will go high to turn off the internal high-side
switch. If the output under-voltage condition continues for
a period of time, the RTQ2965 enters output under-voltage
protection with hiccup mode and discharges the CSS/TR
by an internal discharging current source ISS_DIS (0.5μA,
typically). During hiccup mode, the device remains
shutdown. After the SS pin voltage is discharged to less
than 54mV (typically), the RTQ2965 attempts to re-start
up again, and the internal charging current source ISS
(1.7μA, typically) gradually increases the voltage on CSS/
TR. The high-side MOSFET will start switching when
voltage difference between SS pin and FB pin is equal to
42mV ( i.e. VSS − VFB = 42mV, typically). If the output
under-voltage condition is not removed, the high-side
MOSFET stops switching when the voltage difference
between SS pin and FB pin is 1.2V ( i.e. VSS − VFB =
1.2V, typically) and then the ISS_DIS discharging current
source begins to discharge CSS/TR. Upon completion of
the soft-start sequence, if the output under-voltage
condition is removed, the converter will resume normal
operation; otherwise, such cycle for auto-recovery will be
repeated until the output under-voltage condition is cleared.
Hiccup mode allows the circuit to operate safely with low
input current and power dissipation, and then resume
normal operation as soon as the over-load or short-circuit
condition is removed. A short circuit protection and
recovery profile is shown in Figure 3.
Since the CSS/TR will be discharged to 54mV when the
RTQ2965 enters output under-voltage protection, the first
discharging time (tSS_DIS1) can be calculated as below :
V 0.054
tSS_DIS1 = CSS SS
ISS_DIS
The equation below assumes that the VFB will be 0 at
short-circuited condition and it can be used to calculate
the CSS/TR discharging time (tSS_DIS2) and charging time
(tSS_CH) during hiccup mode.
tSS_DIS2 = CSS
tSS_CH = CSS
1.146
ISS_DIS
VOUT
2V/DIV
Short Removed
Short Applied
VPGOOD
4V/DIV
VSS
4V/DIV
I SW
3A/DIV
Figure 3. Short-Circuit Protection and Recovery
Output Over-Voltage Protection
The RTQ2965 includes an output over-voltage protection
(OVP) circuit to limit output voltage. Since the VFB is lower
than the reference voltage (VREF) at over-load or shortcircuited condition, the COMP voltage will be high to
demand maximum output current. Once the over-load or
short-circuited condition is removed, the COMP voltage
resumes to the normal voltage to regulate the output
voltage. The output voltage leads to the possibility of an
output overshoot if the load transient is faster than the
COMP voltage transient response, especially for small
output capacitance. If the VFB goes above the 109% of
the reference voltage, the high-side MOSFET will be forced
off to limit the output voltage. When the VFB drops lower
than the 106% of the reference voltage, the high-side
MOSFET will be resumed.
Pin-Short Protection
The RTQ2965 provides pin-short protection for neighbor
pins. The internal protection fuse will be burned out to
prevent IC smoke, fire and spark when BOOT pin is
shorted to VIN pin. The EN pin with high-voltage rating
can avoid IC burn-out when EN pin is shorted to VIN pin.
The hiccup mode protection will be triggered to avoid IC
burn-out when SW pin is shorted to ground during internal
high-side MOSFET turns on.
1.146
ISS_CH
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RTQ2965-QA
Over-Temperature Protection
The RTQ2965 includes an over-temperature protection
(OTP) circuitry to prevent overheating due to excessive
power dissipation. The OTP will shut down switching
operation when junction temperature exceeds a thermal
shutdown threshold TSD. Once the junction temperature
cools down by a thermal shutdown hysteresis (ΔTSD), the
IC will resume normal operation with a complete soft-start.
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Absolute Maximum Ratings
Supply Voltage, VIN -----------------------------------------------------------------------------------------------------Enable Voltage, EN ------------------------------------------------------------------------------------------------------Switch Voltage, SW -----------------------------------------------------------------------------------------------------SW (t ≤ 100ns) ------------------------------------------------------------------------------------------------------------SW (t ≤ 5ns) ---------------------------------------------------------------------------------------------------------------Power Good Voltage, PGOOD ----------------------------------------------------------------------------------------BOOT to SW (BOOT−SW) ---------------------------------------------------------------------------------------------All Other Pins -------------------------------------------------------------------------------------------------------------Lead Temperature (Soldering, 10 sec.) ------------------------------------------------------------------------------Junction Temperature ----------------------------------------------------------------------------------------------------Storage Temperature Range --------------------------------------------------------------------------------------------
ESD Ratings
(Note 1)
(Note 2)
ESD Susceptibility
HBM (Human Body Model) ---------------------------------------------------------------------------------------------- 2kV
Recommended Operating Conditions
−0.3V to 65V
−0.3V to 65V
−0.6V to 65V
−5V to 70V
−7V to 70V
−0.3V to 65V
−0.3V to 6V
−0.3V to 6V
260°C
150°C
−65°C to 150°C
(Note 3)
Supply Input Voltage, VIN ----------------------------------------------------------------------------------------------- 4V to 60V
Output Voltage ------------------------------------------------------------------------------------------------------------- 0.8V to VIN
Junction Temperature Range -------------------------------------------------------------------------------------------- −40°C to 150°C
Thermal Information
(Note 4 and Note 5)
Thermal Parameter
WDFN-10L 4x4
SOP-8
(Exposed pad)
Unit
JA
Junction-to-ambient thermal resistance (JEDEC
standard)
31.7
30.4
C/W
JC(Top)
Junction-to-case (top) thermal resistance
46.4
73.9
C/W
JC(Bottom)
Junction-to-case (bottom) thermal resistance
4.1
3.4
C/W
JA(EVB)
Junction-to-ambient thermal resistance (specific
EVB)
30.4
30
C/W
JC(Top)
Junction-to-top characterization parameter
3.4
5
C/W
JB
Junction-to-board characterization parameter
13
13.2
C/W
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RTQ2965-QA
Electrical Characteristics
(VIN = 12V, TA = TJ = −40°C to 125°C, unless otherwise specified)
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
4
--
60
V
Supply Voltage
Input Operating Voltage
VIN
After soft-start is finished
VIN Under-Voltage Lockout
Threshold
VUVLOH
VIN rising
4.1
4.3
4.5
VUVLOL
VIN falling
3.8
3.9
4
Shutdown Current
ISHDN
VEN = 0V
--
2.25
5
A
Quiescent Current
IQ
VEN = 2V, VFB = 0.83V, not
switching
--
100
135
A
1.1
1.2
1.3
V
VIN = 12V, TA = 25C
--
540
--
s
VTH_EN + 50mV
--
4.6
--
A
VTH_EN 50mV
0.58
1.2
1.8
A
2.2
3.4
4.5
A
0.792
0.8
0.808
V
--
70
140
m
--
50
--
nA
Normal operation
2A < ICOMP < 2A
VCOMP = 1V
--
440
--
During SS,
2A < ICOMP < 2A
VCOMP = 1V, VFB = 0.4V
--
77
--
VFB = 0.8V
--
10000
--
V/V
--
2500
--
kHz
--
30
--
A
--
17
--
A/V
6.375
7.5
8.625
A
V
Enable Voltage
Enable Threshold Voltage
VTH_EN
Enable to COMP Active
Pull-Up Current
IEN
Hysteresis Current
IEN_Hys
Reference Voltage
Reference Voltage
VREF
Internal MOSFET
High-Side Switch OnResistance
RDS(ON)_H
VIN = 12V, VBOOT VSW = 5V
Error Amplifier
Input Current
Error Amplifier TransConductance
gm
Error Amplifier DC Gain
Error Amplifier Bandwidth
Source/Sink Current
COMP to Current Sense
Trans-Conductance
VCOMP = 1V, 100mV overdrive
gm_cs
A/V
Current Limit
Current Limit
ILIM
f SW = 500kHz,
VOUT = 5V
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
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is a registered trademark of Richtek Technology Corporation.
DSQ2965-QA-01 March 2021
RTQ2965-QA
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
IOUT = 1A
--
120
135
ns
On-Time Timer Control
Minimum On-Time
tON_MIN
Timing Resistor and External Clock
Switching Frequency 1
f SW1
RRT/SYNC = 1M
90
105
120
kHz
Switching Frequency 2
f SW2
RRT/SYNC = 200k
450
500
550
kHz
Switching Frequency 3
f SW3
RRT/SYNC = 37.4k
2200
2450
2700
kHz
0.3
--
2.2
MHz
--
20
--
ns
VIH_SYNC
--
1.55
2
VIL_SYNC
0.5
1.2
--
--
70
--
ns
VSS/TR = 0.4V, RTQ2965GQW
--
1.7
--
A
SS/TR to FB Offset
VSS/TR = 0.4V, RTQ2965GQW
--
42
--
mV
SS/TR-to-Reference Crossover
98% nominal, RTQ2965GQW
--
1.16
--
V
SS/TR Discharge Voltage
VFB = 0V, RTQ2965GQW
--
54
--
mV
10% to 90%, RTQ2965GSP
1.4
2
2.6
ms
VTH_PGLH1
VFB rising, % of VREF, PGOOD
from low to high, RTQ2965GQW
90
94
98
VTH_PGHL1
VFB rising, % of VREF, PGOOD
from high to low, RTQ2965GQW
105
109
113
VTH_PGHL2
VFB falling, % of VREF, PGOOD
from high to low, RTQ2965GQW
88
92
96
VTH_PGLH2
VFB falling, % of VREF, PGOOD
from low to high, RTQ2965GQW
102
106
110
VFB falling, RTQ2965GQW
--
2
--
%
VPGOOD = 5.5V, TA = 25°C,
RTQ2965GQW
--
10
500
nA
On-Resistance
IPGOOD = 3mA, VFB < 0.79V,
RTQ2965GQW
--
45
--
Minimum VIN for defined output
VPGOOOD < 0.5V, IPGOOD = 100A,
RTQ2965GQW
--
0.9
2
V
SYNC Frequency Range
External clock
Minimum Sync Pulse Width
SYNC Threshold Voltage
RT/SYNC Falling Edge to SW
Rising Edge Delay
V
Soft-Start and Tracking
Internal Charge Current
ISS
Internal Soft-Start Time
Soft-Start Period
Power Good
Power Good Threshold
Power Good Hysteresis
Power Good Leakage Current
ILK_PGOOD
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DSQ2965-QA-01 March 2021
%
is a registered trademark of Richtek Technology Corporation.
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13
RTQ2965-QA
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
SSP
--
+6
--
%
Thermal Shutdown
TSD
--
175
--
°C
Thermal Shutdown
Hysteresis
TSD
--
15
--
°C
--
0.4
--
V
Spread Spectrum
Spread-Spectrum Range
Thermal Shutdown
Output Under-Voltage Protection
UVP Trip Threshold
VUVP
UVP detect
Note 1. Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device.
These are stress ratings only, and functional operation of the device at these or any other conditions beyond those
indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating
conditions may affect device reliability.
Note 2. Devices are ESD sensitive. Handling precaution is recommended.
Note 3. The device is not guaranteed to function outside its operating conditions.
Note 4. For more information about thermal parameter, see the Application and Definition of Thermal Resistances report,
AN061.
Note 5. θJA(EVB), ψJC(Top) and ψJB are measured on a high effective-thermal-conductivity four-layer test board which is in size of
70mm x 50mm; furthermore, outer layers with 2 oz. Cu and inner layers with 1 oz. Cu. Thermal resistance/parameter
values may vary depending on the PCB material, layout, and test environmental conditions.
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
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is a registered trademark of Richtek Technology Corporation.
DSQ2965-QA-01 March 2021
RTQ2965-QA
Typical Application Circuit
300kHz, 3.3V, 5A Step-Down Converter
5V
DBOOT(*)
RTQ2965GSP
VIN
4.5V to 60V
2
C1
2.2µF
C2
2.2µF
C3
2.2µF
C4
2.2µF
VIN
C5
0.1µF
3
Enable Signal
BOOT
SW
R3
5.36k
6
CBOOT
0.1µF
L1
7.8µH
8
D1
SS5P6
EN
FB
C6
10nF
1
5
VOUT
3.3V/5A
R1
31.6k
C8
47µF
C9
47µF
C10
47µF
R2
10k
COMP
C7
220pF
RT/SYNC
4
RRT
332k
GND PAD
7
9 (Exposed pad)
fSW = 300kHz
L1 = 744325780
C8/C9/C10 = GRT32EC81C476KE13L
C1/C2/C3/C4 = HMK316AC7225KL-TE
400kHz, 5V, 5A Step-Down Converter
5V
DBOOT(*)
CBOOT
0.1µF
RTQ2965GSP
VIN
8V to 60V
2
C1
2.2µF
C2
2.2µF
C3
2.2µF
VIN
C5
0.1µF
C4
2.2µF
3
Enable Signal
BOOT
SW
R3
10.5k
6
D1
SS5P6
DSQ2965-QA-01 March 2021
R1
52.3k
C8
47µF
C9
47µF
C10
47µF
COMP
RT/SYNC
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
5
VOUT
5V/5A
R2
10k
C7
100pF
fSW = 400kHz
L1 = Cyntec-VCHA075D-6R8MS6
C8/C9/C10 = GRT32EC81C476KE13L
C1/C2/C3/C4 = HMK316AC7225KL-TE
L1
6.8µH
8
EN
FB
C6
8.2nF
1
4
RRT
249k
GND PAD
7
9 (Exposed pad)
is a registered trademark of Richtek Technology Corporation.
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15
RTQ2965-QA
400kHz, 12V, 5A Step-Down Converter
5V
DBOOT(*)
RTQ2965GSP
VIN
14V to 60V
2
C1
2.2µF
C3
2.2µF
C2
2.2µF
VIN
BOOT
C5
0.1µF
C4
2.2µF
3
Enable Signal
SW
L1
15µH
8
D1
SS5P6
EN
FB
C6
8.2nF
1
CBOOT
0.1µF
5
VOUT
12V/5A
R1
140k
C8
10µF
C9
10µF
C10
10µF
C11
10µF
C12
10µF
C13
10µF
R2
10k
R3
8.87k
6
COMP
C7
100pF
RT/SYNC
4
RRT
249k
GND PAD
7
9 (Exposed pad)
fSW = 400kHz
L1 = Cyntec-VCHA105D-150MS6
C8/C9/C10/C11/C12/C13 = UMK325AB7106KM
C1/C2/C3/C4 = HMK316AC7225KL-TE
300kHz, 3.3V, 5A Step-Down Converter
5V
DBOOT(*)
RTQ2965GQW
2
VIN
4.5V to 60V
C1
2.2µF
C2
2.2µF
C3
2.2µF
C4
2.2µF
C5
0.1µF
3
Enable Signal
10
PWRGD
RRT
332k
C6
10nF
VIN
R3
C7 5.36k
5
7
SW
PGOOD
1
L1
7.8µH
9
D1
SS5P6
EN
FB
6
COMP
fSW = 300kHz
L1 =744325780
C8/C9/C10 = GRT32EC81C476KE13L
C1/C2/C3/C4 = HMK316AC7225KL-TE
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
VOUT
3.3V/5A
R1
31.6k
C8
47µF
C9
47µF
C10
47µF
R2
10k
RT/SYNC
220pF
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16
BOOT
CBOOT
0.1µF
SS/TR 4
CSS
0.01µF
GND PAD
8
11 (Exposed pad)
is a registered trademark of Richtek Technology Corporation.
DSQ2965-QA-01 March 2021
RTQ2965-QA
400kHz, 5V, 5A Step-Down Converter
5V
DBOOT(*)
CBOOT
0.1µF
RTQ2965GQW
VIN
8V to 60V
2
C1
2.2µF
C3
2.2µF
C2
2.2µF
3
Enable Signal
10
RRT
249k
C6
BOOT
C5
0.1µF
C4
2.2µF
PWRGD
VIN
R3
8.2nF C7 10.5k
5
7
SW
1
L1
6.8µH
9
D1
SS5P6
EN
FB
PGOOD
6
VOUT
5V/5A
R1
52.3k
C8
47µF
C9
47µF
C10
47µF
R2
10k
RT/SYNC
SS/TR 4
COMP
CSS
0.01µF
GND PAD
8
11 (Exposed pad)
100pF
fSW = 400kHz
L1 = Cyntec-VCHA075D-6R8MS6
C8/C9/C10 = GRT32EC81C476KE13L
C1/C2/C3/C4 = HMK316AC7225KL-TE
400kHz, 12V, 5A Step-Down Converter
5V
DBOOT(*)
RTQ2965GQW
VIN
14V to 60V
2
C1
2.2µF
C2
2.2µF
C3
2.2µF
3
Enable Signal
10
RRT
249k
C6
BOOT
C5
0.1µF
C4
2.2µF
PWRGD
VIN
R3
8.2nF C7 8.87k
5
7
SW
1
CBOOT
0.1µF
L1
15µH
9
D1
SS5P6
EN
FB
PGOOD
6
100pF
R1
140k
C8
10µF
C9
10µF
C10
10µF
C11
10µF
C12
10µF
C13
10µF
R2
10k
RT/SYNC
COMP
VOUT
12V/5A
SS/TR 4
CSS
0.01µF
GND PAD
8
11 (Exposed pad)
fSW = 400kHz
L1 = Cyntec-VCHA105D-150MS6
C8/C9/C10/C11/C12/C13 = UMK325AB7106KM
C1/C2/C3/C4 = HMK316AC7225KL-TE
* : While the input voltage is below 5.5V or duty ratio is higher than 65%, an external bootstrap diode must be added
between an external 5V voltage supply and the BOOT pin to enhance the drive capability for the high-side MOSFET.
Refer to the Application Information for more details on the external bootstrap diode design.
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DSQ2965-QA-01 March 2021
is a registered trademark of Richtek Technology Corporation.
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17
RTQ2965-QA
Typical Operating Characteristics
Efficiency vs. Output Current
Efficiency vs. Output Current
100
100
90
90
70
60
50
40
VIN = 12V, VOUT = 5V
fSW = 2.5MHz, L= VCMT063T-1R5MN5, 1.5μH
fSW = 1MHz, L= VCHA075D-3R3MS6, 3.3μH
fSW = 400kHz, L= VCHA075D-6R8MS6, 6.8μH
fSW = 100kHz, L= 74435573300, 33μH
30
20
10
0
0.001
0.01
0.1
1
VIN
VIN
VIN
VIN
VIN
80
Freq = 100k
Freq = 400k
Freq = 1M
Freq = 2.5M
Efficiency (%)
Efficiency (%)
80
70
60
40
30
20
VOUT = 12V, fSW = 400kHz,
L = VCHA105D-150MS6, 15μH
10
0
0.001
10
0.01
Efficiency vs. Output Current
1
10
Efficiency vs. Output Current
100
90
VIN
VIN
VIN
VIN
VIN
VIN
VIN
VIN
70
60
50
40
30
=
=
=
=
=
=
=
=
80
8V
12V
13.5V
18V
24V
36V
48V
60V
Efficiency (%)
80
Efficiency (%)
0.1
Output Current (A)
90
20
70
VIN
VIN
VIN
VIN
VIN
VIN
VIN
VIN
VIN
60
50
40
30
20
VOUT = 5V, fSW = 400kHz
L = VCHA075D-6R8MS6, 6.8μH
10
0
0.001
0.01
0.1
1
VOUT = 3.3V, fSW = 300kHz
L = 744325780, 7.8μH
10
0
0.001
10
0.01
Output Current (A)
0.1
=
=
=
=
=
=
=
=
=
4.5V
8V
12V
13.5V
18V
24V
36V
48V
60V
1
10
Output Current (A)
Output Voltage vs. Input Voltage
Output Voltage vs. Output Current
5.06
5.15
5.05
Output Voltage (V)
5.10
Output Voltage (V)
14V
24V
36V
48V
60V
50
Output Current (A)
100
=
=
=
=
=
5.05
5.00
4.95
4.90
VIN = 12V, VOUT = 5V
5.04
5.03
5.02
5.01
5.00
4.99
IOUT = 5A, VOUT = 5V
4.98
4.85
0
1
2
3
4
Output Current (A)
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
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18
5
5
10
15
20
25
30
35
40
45
50
55
60
Input Voltage (V)
is a registered trademark of Richtek Technology Corporation.
DSQ2965-QA-01 March 2021
RTQ2965-QA
Current Limit vs. Input Voltage
Switching Frequency vs. Temperature
9.0
Switching Frequency (kHz)1
120
Current Limit (A)
8.5
8.0
7.5
7.0
6.5
115
110
105
100
95
VIN = 12V, VOUT = 5V
IOUT = 2.5A, RRT/SYNC = 1MΩ
VOUT = 5V, fSW = 500kHz, L = 5.6μH
6.0
90
6
12
18
24
30
36
42
48
54
60
-50
-25
0
Input Voltage (V)
Switching Frequency vs. Temperature
75
100
125
Switching Frequency vs. Temperature
2700
530
510
490
470
VIN = 12V, VOUT = 5V
IOUT = 2.5A, RRT/SYNC = 200kΩ
Switching Frequency (kHz)1
Switching Frequency (kHz)1
50
Temperature (°C)
550
450
2600
2500
2400
2300
VIN = 12V, VOUT = 5V
IOUT = 2.5A, RRT/SYNC = 37.4kΩ
2200
-50
-25
0
25
50
75
100
125
-50
-25
0
Temperature (°C)
25
50
75
100
125
Temperature (°C)
Quiescent Current vs. Temperature
Shutdown Current vs. Temperature
135
5.0
4.5
125
Shutdown Current (μA)1
Quiescent Current (μA)
25
115
105
95
85
75
VIN = 12V
65
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
VIN = 12V
0.0
-50
-25
0
25
50
75
100
Temperature (°C)
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DSQ2965-QA-01 March 2021
125
-50
-25
0
25
50
75
100
125
Temperature (°C)
is a registered trademark of Richtek Technology Corporation.
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19
RTQ2965-QA
UVLO Threshold vs. Temperature
Enable Threshold vs. Temperature
1.30
4.6
Enable Threshold (V)
UVLO Threshold (V)
5.0
Rising
4.2
Falling
3.8
3.4
1.26
1.22
1.18
1.14
VOUT = 1V
VOUT = 1V
3.0
1.10
-50
-25
0
25
50
75
100
125
-50
-25
0
Temperature (°C)
Output Voltage vs. Temperature
75
100
125
Current Limit vs. Temperature
9.0
High-Side MOSFET
8.5
Current Limit (A)
5.05
Output Voltage (V)
50
Temperature (°C)
5.10
5.00
4.95
4.90
4.85
8.0
7.5
7.0
6.5
6.0
VIN = 12V, VOUT = 5V
fSW = 500kHz, L= 5.6μH
5.5
VIN = 12V, VOUT = 5V, IOUT = 2.5A
4.80
-50
-25
0
25
50
75
100
5.0
125
-50
-25
0
25
50
75
Temperature (°C)
Temperature (°C)
Load Transient Response
Output Ripple Voltage
100
125
VOUT
(10mV/Div)
IOUT
(1A/Div)
VOUT
(200mV/Div)
25
VIN = 12V, VOUT = 5V
IOUT = 2.5 to 5A, fSW = 400kHz
COUT = 47μF x 3, L = 6.8μH
VSW
(5V/Div)
VIN = 12V, VOUT = 5V, IOUT = 1mA, fSW = 400kHz
Time (100μs/Div)
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Time (200μs/Div)
is a registered trademark of Richtek Technology Corporation.
DSQ2965-QA-01 March 2021
RTQ2965-QA
Output Ripple Voltage
Power On from EN
VIN = 12V, VOUT = 5V, IOUT =5A, fSW = 400kHz
VEN
(2V/Div)
VOUT
(10mV/Div)
VSS/TR
(1V/Div)
VIN = 12V, VOUT = 5V
IOUT = 5A, fSW = 400kHz
VOUT
(2V/Div)
VPGOOD
(5V/Div)
VSW
(5V/Div)
Time (4μs/Div)
Time (2ms/Div)
Power Off from EN
Power On from VIN
VEN
(2V/Div)
VSS/TR
(5V/Div)
VOUT
(2V/Div)
VIN = 12V, VOUT = 5V
IOUT = 5A, fSW = 400kHz
VPGOOD
(5V/Div)
VIN
(4V/Div)
VSS/TR
(2V/Div)
VIN = 12V, VOUT = 5V
IOUT = 5A, fSW = 400kHz
VOUT
(2V/Div)
VPGOOD
(5V/Div)
Time (200μs/Div)
Time (4ms/Div)
Power Off from VIN
Start-Up Dropout Performance
VIN
(4V/Div)
VIN
VSS/TR
(5V/Div)
VOUT
(2V/Div)
VIN = 12V, VOUT = 5V
IOUT = 5A, fSW = 400kHz
VPGOOD
(5V/Div)
VOUT
VIN
(2V/Div)
VOUT
(2V/Div)
VOUT = 5V, IOUT = 0.1A, 50Ω, EN pin floats
Time (4ms/Div)
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DSQ2965-QA-01 March 2021
Time (100ms/Div)
is a registered trademark of Richtek Technology Corporation.
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21
RTQ2965-QA
Start-Up Dropout Performance
Radiated EMI Performance with Peak Limits
VIN
VOUT
VIN
(2V/Div)
VOUT
(2V/Div)
VOUT = 5V, IOUT = 1A, 5Ω, EN pin floats
Time (100ms/Div)
Emission Level (dBμV/m)
60
CISPR25 Class 5 Peak Limit
Vertical Polarization
Horizontal Polarization
55
50
45
40
35
30
25
20
15
10
5
0
-5
VOUT = 5V, IOUT = 5A (1Ω)
with EMI Filter
0.1
1
10
100
1000
Frequency (MHz)
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
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is a registered trademark of Richtek Technology Corporation.
DSQ2965-QA-01 March 2021
RTQ2965-QA
Application Information
Switching Frequency Setting
The RTQ2965 offers adjustable switching frequency setting
and the switching frequency can be set by using external
resistor RRT/SYNC. The switching frequency range is from
100kHz to 2.5MHz. The selection of the operating
frequency is a trade-off between efficiency and component
size. High frequency operation allows the use of smaller
inductor and capacitor values. Operation at lower
frequencies improves efficiency by reducing internal gate
charge and transition losses, but requires larger inductance
values and/or capacitance to maintain low output ripple
voltage. The additional constraints on operating frequency
are the minimum on-time and minimum off-time. The
minimum on-time, tON_MIN, is the smallest duration of time
in which the high-side switch can be in its “on” state.
The minimum on-time of the RTQ2965 is 120ns (typically).
In continuous mode operation, the maximum operating
frequency, fSW_MAX, can be derived from the minimum ontime according to the formula below :
VOUT
fSW_MAX =
tON_MIN VIN_MAX
where VIN_MAX is the maximum operating input voltage.
The minimum off-time, tOFF_MIN, is the smallest amount of
time that the RTQ2965 is capable of tripping the current
comparator and turning the high-side MOSFET back off.
The minimum off-time of the RTQ2965 is 130ns (typically).
If the switching frequency should be constant, the required
off-time needs to be larger than minimum off-time. Below
shows minimum off-time calculation with loss terms
consideration :
VOUT + IOUT_MAX RL + VD
1
VIN_MIN IOUT_MAX RDS(ON)_H + VD
tOFF_MIN
fSW
where RDS(ON)_H is the on-resistance of the high-side
MOSFET; VD is the forward conduction voltage of the
freewheel diode; RL is the DC resistance of inductor.
The switching frequency fSW is set by the external resistor
RRT/SYNC connected between the RT/SYNC pin and
ground. The failure mode and effects analysis (FMEA)
consideration is applied to the RT/SYNC pin setting to
avoid abnormal switching frequency operation at failure
conditions. It includes failure scenarios of short-circuit to
ground and the pin is left open. The switching frequency
will be 900kHz (typically) when the RT/SYNC pin is
shorted to ground, and 240kHz (typically) when the pin is
left open. The equation below shows the relation between
setting frequency and the RRT/SYNC value.
120279
RRT/SYNC (k ) =
fSW1.033
where fSW (kHz) is the desired setting frequency. It is
recommended to use 1% tolerance or better, and the
temperature coefficient of 100 ppm or less resistors. Figure
4 shows the relationship between switching frequency and
the RRT/SYNC resistor.
1200
1000
RRT/SYNC (k Ω )
A general RTQ2965 application circuit is shown in typical
application circuit section. External component selection
is largely driven by the load requirement and begins with
the switching frequency selection by using external resistor
RRT/SYNC. Next, the inductor L, the input capacitor CIN,
the output capacitor COUT and freewheel diode are chosen.
Next, feedback resistors and compensation circuit are
selected to set the desired output voltage and crossover
frequency, and the bootstrap capacitor CBOOT can be
selected. Finally, the remaining optional external
components can be selected for functions such as the
EN, external soft-start, PGOOD, and synchronization.
800
600
400
200
0
0
500
1000
1500
2000
2500
f SW (kHz)
Figure 4. Switching Frequency vs. RRT/SYNC
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Inductor Selection
The inductor selection trade-offs among size, cost,
efficiency, and transient response requirements. Generally,
three key inductor parameters are specified for operation
with the device: inductance value (L), inductor saturation
current (ISAT), and DC resistance (DCR).
A good compromise between size and loss is a 30% peakto-peak ripple current to the IC rated current. The switching
frequency, input voltage, output voltage, and selected
inductor ripple current determines the inductor value as
follows :
V
(VIN VOUT )
L = OUT
VIN fSW IL
Larger inductance values result in lower output ripple
voltage and higher efficiency, but a slightly degraded
transient response. This results in additional phase lag in
the loop and reduces the crossover frequency. As the ratio
of the slope-compensation ramp to the sensed-current
ramp increases, the current-mode system tilts towards
voltage-mode control. Lower inductance values allow for
smaller case size, but the increased ripple lowers the
effective current limit threshold and increases the AC
losses in the inductor. It also causes insufficient slope
compensation and ultimately loop instability as duty cycle
approaches or exceeds 50%. When duty cycle exceeds
50%, below condition needs to be satisfied :
V
4 fSW OUT
L
A good compromise among size, efficiency, and transient
response can be achieved by setting an inductor current
ripple (ΔIL) with about 10% to 50% of the maximum rated
output current (5A).
To enhance the efficiency, choose a low-loss inductor
having the lowest possible DC resistance that fits in the
allotted dimensions. The inductor value determines not
only the ripple current but also the load-current value at
which DCM/CCM switchover occurs. The selected inductor
should have a saturation current rating greater than the
peak current limit of the device. The core must be large
enough not to saturate at the peak inductor current (IL_PEAK) :
V
(VIN VOUT )
IL = OUT
VIN fSW L
IL_PEAK = IOUT_MAX + 1 IL
2
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The current flowing through the inductor is the inductor
ripple current plus the output current. During power up,
faults, or transient load conditions, the inductor current
can increase above the calculated peak inductor current
level calculated above . In transient conditions, the inductor
current can increase up to the switch current limit of the
device. For this reason, the most conservative approach
is to specify an inductor with a saturation current rating
which is equal to or greater than the switch current limit
rather than the peak inductor current. It is recommended
to use shielded inductors for good EMI performance.
Input Capacitor Selection
Input capacitance, CIN, is needed to filter the pulsating
current at the drain of the high-side MOSFET. The CIN
should be sized to do this without causing a large variation
in input voltage. The peak-to-peak voltage ripple on input
capacitor can be estimated as equation below :
+ ESR IOUT
VCIN = D IOUT 1 D
CIN fSW
where
V
D = OUT
VIN
Figure 5 shows the CIN ripple current flowing through the
input capacitors and the resulting voltage ripple across
the capacitors.
For ceramic capacitors, the equivalent series resistance
(ESR) is very low, the ripple which is caused by ESR can
be ignored, and the minimum value of effective input
capacitance can be estimated as equation below :
CIN_MIN = IOUT_MAX
D 1 D
VCIN_MAX fSW
where ΔVCIN_MAX is maximum input ripple voltage.
VCIN
CIN Ripple Voltage
VESR = IOUT x ESR
(1-D) x IOUT
CIN Ripple Current
D x IOUT
D x tSW (1-D) x tSW
Figure 5. CIN Ripple Voltage and Ripple Current
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In addition, the input capacitor needs to have a very low
ESR and must be rated to handle the worst-case RMS
input current. The RMS ripple current (IRMS) of the regulator
can be determined by the input voltage (VIN), output voltage
(VOUT), and rated output current (IOUT) as the following
equation :
V
VIN
IRMS IOUT _MAX OUT
1
VIN
VOUT
From the above, the maximum RMS input ripple current
occurs at maximum output load, which will be used as
the requirements to consider the current capabilities of
the input capacitors. The maximum ripple voltage usually
occurs at 50% duty cycle, that is, VIN = 2 x VOUT. It is
common to use the worse IRMS ≅ 0.5 x IOUT_MAX at VIN = 2
x VOUT for design. Note that ripple current ratings from
capacitor manufacturers are often based on only 2000
hours of life which makes it advisable to further de-rate
the capacitor, or choose a capacitor rated at a higher
temperature than required.
Several capacitors may also be paralleled to meet size,
height and thermal requirements in the design. For low
input voltage applications, sufficient bulk input capacitance
is needed to minimize transient effects during output load
changes.
Ceramic capacitors are ideal for switching regulator
applications because of its small size, robustness, and
very low ESR. However, care must be taken when these
capacitors are used at the input. A ceramic input capacitor
combined with trace or cable inductance forms a high
quality (under damped) tank circuit. If the RTQ2965 circuit
is plugged into a live supply, the input voltage can ring to
twice its nominal value, possibly exceeding the device's
rating. This situation is easily avoided by placing the low
ESR ceramic input capacitor in parallel with a bulk
capacitor with higher ESR to damp the voltage ringing.
The input capacitor should be placed as close as possible
to the VIN pin with a low inductance connection to the
GND of the IC. The VIN pin must be bypassed to ground
with a minimum value of effective capacitance 3μF. For
400kHz switching frequency application, two 4.7μF, X7R
capacitors can be connected between the VIN pin and the
GND pin. The larger input capacitance is required when a
lower switching frequency is used. For filtering high
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frequency noise, an additional small 0.1μF capacitor should
be placed close to the part and the capacitor should be
0402 or 0603 in size. X7R capacitors are recommended
for best performance across temperature and input voltage
variations.
Output Capacitor Selection
The selection of COUT is determined by considering to
satisfy the voltage ripple and the transient loads. The peakto-peak output ripple, ΔVOUT, is determined by :
1
VOUT = IL ESR +
8 fSW COUT
Where the ΔIL is the peak-to-peak inductor ripple current.
The highest output ripple is at maximum input voltage
since ΔIL increases with input voltage. Multiple capacitors
placed in parallel may be needed to meet the ESR and
RMS current handling requirements.
Regarding to the transient loads, the VSAG and VSOAR
requirement should be taken into consideration for
choosing the effective output capacitance value. The
amount of output sag/soar is a function of the crossover
frequency factor at PWM, and can be calculated from
below equation :
1
VSAG = VSOAR = IOUT ESR +
2 COUT fC
Ceramic capacitors have very low equivalent series
resistance (ESR) and provide the best ripple performance.
The X7R dielectric capacitor is recommended for the best
performance across temperature and input voltage
variations. The variation of the capacitance value with
temperature, DC bias voltage and switching frequency
needs to be taken into consideration. For example, the
capacitance value of a capacitor decreases as the DC bias
across the capacitor increases. Be careful to consider the
voltage coefficient of ceramic capacitors when choosing
the value and case size. Most ceramic capacitors lose
50% or more of their rated values when used near their
rated voltage.
Transient performance can be improved with a higher value
output capacitor. Increasing the output capacitance will
also decrease the output voltage ripple.
Freewheel Diode Selection
When the high-side MOSFET turns off, inductor current
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is supplied through the external low-side diode, freewheel
diode, connected between the SW pin and GND.
The reverse voltage rating of freewheel diode should be
equal to or greater than the VIN_MAX. The maximum average
forward rectified current of freewheel diode should be equal
to or greater than the maximum load current. Considering
the efficiency performance, the diode must have a
minimum forward voltage and reverse recovery time. So
Schottky Diodes are recommended to be freewheel diode.
Restriction of Forward Voltage (V)
The selected forward voltage of Schottky Diode must be
less than the restriction of forward voltage in Figure 6 at
operating temperature range to avoid the IC malfunction.
output voltage is set according to the following equation :
R1
VOUT = VREF 1 +
R2
where the reference voltage VREF, is 0.8V (typically).
VOUT
R1
FB
RTQ2965
R2
GND
Figure 7. Output Voltage Setting
The placement of the resistive divider should be within
5mm of the FB pin. The resistance of R2 should not be
larger than 80kΩ for noise immunity consideration. The
resistance of R1 can then be obtained as below :
R2 (VOUT VREF )
R1 =
VREF
1.45
1.40
1.35
1.30
1.25
1.20
For better output voltage accuracy, the divider resistors
(R1 and R2) with ±1% tolerance or better should be used.
1.15
1.10
Compensation Network Design
1.05
1.00
-50
-25
0
25
50
75
100
125
150
Temperature (°C)
Figure 6. Restriction of Forward Voltage vs. Temperature
The losses of freewheel diode must be considered in order
to ensure sufficient power rating for diode selection. The
conduction loss in the diode is determined by the forward
voltage of the diode, and the switching loss in the diode
can be determined by the junction capacitor of the diode.
The power dissipation of the diode can be calculated as
following formula
V
PD = PD_CON + PD_SW = IOUT VD 1 OUT
V
IN
1
2
+ CJ VIN + VD fSW
2
where CJ is the junction capacitance of the freewheel diode.
Output Voltage Programming
The output voltage can be programmed by a resistive divider
from the output to ground with the midpoint connected to
the FB pin. The resistive divider allows the FB pin to sense
a fraction of the output voltage as shown in Figure 7. The
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The purpose of loop compensation is to ensure stable
operation while maximizing the dynamic performance. An
undercompensated system may result in unstable
operation. Typical symptoms of an unstable power supply
include: audible noise from the magnetic components or
ceramic capacitors, jittering in the switching waveforms,
oscillation of output voltage, overheating of power MOSFET
and so on.
In most cases, the peak current mode control architecture
used in the RTQ2965 only requires two external
components to achieve a stable design as shown in Figure
8. The compensation can be selected to accommodate
any capacitor type or value. The external compensation
also allows the user to set the crossover frequency and
optimize the transient performance of the device. At around
the crossover frequency, the peak current mode control
(PCMC) equivalent circuit of Buck converter can be
simplified as shown in Figure 9. The method presented
here is easy to calculate and ignore the effects of the
internal slope compensation. Since the slope
compensation is ignored, the actual crossover frequency
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is usually lower than the crossover frequency used in the
calculations. It is always necessary to make a
measurement before releasing the design for final
production. Though the models of power supplies are
theoretically correct, they cannot take full account of the
circuit parasitic and component nonlinearity, such as the
ESR variations of output capacitors, the nonlinearity of
inductors and capacitors, etc. Also, circuit PCB noise and
limited measurement accuracy may also cause
measurement errors. A Bode plot is ideally measured with
a network analyzer while Richtek application note AN038
provides an alternative way to check the stability quickly
and easily. Generally, follow the steps below to calculate
the compensation components :
1. Set up the crossover frequency, f C. For stability
purposes, the target is to have a loop gain slope that
is −20dB/decade from a very low frequency to beyond
the crossover frequency. In general, one-twentieth to
one-tenth of the switching frequency (5% to 10% of
fsw) is recommended to be the crossover frequency.
Do “NOT” design the crossover frequency over 80kHz
with the RTQ2965. For dynamic purposes, the higher
the bandwidth, the faster the load transient response.
The downside of the high bandwidth is that it increases
the susceptibility of the regulators to board noise which
ultimately leads to excessive falling edge jitter of the
switch node voltage.
2. RCOMP can be determined by :
2 fC VOUT COUT
2 fC COUT
RCOMP =
=
gm VREF gm_cs
gm gm_cs
R1 + R2
R2
where gm is the error amplifier gain of transconductance (440μA/V) ; gm_cs is COMP to current
sense trans-conductance (17A/V); the variation of COUT
with temperature, DC bias voltage and switching
frequency needs to be taken into consideration.
3. A compensation zero can be placed at or before the
dominant pole of buck which is provided by output
capacitor and maximum output loading (RL). Calculate
CCOMP :
CCOMP =
RL COUT
RCOMP
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4. The compensation pole is set to the frequency at the
ESR zero or 1/2 of the operating frequency. Output
capacitor and its ESR provide a zero, and optional
CCOMP2 can be used to cancel this zero.
R
COUT
CCOMP2 = ESR
RCOMP
If 1/2 of the operating frequency is lower than the ESR
zero, the compensation pole is set at 1/2 of the
operating frequency.
1
CCOMP2 =
fsw
RCOMP
2
2
Note: Generally, CCOMP2 is an optional component used
to enhance noise immunity.
COMP
RCOMP
CCOMP2
RTQ2965
(option)
CCOMP
GND
Figure 8. External Compensation Components
VOUT
RESR
gm_cs
RL
COUT
VCOMP
-
(option)
RCOMP
R1
EA
+
CCOMP2
VFB
VREF
R2
CCOMP
Figure 9. Simplified Equivalent Circuit of Buck with
PCMC
Bootstrap Driver Supply
The bootstrap capacitor (CBOOT) between the BOOT pin
and the SW pin is used to create a voltage rail above the
applied input voltage, VIN. Specifically, the bootstrap
capacitor is charged through an internal diode to an internal
voltage source each time when the low-side freewheel
diode conducts. The charge on this capacitor is then used
to supply the required current during the remainder of the
switching cycle. For most applications, a 0.1μF, 0603
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ceramic capacitor with X7R is recommended, and the
capacitor should have a 6.3 V or higher voltage rating.
External Bootstrap Diode
It has to add an external bootstrap diode between an
external 5V voltage supply and the BOOT pin to enhance
the drive capability for the high-side MOSFET and improve
efficiency when the input voltage is below 5.5V or duty
ratio is higher than 65%. The recommended application
circuit is shown in Figure 10. The bootstrap diode can be
a low-cost one, such as 1N4148. The external 5V can be
a fixed 5V voltage supply from the system, or a 5V output
voltage generated by the RTQ2965. Note that the VBOOT−
must be lower than 5.5V. Figure 11 shows efficiency
comparison between with and without bootstrap diode.
SW
5V
DBOOT
BOOT
CBOOT
0.1µF
RTQ2965
SW
Figure 10. External Bootstrap Diode
100
VIN = 4.5V, VOUT = 3.3V,
L = 744325780, 7.8μH,
fSW = 300kHz
98
Efficiency (%)
96
94
External Bootstrap Resistor (Option)
The gate driver of an internal high-side MOSFET, utilized
as a high-side switch, is optimized for turning on the
switch. The gate driver is not only fast enough for reducing
switching power loss, but also slow enough for minimizing
EMI. The EMI issue is worse when the switch is turned
on rapidly due to induced high di/dt noises. When the
high-side MOSFET is turned off, the SW node will be
discharged relatively slow by the inductor current because
the presence of the dead time when both the high-side
MOSFET and low-side freewheel diode are turned off.
In some cases, it is desirable to reduce EMI further, even
at the expense of some additional power dissipation. The
turn-on rate of the high-side MOSFET can be slowed by
placing a small bootstrap resistor RBOOT between the
BOOT pin and the external bootstrap capacitor as shown
in Figure 12. The recommended range for the RBOOT is
several ohms to 10 ohms, and it could be 0402 or 0603 in
size.
This will slow down the rates of the high-side switch turnon and the rise of VSW. In order to improve EMI performance
and enhancement of the internal high-side MOSFET, the
recommended application circuit is shown in Figure 13,
which includes an external bootstrap diode for charging
the bootstrap capacitor and a bootstrap resistor RBOOT
placed between the BOOT pin and the capacitor/diode
connection.
92
90
BOOT
88
With Bootstrap Diode (1N4148)
Without Bootstrap Diode
86
RBOOT
CBOOT
RTQ2965
SW
84
82
Figure 12. External Bootstrap Resistor at the BOOT Pin
80
0
1
2
3
4
5
5V
Output Current (A)
Figure 11. Efficiency Comparison between with and
without Bootstrap Diode
BOOT
RBOOT
DBOOT
CBOOT
RTQ2965
SW
Figure 13. External Bootstrap Diode and Resistor at the
BOOT Pin
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EN Pin for Start-Up and UVLO Adjustment
For automatic start-up, the EN pin has an internal pull-up
current source IEN (1.2μA, typically) that enables operation
of the RTQ2965 when the EN pin floats. If the EN voltage
rises above the VTH_EN (1.2V, typically) and the VIN voltage
is higher than VUVLOH (4.3V, typically), the device will be
turned on, that is, switching is enabled and soft-start
sequence is initiated. If the high UVLO is required, the
EN pin can be used to adjust the under-voltage lockout
(UVLO) threshold and hysteresis. There is an additional
hysteresis current source IEN_Hys (3.4μA, typically) which
is sourced out of the EN pin when the EN pin voltage
exceeds VTH_EN. When the EN pin drops below VTH_EN,
the IEN_Hys is removed. Therefore, the EN pin can be
externally connected to VIN by adding two resistors, RENH
and RENL to achieve UVLO adjustment as shown in Figure
14.
According to the desired start and stop input voltage, the
resistance of REN1 and REN2 can be obtained as below :
REN1 =
REN2
VStart VStop
IEN_Hys
VTH_EN
=
VStart VTH_EN
REN1
+ IEN
where IEN is the enable pull-up current source value before
the EN pin voltage exceeds the VTH_EN (1.2μA typically).
The EN pin, with high-voltage rating, supports wide input
voltage range to adjust the VIN UVLO.
VIN
REN1
EN
REN2
RTQ2965
GND
Figure 14. Resistive Divider for Under-Voltage Lockout
Threshold Setting
the SS/TR pin to ground or controlled by external ramp
voltage to SS/TR pin. An internal current source ISS (1.7μA,
typically) charges an external capacitor to build a softstart ramp voltage. The internal charging current source
ISS gradually increases the voltage on CSS/TR, and the highside MOSFET will start switching if voltage difference
between SS/TR pin and FB pin is equal to 42mV ( i.e.
VSS/TR − VFB = 42mV, typically) during power-up period.
The FB voltage will track the SS/TR pin ramp voltage with
a SS/TR to FB offset voltage (42mV, typically) during softstart interval. The typical soft-start time (tSS) which is the
duration of VOUT rises from 10% to 90% of setting value is
calculated as follows :
V
0.8
t SS = CSS/TR REF
ISS
If a heavy load is added to the output with large
capacitance, the output voltage will never enter regulation
because of UVP. Thus, the device remains in hiccup
operation. The CSS/TR should be large enough to ensure
soft-start period ends after COUT is fully charged.
ISS VOUT
CSS/TR COUT
0.8 ICOUT_CHG
where ICOUT_CHG is the COUT charge current which is
related to the switching frequency, inductance, high-side
MOSFET peak current limit and load current.
Power-Good Output
The RTQ2965GQW features an open-drain power-good
output (PGOOD) to monitor the output voltage status. The
PGOOD pin is an open-drain power-good indication output
and is to be connected to an external voltage source
through a pull-up resistor.
It is recommended to use pull-up resistance between the
values of 1 and 10kΩ to reduce the switching noise
coupling to PGOOD pin.
Synchronization
Soft-Start and Tracking Control
The RTQ2965GQW provides adjustable soft-start function.
The soft-start function is used to prevent large inrush
current while converter is being powered-up. The
RTQ2965GQW provides an SS/TR pin so that the softstart time can be programmed by selecting the value of
the external soft-start capacitor CSS/TR connected from
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The RTQ2965 can be synchronized with an external clock
ranging from 300kHz to 2.2MHz which is applied to the
RT/SYNC pin. The minimum synchronous pulse width of
the external clock must be larger than 20ns and the
amplitude should have valleys that are below 0.5V and
peaks above 2V (up to 6V). The rising edge of the SW will
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RTQ2965-QA
be synchronized to the falling edge of the RT/SYNC pin
signal.
The switching frequency control of the RTQ2965 will switch
from the RT resistor setting mode to the synchronization
mode when the external clock is applied to the RT/SYNC
pin. The RTQ2965 transitions from the RT resistor setting
mode to the synchronization mode within 60
microseconds. Figure 15 and Figure 16 show the device
synchronized to an external system clock in power saving
mode (PSM) and continuous conduction mode (CCM).
The sub-harmonic oscillation may occur for duty cycle
greater than 50% in CCM at synchronization mode. By
choosing a larger inductor, more slope compensation can
be achieved and the risk of such sub-harmonic oscillations
is eliminated.
The switching frequency of synchronization should be
equal to or higher than the frequency set with the RT
resistor. For example, if the switching frequency of
synchronization will be 500kHz and higher, the RRT/SYNC
should be selected for 500kHz. Be careful to design the
compensation network and inductance for switching
frequency controlled by both RT resistor setting mode
and the synchronization mode.
Figure 16. Synchronization Mode in CCM
Thermal Considerations
In many applications, the RTQ2965 does not generate
much heat due to its high efficiency and low thermal
resistance of its WDFN-10L 4x4 and SOP-8 (Exposed pad)
packages. However, in applications which the RTQ2965
runs at a high ambient temperature and high input voltage
or high switching frequency, the generated heat may
exceed the maximum junction temperature of the part.
The junction temperature should never exceed the
absolute maximum junction temperature TJ(MAX), listed
under Absolute Maximum Ratings, to avoid permanent
damage to the device. If the junction temperature reaches
approximately 175°C, the RTQ2965 stops switching the
high-side MOSFET until the temperature cools down by
15°C.
The maximum power dissipation can be calculated by
the following formula :
PD(MAX) = (TJ(MAX) − TA) / θJA(EFFECTIVE)
Figure 15. Synchronization Mode in PSM
where TJ(MAX) is the maximum allowed junction temperature
of the die. For recommended operating condition
specifications, the maximum junction temperature is
150°C. T A is the ambient operating temperature,
θJA(EFFECTIVE) is the system-level junction to ambient
thermal resistance. It can be estimated from thermal
modeling or measurements in the system.
The thermal resistance of the device strongly depends on
the surrounding PCB layout and can be improved by
providing a heat sink of surrounding copper ground. The
addition of backside copper with thermal vias, stiffeners,
and other enhancements can also help reduce thermal
resistance. Carefully select the freewheel diode to ensure
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that thermal performance will not be limited by the
freewheel diode.
If the application calls for a higher ambient temperature
and may exceed the recommended maximum junction
temperature of 150°C, care should be taken to reduce the
temperature rise of the part by using a heat sink or air
flow.
Note that the over-temperature protection is intended to
protect the device during momentary overload conditions.
The protection is activated outside of the absolute
maximum range of operation as a secondary fail-safe and
therefore should not be relied upon operationally.
Continuous operation above the specified absolute
maximum operating junction temperature may impair
device reliability or permanently damage the device.
Place freewheel diode, D1, and inductor, L1, as close to
the IC as possible to reduce the area size of the SW
exposed copper to reduce the electrically coupling from
this voltage.
Connect the feedback sense network behind via of output
capacitor.
Place the feedback components RFB1 / RFB2 / CFF near
the IC.
Place the compensation components RCP1 / CCP1 / CCP2
near the IC.
The RT/SYNC resistor, RRT/SYNC, should be placed as
close to the IC as possible because to the RT/SYNC
pin is sensitive to noise.
Figure 17 and Figure 18 are the RTQ2965GQW layout
examples.
Layout Guidelines
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the RTQ2965 :
Four-layer or six-layer PCB with maximum ground plane
is strongly recommended for good thermal performance.
Keep the traces of the main current paths wide and
short.
Place high frequency decoupling capacitor CIN5 as close
to the IC as possible to reduce the loop impedance and
minimize switch node ringing.
Place bootstrap capacitor, CBOOT, as close to the IC as
possible. Routing the trace with width of 20mil or wider.
Place multiple vias under the device near VIN and GND,
and close to input capacitors to reduce parasitic
inductance and improve thermal performance. To keep
thermal resistance low, extend the ground plane as much
as possible. Add thermal vias under and near the
RTQ2965 to additional ground planes within the circuit
board and on the bottom side.
The high frequency switching nodes, SW and BOOT,
should be as small as possible. Keep analog
components away from the SW and BOOT nodes.
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DSQ2965-QA-01 March 2021
is a registered trademark of Richtek Technology Corporation.
www.richtek.com
31
L1
COUT2
COUT1
COUT3
SW should be connected to
inductor / diode by wide and
short trace. Keep sensitive
components away from this
trace. Reducing area of SW
trace as possible.
D1
CIN1
CIN2
CIN3
CIN4
CIN5
CCP2
REN1
Input capacitors must be
placed as close to IC
VIN-GND as possible.
RCP1
REN2
CSS
RRT
RFB2
The exposed pad must be soldered to a large
GND plane and add 6 thermal vias with
0.25mm diameter on exposed pad for thermal
dissipation.
RFB1
CFF
CCP1
The feedback and compensation
components must be connected
as close to the device as possible.
Top Layer
Figure 17. Layout Guide for RTQ2965GQW (Top Layer)
Place the CBOOT on another layer and
connect by short trace. Keep sensitive
components away from this trace.
Bottom Layer
Figure 18. Layout Guide for RTQ2965GQW (Bottom Layer)
RTQ2965-QA
Outline Dimension
H
A
M
EXPOSED THERMAL PAD
(Bottom of Package)
Y
J
X
B
F
C
I
D
Dimensions In Millimeters
Symbol
Dimensions In Inches
Min
Max
Min
Max
A
4.801
5.004
0.189
0.197
B
3.810
4.000
0.150
0.157
C
1.346
1.753
0.053
0.069
D
0.330
0.510
0.013
0.020
F
1.194
1.346
0.047
0.053
H
0.170
0.254
0.007
0.010
I
0.000
0.152
0.000
0.006
J
5.791
6.200
0.228
0.244
M
0.406
1.270
0.016
0.050
X
2.000
2.300
0.079
0.091
Y
2.000
2.300
0.079
0.091
X
2.100
2.500
0.083
0.098
Y
3.000
3.500
0.118
0.138
Option 1
Option 2
8-Lead SOP (Exposed Pad) Plastic Package
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34
is a registered trademark of Richtek Technology Corporation.
DSQ2965-QA-01 March 2021
RTQ2965-QA
2
1
2
1
DETAIL A
Pin #1 ID and Tie Bar Mark Options
Note : The configuration of the Pin #1 identifier is optional,
but must be located within the zone indicated.
Symbol
Dimensions In Millimeters
Dimensions In Inches
Min
Max
Min
Max
A
0.700
0.800
0.028
0.031
A1
0.000
0.050
0.000
0.002
A3
0.175
0.250
0.007
0.010
b
0.250
0.350
0.010
0.014
D
3.900
4.100
0.154
0.161
D2
3.250
3.350
0.128
0.132
E
3.900
4.100
0.154
0.161
E2
2.550
2.650
0.100
0.104
0.800
e
L
0.350
0.031
0.450
0.014
0.018
W-Type 10L DFN 4x4 Package
Copyright © 2021 Richtek Technology Corporation. All rights reserved.
DSQ2965-QA-01 March 2021
is a registered trademark of Richtek Technology Corporation.
www.richtek.com
35
RTQ2965-QA
Footprint Information
Package
PSOP-8
Option1
Option2
Number of Pin
8
Footprint Dimension (mm)
P
A
B
C
D
1.27
6.80
4.20
1.30
0.70
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36
Sx
Sy
2.30
2.30
3.40
2.40
M
4.51
Tolerance
±0.10
is a registered trademark of Richtek Technology Corporation.
DSQ2965-QA-01 March 2021
RTQ2965-QA
Footprint Dimension (mm)
Package
Number of
Pin
P
A
B
C
D
Sx
Sy
M
V/W/U/XDFN4x4-10
10
0.80
4.80
3.10
0.85
0.40
3.40
2.70
3.60
Tolerance
±0.05
Richtek Technology Corporation
14F, No. 8, Tai Yuen 1st Street, Chupei City
Hsinchu, Taiwan, R.O.C.
Tel: (8863)5526789
Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should
obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot
assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be
accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third
parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.
DSQ2965-QA-01 March 2021
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