SC173AMLTRT

SC173A POWER MANAGEMENT Features               3A EcoSpeedTM Synchronous Step-Down Regulator with Automatic Power Save Description The SC173A is an integrated, synchronous 3A EcoSpeedTM step-down regulator, which incorporates Semtech’s advanced, patented adaptive on-time architecture to achieve best-in-class performance in dynamic point-of-load applications. The input voltage range is 3V to 5.5V with a programmable output voltage from 0.75V up to 95% x VIN. The device features low-RDS(ON) internal switches and automatic power save for high efficiency across the output load range. Adaptive on-time control provides programmable pseudo-fixed frequency operation and excellent transient performance. The switching frequency can be set from 200kHz to 1MHz - allowing the designer to reduce external LC filtering and minimize light load (standby) losses. Additional features include cycle-by-cycle current limit, soft start, input UVLO and output OV protection, and over temperature protection. The open-drain PGOOD pin provides output status. Standby current is less than 10μA when disabled. The device is available in a low profile, thermally enhanced MLPD-3x3mm 10-pin package. VIN: 3V to 5.5V VOUT: 0.75V to 95% x VIN IOUT: Up to 3A Low RDS(ON) Switches Up to 96% Peak Efficiency Enable High Threshold: 1V Compatible with Low Voltage Logic High Output Accuracy Small Ceramic Capacitors Power Good Pin (Open-Drain) Patented Adaptive On-Time Control: Excellent Transient Response Programmable Pseudo-fixed Frequency Fault Protection Features: Cycle-by-Cycle Current Limit Short Circuit Protection Over and Under Output Voltage Protection Over-Temperature Internal Soft start Smart Power Save Ultra-Small Lead-Free 3x3mm, 10-Pin MLPD Package Fully WEEE and RoHS Compliant Networking Equipment, Embedded Systems Medical Equipment, Office Automation Instrumentation, Portable Systems Consumer Devices (DTV, Set-top Box, ... ) 5V POL Converters Applications      Typical Application Circuit 3 to 5.5V VIN BST LX VOUT = 0.75V to 95% VIN SC173A VDD FB PGOOD Power Good Enable EN PGND TON AGND December 16 , 2010 © 2010 Semtech Corporation 1 SC173A Pin Configuration Ordering Information Device SC173AMLTRT(1) Top Mark 173A Package(2) MLPD-10 3x3 BST VIN LX PGND PGOOD VDD AGND TON EN FB SC173AEVB Evaluation Board Notes: 1) Available in tape and reel packaging only. A reel contains 3000 devices. 2) Available in lead-free packaging only. WEEE compliant and Halogen free. This component and all homogenous sub-components are RoHS compliant. 10 Pin MLPD θJA= 40°C/W. Marking Information TOP MARKING 173A yyww xxxx yyww = Date Code (Example: 0952) xxxx = Semtech Lot Number (Example: 3901) © 2010 Semtech Corporation 2 SC173A Absolute Maximum Ratings LX to GND(3)……………………… - 0.3(DC) to +6.0V(DC) Max VIN to PGND, EN to AGND …………………… -0.3 to +6.0V BST to LX ………………………………………… -0.3 to +6.0V BST to PGND…………………………………… -0.3 to +12V VDD to AGND, VOUT to AGND ………………… -0.3V to +6.0V FB, PGOOD, TON …………………………… -0.3 to VDD + 0.3V AGND to PGND………………………………… -0.3 to +0.3V Peak IR Reflow Temperature ……………….…………… 260°C ESD Protection Level(2) ………………………………………1kV Recommended Operating Conditions Supply Input Voltage……………………………… 3V to 5.5V Maximum Continuous Output Current …………………… 3A Thermal Information Storage Temperature …………………………… -60 to +150°C Maximum Junction Temperature ……………………… 150°C Operating Junction Temperature ……………… -40 to +125°C Thermal Resistance, Junction to Ambient(1) ………… 40°C/W Exceeding the above specifications may result in permanent damage to the device or device malfunction. Operation outside of the parameters specified in the Electrical Characteristics section is not recommended. NOTES(1) Calculated from package in still air, mounted to 3” x 4.5”, 4 layer FR4 PCB with thermal vias under the exposed pad per JESD51 standards. (2) Tested according to JEDEC standard JESD22-A114-B (3) Due to parasitic board inductance, the transient LX pin voltage at the point of measurement may appear larger than that which exists on silicon. The device is designed to tolerate the short duration transient voltages that will appear on the LX pin due to the deadtime diode conduction, for inductor currents up to the current limit setting of the device. See application section for details. Electrical Characteristics Parameter Input Supplies VIN, VDD Input Voltage VDD UVLO Threshold VDD UVLO Hysteresis Unless specified: VIN =5V, TA=+25°C for Typ, -40°C to +85°C for Min and Max, TJ < 125°C Symbol Conditions Min Typ Max Units 3 Rising UVLO V TH 2.75 100 EN= 0V 2.85 200 5 500 5.5 2.98 V V mV 15 μA VIN, VDD Supply Current IOUT=0A(1) Controller FB On-Time Threshold Frequency Programming Range FB Input Bias Current See RTON Calculation FB=VDD or 0V 0.7425 200 -1 0.75 0.7575 1000 +1 V kHz μA Timing On-Time Minimum On-Time(1) Minimum Off-Time(1) © 2010 Semtech Corporation In Continuous Conduction VIN=5V, VOUT=3V, RTON=200kΩ 2.7 3 80 250 3.3 μs ns ns 3 SC173A Electrical Characteristics (continued) Unless specified: VIN =5V, TA=+25°C for Typ, -40°C to +85°C for Min and Max, TJ < 125°C Parameter Power Good Symbol Conditions Min Typ Max Units Power Good Signal Threshold High Power Good Threshold Power Good Signal Threshold Low VDD=3V PGOOD Delay Time(1) VDD=5V Noise Immunity Delay Time Leakage Power Good On-Resistance 116 86 120 90 1 124 %VOUT 93 ms 2 5 1 10 20 µs µA Ω Fault Protection Output Under-Voltage Fault Output Over-Voltage Fault Smart PowerSave Protection Threshold OV, UV Fault Noise Immunity Delay Over-Temperature Shutdown OT Latched FB with Respect to REF, 8 Consecutive Clocks FB with Respect to REF FB with Respect to REF -30 +16 +7 -25 +20 +10 5 150 -20 +24 +13 % % % μs °C Enable Output Enabled Output Disabled EN Input Bias Current Enable Pin Floating Voltage EN = VDD or 0V EN floating 39 0.5 41 1 0.4 8.0 44 V V μA %VDD © 2010 Semtech Corporation 4 SC173A Electrical Characteristics (continued) Unless specified: VIN =5V, TA=+25°C for Typ, -40°C to +85°C for Min and Max, TJ < 125°C Parameter Gate Drivers BST Switch On resistance Symbol Conditions Min Typ Max Units 25 45 Ω Internal Power MOSFETs Valley Current Limit, VDD=5V Current Limit Valley Current Limit, VDD=3V LX Leakage Current Switch Resistance Low Side Non-overlap time (1) Note: (1) Typical value from EVB, not ATE tested. 50 30 75 ns VIN=5.5V, LX=0V, High Side High Side 3 3.5 1 60 10 85 mΩ µA 3.5 A © 2010 Semtech Corporation 5 SC173A Pin Descriptions (MLPD-10) Pin # 1 2 3 4 5 6 7 8 9 10 Pin Name BST VIN LX PGND PGOOD FB EN TON AGND VDD PAD Pin Function Bootstrap pin. A capacitor is connected between BST to LX to develop the floating voltage for the high-side gate drive. Power input supply voltage. Switching (Phase) node. Power ground. Open-drain Power Good indicator. High impedance indicates power is good. An external pull-up resistor is required. Feedback input for switching regulator. Connect to an external resistor divider from the output to program the output voltage. Enable input for the switching regulator. Pull EN above 1V or float it to enable the part with automatic power save mode enabled. Connect EN to AGND to disable the switching regulator. On-time set input. Set the on-time by a series resistor to AGND. Analog Ground. Input power for internal control circuit. Needs at least 2.2mF decoupling capacitor from this pin to AGND. Thermal pad for heatsinking purposes. Connect to ground plane using multiple vias. Not connected internally. © 2010 Semtech Corporation 6 SC173A Block Diagram 10 VDD 9 AGND Reference 7 5 PGOOD Control 1 EN VDD BST VIN 2 Soft Start FB On-Time Generator Gate Drive Control VDD LX 3 6 8 TON PGND Zero-Cross Valley Current Limit R 4 © 2010 Semtech Corporation 7 SC173A Typical Characteristics Efficiency vs Output Current 100.0 3.339 3.330 Output Voltage vs Output Current Vin=5V, Vo=3.305V, LOUT: DS84LC-B1015AS-2R2N COUT=22mFx2 RTON = 80.6kOhm Output Voltage (V) 3.00 95.0 Efficiency (%) 3.322 3.313 3.304 3.296 3.287 3.279 3.270 90.0 85.0 Vin=5V, Vo=3.3V, LOUT: DS84LC-B1015AS-2R2N 80.0 COUT=22mFx2 RTON = 80.6kOhm 75.0 0.00 0.75 Output Current (A) 1.50 2.25 0.00 0.75 Output Current (A) 1.50 2.25 3.00 Efficiency vs Output Current 95.0 90.0 1.218 Output Voltage vs Output Current Efficiency (%) 85.0 80.0 75.0 70.0 65.0 60.0 55.0 0.01 Vin=5V, Vo=1.2V, LOUT: DS84LC-B1015AS-2R2N COUT=22mF RTON = 54.9kOhm Output Voltage (V) 1.216 Vin=5V, Vo=1.212V, LOUT: DS84LC-B1015AS-2R2N COUT=22mF RTON = 54.9kOhm 1.214 1.212 1.210 0.76 Output Current (A) 1.51 2.26 3.01 0.00 0.60 1.20 1.80 2.40 3.00 Output Current (A) Efficiency vs Output Current 95.0 90.0 1.216 Output Voltage vs Output Current Vo=1.212V, RTON = 54.9kOhm LOUT: DS84LC-B1015AS-2R2N COUT:22mF Output Voltage (V) Efficiency (%) 85.0 80.0 75.0 70.0 65.0 0.00 1.214 1.211 1.209 1.206 1.204 1.201 0.00 Vo=1.2V, RTON = 54.9kOhm LOUT:DS84LC-B1015AS-2R2N COUT:22mF Red:Vin = 3.5V Green: Vin = 4.0V Blue: Vin = 5.0V Black: Vin=5V Red: Vin=4V Blue: Vin=3.5V 0.60 Output Current (A) 1.20 1.80 2.40 3.00 0.75 Output Current (A) 1.50 2.25 3.00 © 2010 Semtech Corporation 8 SC173A Typical Characteristics FB Voltage vs Temperature 0.755 Start up waveform ( VIN=5V, VOUT=1.2V, IOUT=3A, Channel 1: 500mV/Div, Channel 4: 1A/Div, Time: 1ms/Div ) 0.753 FB Voltage (V) 0.750 Black:VDD=5.0V Red: VDD=3.0V 0.748 0.745 -40 -15 10 35 60 85 110 135 Temperature (°C) Load Transient Test (VIN=5V, VOUT=1.2V, IOUT= 0A to 3A, LOUT=1.0 H,COUT=2x22 F,Channel 1: 50mV/Div, Channel 2:5V/Div,Channel 4:2A/Div,Time:20 s/Div) IVIN Input Current In Shutdown vs Temperature 3.5 IVIN Input Current In Shutdown ( A) 3.0 2.5 2.0 1.5 1.0 0.5 0.0 -50 -25 0 25 50 75 100 125 Temperature (°C) VIN =5V Blue: VLX=GND Black: VLX=VIN Load Transient Test (VIN=5V, VOUT=1.2V, IOUT= 3A to 0A, LOUT=1.0 H, COUT=2x22 F, Channel 1: 50mV/Div,Channel 2:5V/Div,Channel 4:2A/Div, Time:20 s/Div) IBST Leakage Current vs Temperature 0.5 0.0 -0.5 -1.0 -1.5 VIN =5V I BST Leakage Current ( A) -2.0 -2.5 -3.0 -50 -25 0 VBST=V IN 25 50 75 100 125 Temperature (°C) © 2010 Semtech Corporation 9 SC173A Typical Characteristics 58 54 On-State Resistance (mΩ ) Low Side Switch On-State Resistance vs T e mperature 64 60 On-State Resistance (mΩ ) High Side Switch On-State Resistance vs T e mperature 50 46 42 38 34 30 -40 -15 10 35 60 85 110 135 Temperature (°C) 56 52 48 44 40 -40 -15 10 35 60 85 110 135 Temperature (°C) Blue: VDD=3.0V Black: VDD=5.0V Blue: VDD=3.0V Black: VDD=5.0V © 2010 Semtech Corporation 10 SC173A Applications Information SC173A Synchronous Buck Converter The SC173A is a step down synchronous buck dc-dc regulator. The SC173A is capable of 3A operation at very high efficiency in a tiny 3x3-10 pin package. The programmable operating frequency range of 200kHz – 1MHz (continuous conduction mode) enables the user to optimize the solution for minimum board space and optimum efficiency. The buck regulator employs pseudo-fixed frequency adaptive on-time control. This control scheme allows fast transient response thereby lowering the size of the power components used in the system. Input Voltage Range The SC173A can operate with an input voltage ranging from 3V to 5.5V. Psuedo-fixed Frequency Adaptive On-time Control The PWM control method used by the SC173A is pseudofixed frequency, adaptive on-time, as shown in Figure 1. The ripple voltage generated at the output capacitor ESR is used as a PWM ramp signal. This ripple is used to trigger the on-time of the controller. VIN CIN Q1 L VLX Q2 ESR TON VFB VLX FB threshold VOUT The adaptive on-time is determined by an internal oneshot timer. When the one-shot is triggered by the output ripple, the device sends a single on-time pulse to the high-side MOSFET. The pulse period is determined by VOUT and VIN; the period is proportional to output voltage and inversely proportional to input voltage. With this adaptive on-time arrangement, the device automatically anticipates the on-time needed to regulate VOUT for the present VIN condition and at the selected frequency. The advantages of adaptive on-time control are: • • • • • Predictable operating frequency compared to other variable frequency methods. Reduced component count by eliminating the error amplifier and compensation components. Reduced component count by removing the need to sense and control inductor current. Fast transient response — the response time is controlled by a fast comparator instead of a typically slow error amplifier. Reduced output capacitance due to fast transient response One-Shot Timer and Operating Frequency The one-shot timer operates as shown in Figure 2. The FB Comparator output goes high when VFB is less than the internal 750mV reference. This feeds into the gate drive and turns on the high-side MOSFET, and also starts the one-shot timer. The one-shot timer uses an internal comparator, timing capacitor, and a low pass filter (LPF) which regenerates VOUT from LX. One comparator input is connected to the filtered LX voltage, the other input is connected to the capacitor. When the on-time begins, the internal capacitor charges from zero volts through a current which is proportional to VIN. When the capacitor voltage reaches VOUT, the on-time is completed and the high-side MOSFET turns off. This method automatically produces an on-time that is proportional to VOUT and inversely proportional to VIN. Under steady-state operation conditions, the switching frequency can be determined from the on-time by the following equation. VOUT fSW = TON × VIN 11 COUT + FB Figure 1 — PWM Control Method, VOUT Ripple © 2010 Semtech Corporation SC173A Applications Information (continued) FB REF + Enable Input The EN input is used to enable or disable the switching regulator. When EN is low (grounded), the switching regulator is off and in its lowest power state. When off, the output power switches are tri-stated. FB PWM S R Q Hi-Side and Lo-Side Gate Drivers VIN Q1 VLX Q2 LPF L ESR COUT + VOUT VOUT VIN On-Shot Timing Generator Time = K x VOUT/VIN When EN is pulled high (above 1V), or permitted to float, the switching regulator turns on with automatic power save enabled. Smart Power Save Protection Active loads may leak current from a higher voltage into the switcher output. Under light load conditions with power save enabled, this can force VOUT to slowly rise and reach the over-voltage threshold, resulting in a hard shutdown. Smart power save prevents this condition. When the FB voltage exceeds 10% above nominal (exceeds 825mV), the device immediately disables powe save, and DL drives high to turn on the low-side MOSFET. This draws current from VOUT through the inductor and causes VOUT to fall. When VFB drops back to the 750mV trip point, a normal TON switching cycle begins. This method prevents a hard OVP shutdown and also cycles energy from VOUT back to VIN. Figure 4 shows typical waveforms for the Smart Power Save feature. VOUT drifts up to due to leakage current flowing into COUT Smart Power Save Threshold (825mV) FB threshold DH and DL off High-side Drive (DH) Single DH on-time pulse after DL turn-off Low-side Drive (DL) DL turns on when Smart PSAVE threshold is reached DL turns off when FB threshold is reached Normal DL pulse after DH on-time pulse RTON Figure 2 — On-Time Generation The SC173A uses an external resistor to set the on-time which indirectly sets the frequency. The on-time can be programmed to provide operating frequency from 200kHz to 1MHz using a resistor between the TON pin and ground. The resistor value is selected by the following equation. R TON = 1 25pF ⋅ fSW VOUT Voltage Selection The switcher output voltage is regulated by comparing VOUT as seen through a resistor divider at the FB pin to the internal 750mV reference voltage, see Figure 3. VOUT R1 R2 To FB pin VOUT discharges via inductor and low-side MOSFET Normal VOUT ripple Figure 3 — Output Voltage Selection Note that this control method regulates the valley of the output ripple voltage, not the DC value. The DC output voltage VOUT is offset by the output ripple according to the following equation.  R V VOUT = 0.75V ⋅  1 + 1  + RIPPLE  R2  2   Figure 4 — Smart Power Save Current Limit Protection The device features fixed current limiting, which is accomplished by using the RDS(ON) of the lower MOSFET for current sensing. While the low-side MOSFET is on, the © 2010 Semtech Corporation 12 SC173A inductor current flows through it and creates a voltage across the RDS(ON). During this time, the voltage across the MOSFET is negative with respect to ground. During this time, If this MOSFET voltage drop exceeds the internal reference voltage, the current limit will activate. The current limit then keeps the low-side MOSFET on and will not allow another high side on-time, until the current in the low-side MOSFET reduces enough to drop below the internal reference voltage once more. This method regulates the inductor valley current at the level shown by ILIM in Figure 5. Inductor Current IPEAK ILOAD ILIM During soft start the regulator turns off the low-side MOSFET on any cycle if the inductor current falls to zero. This prevents negative inductor current, allowing the device to start into a pre-biased output. Power Good Output The power good (PGOOD) output is an open-drain output which requires a pull-up resistor. When the output voltage is 10% below the nominal voltage, PGOOD is pulled low. It is held low until the output voltage returns to the nominal voltage. PGOOD is held low during soft start and activated approximately 1ms after VOUT reaches regulation. The total PGOOD delay is typically 2ms. PGOOD will transition low if the VFB pin exceeds +20% of nominal, which is also the over-voltage shutdown threshold (900mV). PGOOD also pulls low if the EN pin is low when VDD is present. Output Over-Voltage Protection Over-Voltage Protection (OVP) becomes active as soon as the device is enabled. The threshold is set at 750mV + 20% (900mV). When VFB exceeds the OVP threshold, DL latches high and the low-side MOSFET is turned on. DL remains high and the controller remains off, until the EN input is toggled or VDD is cycled. There is a 5μs delay built into the OVP detector to prevent false transitions. PGOOD is also low after an OVP event. Output Under-Voltage Protection Time Figure 5 — Valley Current Limit Setting the valley current limit to a value of ILIM results in a peak inductor current of ILIM plus the peak-to-peak ripple current. In this situation, the average (load) current through the inductor will be ILIM plus one half the peakto-peak ripple current. Soft start of PWM Regulator Soft start is achieved in the PWM regulator by using an internal voltage ramp as the reference for the FB comparator. The voltage ramp is generated using an internal charge pump which drives the reference from zero to 750mV in ~1.8mV increments, using an internal ~500kHz oscillator. When the ramp voltage reaches 750mV, the ramp is ignored and the FB comparator switches over to a fixed 750mV threshold. During soft start the output voltage tracks the internal ramp, which limits the start-up inrush current and provides a controlled soft start profile for a wide range of applications. Typical soft start ramp time is 0.85ms. When VFB falls to 75% of its nominal voltage (falls to 562.5mV) for eight consecutive clock cycles, the switcher is shut off and the DH and DL drives are pulled low to turn off the MOSFETs. The controller stays off until EN is toggled or VDD is cycled. VDD UVLO, and POR Under-Voltage Lock-Out (UVLO) circuitry inhibits switching and tri-states the power FETs until VDD rises above 2.9V. An internal Power-On Reset (POR) occurs when VDD exceeds 2.9V, which resets the fault latch and soft start counter to begin the soft start cycle. The SC173A then begins a soft start cycle. The PWM will shut off if VDD falls below 2.7V. 13 © 2010 Semtech Corporation SC173A Applications Information (continued) setting the frequency) using the following equation. 1 25pF ⋅ fSW Design Procedure When designing a switch mode supply the input voltage range, load current, switching frequency, and inductor ripple current must be specified. The maximum input voltage (VINMAX) is the highest specified input voltage. The minimum input voltage ( VINMIN) is determined by the lowest input voltage after evaluating the voltage drops due to connectors, fuses, switches, and PCB traces. The following parameters define the design. R TON = Calculating RTON results in the following solution. RTON=50kW, we use RTON=49.9kW in real application. Inductor Selection In order to determine the inductance, the ripple current must first be defined. Low inductor values result in smaller size but create higher ripple current which can reduce efficiency. Higher inductor values will reduce the ripple current/voltage and for a given DC resistance are more efficient. However, larger inductance translates directly into larger packages and higher cost. Cost, size, output ripple, and efficiency are all used in the selection process. The ripple current will also set the boundary for power save operation. The switching will typically enter power save mode when the load current decreases to 1/2 of the ripple current. For example, if ripple current is 3A then power save operation will typically start for loads less than 1.5A. If ripple current is set at 40% of maximum load current, then power save will start for loads less than 20% of maximum current. The inductor value is typically selected to provide a ripple current that is between 25% to 50% of the maximum load current. This provides an optimal trade-off between cost, efficiency, and transient performance. During the DH on-time, voltage across the inductor is (VIN - VOUT). The equation for determining inductance is shown next. TON = VOUT VINMAX ⋅ fSW • • • • Nominal output voltage (VOUT) Static or DC output tolerance Transient response Maximum load current (IOUT) There are two values of load current to evaluate — continuous load current and peak load current. Continuous load current relates to thermal stresses which drive the selection of the inductor and input capacitors. Peak load current determines instantaneous component stresses and filtering requirements such as inductor saturation, output capacitors, and design of the current limit circuit. The following values are used in this design. VIN = 5V + 10% VOUT = 1.0V + 4% fSW = 800kHz Load = 3A maximum • • • • Frequency Selection Selection of the switching frequency requires making a trade-off between the size and cost of the external filter components (inductor and output capacitor) and the power conversion efficiency. The desired switching frequency is 800kHz which results from using components selected for optimum size and cost . A resistor (RTON) is used to program the on-time (indirectly © 2010 Semtech Corporation L= (VIN - VOUT) × TON IRIPPLE 14 SC173A Applications Information (continued) Example In this example, the inductor ripple current is set equal to 30% of the maximum load current. Therefore ripple current will be 30% x 3A or 0.9A. To find the minimum inductance needed, use the VIN and TON values that correspond to VINMAX. TON_VINMAX = 1V = 227ns 5.5V ⋅ 800kHz The maximum ripple current of 0.511A creates a ripple voltage across the ESR. The maximum ESR value allowed is shown by the following equations. ESR MAX = IRIPPLEMAX VRIPPLE = 40mV 0.51A ESRMAX = 78.3 mΩ The output capacitance is chosen to meet transient requirements. A worst-case load release, from maximum load to no load at the exact moment when inductor current is at the peak, determines the required capacitance. If the load release is instantaneous (load changes from maximum to zero in < 1µs), the output capacitor must absorb all the inductor’s stored energy. This will cause a peak voltage on the capacitor according to the following equation. COUTMIN = L × (IOUT + 1 × IRIPPLEMAX )2 2 (VPEAK )2 - (VOUT )2 L= (5.5V - 1V) • 227ns = 1.14mH 0.9A A larger value of 2µH is selected. This will decrease the maximum IRIPPLE to 0.511A. Note that the inductor must be rated for the maximum DC load current plus 1/2 of the ripple current. The ripple current under minimum VIN conditions is also checked using the following equations. TON_VINMIN = 1V = 277ns 4.5V × 800kHz (VIN - VOUT ) × TON L Assuming a peak voltage VPEAK of 1.050V (50mV rise upon load release), and a 3A load release, the required capacitance is shown by the next equation. 2 m H × (3A + 1 × 0.511A) 2 (1.05V) 2 - (1.0V) 2 2 IRIPPLE = IRIPPLE_VINMIN = COUT MIN = = 207 m F (4.5V - 1V) × 277ns = 0.485A 2mH Capacitor Selection The output capacitors are chosen based on required ESR and capacitance. The maximum ESR requirement is controlled by the output ripple requirement and the DC tolerance. The output voltage has a DC value that is equal to the valley of the output ripple plus 1/2 of the peakto-peak ripple. Change in the output ripple voltage will lead to a change in DC voltage at the output. The design goal is for the output voltage regulation to be ±4% under static conditions. The internal 750mV reference tolerance is 1%. Assuming a 1% tolerance from the FB resistor divider, this allows 2% tolerance due to VOUT ripple. Since this 2% error comes from 1/2 of the ripple voltage, the allowable ripple is 4%, or 40mV for a 1V output. © 2010 Semtech Corporation If the load release is relatively slow, the output capacitance can be reduced. At heavy loads during normal switching, when the FB pin is above the 750mV reference, the DL output is high and the low-side MOSFET is on. During this time, the voltage across the inductor is approximately -VOUT. This causes a down-slope or falling di/dt in the inductor. If the load di/dt is not much faster than the -di/dt in the inductor, then the inductor current will tend to track the falling load current. This will reduce the excess inductive energy that must be absorbed by the output capacitor, therefore a smaller capacitance can be used. The following can be used to calculate the needed capacitance for a given dILOAD/dt. Peak inductor current is shown by the next equation. 15 SC173A Applications Information (continued) 1 ILPK = 3A + × 0.511A = 3.26A 2 increase the ESR of the output capacitors. It is also imperative to provide a proper PCB layout as discussed in the Layout Guidelines section. Another way to eliminate doubling-pulsing is to add a small (~ 10pF) capacitor across the upper feedback resistor, as shown in Figure 6. This capacitor should be left unpopulated unless it can be confirmed that doublepulsing exists. Adding the CTOP capacitor will couple more ripple into FB to help eliminate the problem. An optional connection on the PCB should be available for this capacitor. CTOP Rate of change of load current is dI LOAD 0.6A = dt 1m s IMAX = maximum load release = 3A L× COUT = ILPK × ILPK I - MAX × dt VOUT dILOAD 2 × ( VPK - VOUT ) 3.26A 3A × 1m s 1V 0.6A 2 × (1.05V - 1V) VOUT R1 R2 To FB pin C OUT = 3.26A × − 2m H × C OUT = 50 m F Figure 6 — Capacitor Coupling to FB Pin ESR loop instability is caused by insufficient ESR. The details of this stability issue are discussed in the ESR Requirements section. The best method for checking stability is to apply a zero-to-full load transient and observe the output voltage ripple envelope for overshoot and ringing. Ringing for more than one cycle after the initial step is an indication that the ESR should be increased. One simple way to solve this problem is to add trace resistance in the high current output path. A side effect of adding trace resistance is a decrease in load regulation. Note that COUT is much smaller in this example, 50µF compared to 207µF based upon a worst-case load release. To meet the two design criteria of minimum 50µF and maximum 78mΩ ESR, select two capacitors rated at 33µF and 15mΩ ESR or less. It is recommended that an additional small capacitor be placed in parallel with COUT in order to filter high frequency switching noise. Stability Considerations Unstable operation is possible with adaptive on-time controllers, and usually takes the form of double-pulsing or ESR loop instability. Double-pulsing occurs due to switching noise seen at the FB input or because the FB ripple voltage is too low. This causes the FB comparator to trigger prematurely after the minimum off-time has expired. In extreme cases the noise can cause three or more successive on-times. Double-pulsing will result in higher ripple voltage at the output, but in most applications it will not affect operation. This form of instability can usually be avoided by providing the FB pin with a smooth, clean ripple signal that is at least 10mVp-p, which may dictate the need to © 2010 Semtech Corporation ESR Requirements A minimum ESR is required for two reasons. One reason is to generate enough output ripple voltage to provide 10mVp-p at the FB pin (after the resistor divider) to avoid double-pulsing. The second reason is to prevent instability due to insufficient ESR. The on-time control regulates the valley of the output ripple voltage. This ripple voltage is the sum of the two voltages. One is the ripple generated by the ESR, the other is the ripple due to capacitive charging 16 SC173A Applications Information (continued) and discharging during the switching cycle. For most applications, the total output ripple voltage is dominated by the output capacitors, typically SP or POSCAP devices. For stability the ESR zero of the output capacitor should be lower than approximately one-third the switching frequency. The formula for minimum ESR is shown by the following equation. ESR MIN 3 = 2 × π × COUT × fSW Output Voltage Dropout The output voltage adjustable range for continuousconduction operation is limited by the fixed 320ns (typical) minimum off-time. When working with low input voltages, the duty-factor limit must be calculated using worst-case values for on and off times. The duty-factor limitation is shown by the next equation. DUTY TON(MIN) TON(MIN) TOFF(MAX ) Using Ceramic Output Capacitors When applications use ceramic output capacitors, the ESR is normally too small to meet the previously stated ESR criteria. In these applications it is necessary to add a small signal injection network as shown in Figure 7. In this network RL and CL filter the LX switching waveform to generate an in-phase ripple voltage comparable to the ripple seen on higher ESR capacitors. CC is a coupling capacitor used to AC couple the generated ripple onto the FB pin. Capacitor CFF is required for min COUT applications. This capacitor introduces a lead/lag into the control with the maximum phase placed at 1/2 fSW for added stability. VIN Q1 VLX RL Q2 CL CC L CFF The inductor resistance and MOSFET on-state voltage drops must be included when performing worst-case dropout duty-factor calculations. System DC Accuracy — VOUT Controller Three factors affect VOUT accuracy: the trip point of the FB error comparator, the ripple voltage variation with line and load, and the external resistor tolerance. The error comparator offset is trimmed so that under static conditions it trips when the feedback pin is 750mV, +1%. The on-time pulse from the SC173A in the design example is calculated to give a pseudo-fixed frequency of 800kHz. Some frequency variation with line and load is expected. This variation changes the output ripple voltage. Because adaptive on-time converters regulate to the valley of the output ripple, ½ of the output ripple appears as a DC regulation error. For example, if the output ripple is 50mV with VIN = 5 volts, then the measured DC output will be 25mV above the comparator trip point. If the ripple increases to 30mV with VIN = 5.5V, then the measured DC output will be 15mV above the comparator trip. The best way to minimize this effect is to minimize the output ripple. To compensate for valley regulation, it may be desirable to use passive droop. Take the feedback directly from the output side of the inductor and place a small amount of trace resistance between the inductor and output capacitor. This trace resistance should be optimized so that at full load the output droops to near the lower regulation limit. Passive droop minimizes the required output capacitance because the voltage excursions due to load steps are reduced as seen at the load. R1 COUT R2 Figure 7 — Signal Injection Circuit The values of RL, CL, CC and CFF are dependent on the conditions of the specific application such as VIN, VOUT, fSW and IOUT. For switching frequencies ranging from 600kHz to 800kHz, calculations plus experimental test results show that the following combination of RL=2.5kW, CL=10nF, CC=68pF and CFF=39pF can be used for many output voltages and loads. © 2010 Semtech Corporation 17 SC173A Applications Information (continued) The use of 1% feedback resistors may result in up to an additional 1% error. If tighter DC accuracy is required, resistors with lower tolerances should be used. The output inductor value may change with current. This will change the output ripple and therefore will have a minor effect on the DC output voltage. The output ESR also affects the output ripple and thus has a minor effect on the DC output voltage. deadtime diode conduction, as long as the transient voltage on PVIN is less than 6.0V. The time duration of the transient LX pin voltage is measured on the voltage portion which is either over 6.0V for positive voltage spike or under -1V for negative voltage spike. The LX voltage is measured from the LX pin to the PGND pin by using a probing loop which is as short as possible to minimize or eliminate the switching noise pick up. Switching Frequency Variation The switching frequency will vary depending on line and load conditions. The line variations are a result of fixed propagation delays in the on-time one-shot, as well as unavoidable delays in the power FET switching. As VIN increases, these factors make the actual DH on-time slightly longer than the ideal on-time. The net effect is that frequency tends to fall slightly with increasing input voltage. The switching frequency also varies with load current as a result of the power losses in the MOSFETs and the inductor. For a conventional PWM constant-frequency converter, as load increases the duty cycle also increases slightly to compensate for IR and switching losses in the MOSFETs and inductor. A adaptive on-time converter must also compensate for the same losses by increasing the effective duty cycle (more time is spent drawing energy from VIN as losses increase). The on-time is essentially constant for a given VOUT and VIN combination, to offset the losses the off-time will tend to reduce slightly as load increases. The net effect is that switching frequency increases slightly with increasing load. Switching Node Voltage Spike Due to parasitic board inductance, the transient LX pin voltage at the point of measurement may appear larger than that which exists on silicon. With an input multilayer ceramic capacitor of 10uF placed less than 3mm away from the PVIN pin, the device is designed and guaranteed to tolerate the short transient voltages, of maximum 20ns duration, that will appear on the LX pin due to the © 2010 Semtech Corporation 18 SC173A Layout Guideline VIN+ C11 VINC5 VO+ C1 1 2 3 L1 C4 C3 C2 R3 C7 R6 R7 SC173A U1 BST VIN LX VDD AGND TON EN PAD R1 C10 10 9 8 7 6 R2 Enable C6 0 4 PGND 0 5 PGOOD FB VO- 0 0 R4 C9 C8 0 Schematic for layout illustration Since the SC173A has integrated switches, special consideration should be given to board layout. Let us use the schematic shown above as an example. The board level layout is illustrated in the following four layers. As shown on the top layer layout, U1 is the switching regulator SC173A. C1 and C11 serve as the decoupling capacitor for the buck converter power train. C11, with a value between 1nF and 10nF, is the high frequency filtering capacitor. It is recommended to put C1 and C11 as close as possible to the SC173A to get the best decoupling performance, with C11 closest. C1, with a value of 10uF, should be placed no more than 3mm away from the VIN pin. L1 is the output filtering inductor. C2, C3 and C4 are the output filtering capacitors. C5 is the boostrap capacitor. Pin 10 (VDD) is the input bias power for the internal circuits. It is recommended to get the power from VIN through an RC filtering network consisted of R1, C6 and C10. The value of R1 can be between 3.01W and 10W and the capacitance of C10 should be above 1mF. C6, with a value of 1nF, is the high frequency filtering ca© 2010 Semtech Corporation pacitor. The locations of C6 and C10 should be as close as possible to pins 9 and 10, with C6 closest, to get the best possible filtering result. R2 is the on-time programming resistor. R2 should be located as close as possible to pin 8 and it should return to analog ground. Pull EN high (above 1V) or permit it to float to enable the part with automatic power save enabled. Connect EN to AGND to disable the switching regulator. Since there are two integrated MOSFETs inside the SC173A that will dissipate a lot of power, to help spread the heat out of the IC more efficiently, there is a thermal pad underneath the SC173A serving as a heat sink. To enlarge the heat sinking area, a large copper plane under the thermal pad as shown on the top layer is recommended. On inner layer 2, a large analog ground plane (AGND) on the right hand side is connected to the thermal pad underneath the SC173A using vias. Thus the heat generated inside the SC173A can be spread through the vias to the 19 SC173A inner layers to expand the heat sinking area. On the bottom layer, the resistor network composed of R3 and R4 determines the output voltage. C7 is the feed forward capacitor which helps to stabilize the circuit. R6 in series with C9 is connected to the LX pin (through the via) to the power ground. C8 is the coupling capacitor which injects the ramp signal generated on C9 to the FB pin of the SC173A. R7 is the pull up resistor for the PGOOD pin. © 2010 Semtech Corporation 20 SC173A VIN+ Top Layer C5 U1 C6 R1 C10 AGND C11 L1 C1 PGND C2 C3 C4 AGND R2 EN/PSV VO+ VO VIN VIN+ Inner Layer 1 LX 1 2 3 PGND 4 5 AGND 10 9 8 7 6 VO+ VO VIN © 2010 Semtech Corporation 21 SC173A VIN+ Inner Layer 2 LX 1 2 3 PGND AGND 10 9 8 7 6 4 5 VO+ VO VIN Bottom Layer 1 LX R7 PGND R6 AGND 10 9 8 7 6 2 3 4 5 C9 C8 R4 C7 R3 VO+ VO VIN © 2010 Semtech Corporation 22 SC173A Typical Application Circuits VIN+ R1 5.11Ohm VINC1 10uF/6.3V C101 0.1uF/6.3V 1uF/6.3V C4 L1 2.0uH R6 9.09k BST VIN LX PGND R4 100k PGOOD SC173A VDD AGND TON EN/PSV FB R3 54.9k 10uF/6.3V C5 0.1uF/6.3V C501 VOUT+ Enable FB VOUT- C6 22uF/6.3v 38p C10 R5 2 .5 k FB C18 68pF R2 15k C19 10n Application Circuit: Buck Converter with 1.2V out and 0 to 3A load current (Vin=5V) VIN+ R1 5.11Ohm VIN- C1 10uF/6.3V C101 0.1uF/6.3V 1uF/6.3V C4 L1 2.0uH 0.1uF/6.3V C501 10uF/6.3V C5 R3 80.6k VOUT+ SC173A VDD BST VIN LX PGND AGND TON EN/PSV FB C2 VOUT22uF/6.3v C6 22uF/6.3v 38p C10 Enable FB R6 51.1k R5 4.32k R4 100k PGOOD FB C18 68pF R2 15k C19 10n Application Circuit: Buck Converter with 3.3V out and 0 to 3A load current (Vin=5V) © 2010 Semtech Corporation 23 SC173A Outline Drawing - MLPD-10 3x3 A D B DIMENSIONS INCHES MILLIMETERS DIM MIN NOM MAX MIN NOM MAX A A1 A2 b D D1 E E1 e L N aaa bbb .031 .039 .000 .002 (.008) .008 .010 .012 .114 .118 .122 .087 .089 .091 .114 .118 .122 .057 .059 .061 .020 BSC .018 .020 .022 10 .003 .004 0.80 1.00 0.00 0.05 (0.20) 0.20 0.25 0.30 2.90 3.00 3.10 2.20 2.25 2.30 2.90 3.00 3.10 1.45 1.50 1.55 0.50 BSC 0.45 0.50 0.55 10 0.08 0.10 E PIN 1 INDICATOR (LASER MARK) A aaa C SEATING PLANE C A1 D1 1 2 A2 LxN E/2 E1 N e bxN bbb CAB D/2 NOTES: 1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES). 2. COPLANARITY APPLIES TO THE EXPOSED PAD AS WELL AS TERMINALS. © 2010 Semtech Corporation 24 SC173A Land Pattern - MLPD-10 3x3 K DIM K (C) H (C) H G G Y P DIM Z Z Y X X P C G H K P X Y Z (.114) (2.90) C DIMENSIONS G INCHES .083MILLIMETERS 2.10 1.40 H (.112) .055 (2.85) 2.20 K .079 .087 2.00 1.50 0.50 P .059 .020 2.25 0.30 X .089 .012 .020 0.50 .031 0.80 Y 0.30 .012 3.70 Z .033 .146 0.85 .146 3.70 DIMENSIONS INCHES MILLIMETERS NOTES: NOTES: 1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES). 2. 1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES). 2. THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY. CONSULT YOURMANUFACTURING GROUP TO ENSURE YOUR YOUR MANUFACTURING GROUP TO ENSURE CONSULT YOUR COMPANY'S MANUFACTURING GUIDELINES MET. MET. COMPANY'S MANUFACTURING GUIDELINES ARE ARE THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY. 3. 3.THERMAL VIAS IN THE LAND PATTERN OF THE EXPOSED PAD PAD THERMAL VIAS IN THE LAND PATTERN OF THE EXPOSED SHALL BE CONNECTED TO A A SYSTEM GROUND PLANE. SHALL BE CONNECTED TOSYSTEM GROUND PLANE. FAILURE TO DO SO MAY COMPROMISE THE THERMAL AND/OR FAILURE TO DO SO MAY COMPROMISE THE THERMAL AND/OR FUNCTIONAL PERFORMANCE OF THE DEVICE. FUNCTIONAL PERFORMANCE OF THE DEVICE. © 2010 Semtech Corporation 25 SC173A © Semtech 2010 All rights reserved. Reproduction in whole or in part is prohibited without the prior written consent of the copyright owner. The information presented in this document does not form part of any quotation or contract, is believed to be accurate and reliable and may be changed without notice. No liability will be accepted by the publisher for any consequence of its use. Publication thereof does not convey nor imply any license under patent or other industrial or intellectual property rights. Semtech assumes no responsibility or liability whatsoever for any failure or unexpected operation resulting from misuse, neglect improper installation, repair or improper handling or unusual physical or electrical stress including, but not limited to, exposure to parameters beyond the specified maximum ratings or operation outside the specified range. SEMTECH PRODUCTS ARE NOT DESIGNED, INTENDED, AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS, DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS. INCLUSION OF SEMTECH PRODUCTS IN SUCH APPLICATIONS IS UNDERSTOOD TO BE UNDERTAKEN SOLELY AT THE CUSTOMER’S OWN RISK. Should a customer purchase or use Semtech products for any such unauthorized application, the customer shall indemnify and hold Semtech and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs damages and attorney fees which could arise. Contact Information Semtech Corporation Power Mangement Products Division 200 Flynn Road, Camarillo, CA 93012 Phone: (805) 498-2111 Fax: (805) 498-3804 www.semtech.com © 2010 Semtech Corporation 26