SC174MLTRT

SC174 POWER MANAGEMENT Features               4A EcoSpeedTM Synchronous Step-Down Regulator with Optional Ultrasonic Power Save Description The SC174 is an integrated, synchronous 4A EcoSpeedTM step-down regulator, which incorporates Semtech’s advanced, patented adaptive on-time architecture to achieve best-in-class performance in dynamic point-ofload applications. The input voltage range is 3V to 5.5V with a programmable output voltage from 0.75V up to 95% x VIN. The device features low-RDS(ON) internal switches and optional PSAVE mode for high efficiency across the output load range. Adaptive on-time control provides programmable pseudo-fixed frequency operation and excellent transient performance. The switching frequency can be set from 200kHz to 1MHz - allowing the designer to reduce external LC filtering and minimize light load (standby) losses. Disabling PSAVE operation reduces output voltage ripple at light load for ceramic output capacitors. Additional features include cycle-by-cycle current limit, soft start, input UVLO and output OV protection, and over temperature protection. The open-drain PGOOD pin provides output status. Standby current is less than 10μA when disabled. The device is available in a low profile, thermally enhanced MLPD-3x3mm 10-pin package. VIN: 3V to 5.5V VOUT: 0.75V to 95% x VIN IOUT: Up to 4A Low RDS(ON) Switches Up to 95% Peak Efficiency High Output Accuracy Small Ceramic Capacitors Power Good Pin (Open-Drain) Patented Adaptive On-Time Control: Excellent Transient Response Programmable Pseudo-fixed Frequency Fault Protection Features: Cycle-by-Cycle Current Limit Short Circuit Protection Over and Under Output Voltage Protection Over-Temperature Internal Soft start Ultrasonic Power Save Smart Power Save Ultra-Small Lead-Free 3x3mm, 10-Pin MLPD Package Fully WEEE and RoHS Compliant Applications      Networking Equipment, Embedded Systems Medical Equipment, Office Automation Instrumentation, Portable Systems Consumer Devices (DTV, Set-top Box, ... ) 5V POL Converters Typical Application Circuit 3 to 5.5V VIN BST LX VOUT = 0.75V to 95% VIN SC174 VDD FB PGOOD Power Good Enable / Power Save EN/PSV PGND TON AGND May 18 , 2010 © 2010 Semtech Corporation 1 SC174 Pin Configuration Ordering Information Device SC174MLTRT(1) Top Mark 174 Package(2) MLPD-10 3x3 Evaluation Board BST VIN LX PGND PGOOD VDD AGND TON EN/PSV FB SC174EVB Notes: 1) Available in tape and reel packaging only. A reel contains 3000 devices. 2) Available in lead-free packaging only. WEEE compliant and Halogen free. This component and all homogenous sub-components are RoHS compliant. 10 Pin MLPD θJA= 40°C/W. Marking Information TOP MARKING 174 yyww xxxx yyww = Date Code (Example: 0952) xxxx = Semtech Lot Number (Example: 3901) © 2010 Semtech Corporation 2 SC174 Absolute Maximum Ratings LX to GND ……………………………………… - 0.3 to +6.0V VIN to PGND, EN/PSV to AGND ………………… -0.3 to +6.0V VIN to VDD…………………………………………………+0.3V BST to LX ………………………………………… -0.3 to +6.0V BST to PGND…………………………………… -0.3 to +12V VDD to AGND, VOUT to AGND………………… -0.3V to +6.0V FB, PGOOD, TON …………………………… -0.3 to VDD + 0.3V AGND to PGND………………………………… -0.3 to +0.3V Peak IR Reflow Temperature ……………….…………… 260°C ESD Protection Level(1) ………………………………………1kV Recommended Operating Conditions Supply Input Voltage……………………………… 3V to 5.5V Maximum Continuous Output Current …………………… 4A Thermal Information Storage Temperature …………………………… -60 to +150°C Maximum Junction Temperature ……………………… 150°C Operating Junction Temperature ……………… -40 to +125°C Thermal Resistance, Junction to Ambient(2) ………… 40°C/W Exceeding the above specifications may result in permanent damage to the device or device malfunction. Operation outside of the parameters specified in the Electrical Characteristics section is not recommended. NOTES(1) This device is ESD sensitive. Use of standard ESD handling precautions is required. (2) Tested according to JEDEC standard JESD22-A114-B. Electrical Characteristics Parameter Input Supplies VIN, VDD Input Voltage VDD UVLO Threshold VDD UVLO Hysteresis Unless specified: VIN =5V, TA=+25°C for Typ, -40°C to +85°C for Min and Max, TJ < 125°C Symbol Conditions Min Typ Max Units 3 Rising UVLO V TH 2.75 100 EN/PSV= 0V 2.90 200 5 500 1000 5.5 2.98 V V mV 15 μA VIN, VDD Supply Current IOUT=0A, PSAVE, fSW=25kHz(1) Forced Continuous Conduction Mode IOUT=0A , fSW=500kHz(1) Controller FB On-Time Threshold Frequency Programming Range FB Input Bias Current See RTON Calculation FB=VDD or 0V 0.7425 200 -1 0.75 0.7575 1000 +1 V kHz μA Timing On-Time Minimum On-Time(1) Minimum Off-Time(1) © 2010 Semtech Corporation Continuous Mode VIN=5V, VOUT=3V, RTON=200kΩ 2.7 3 80 250 3.3 μs ns ns 3 SC174 Electrical Characteristics (continued) Unless specified: VIN =5V, TA=+25°C for Typ, -40°C to +85°C for Min and Max, TJ < 125°C Parameter Soft start Soft start Time(1) Symbol Conditions Min Typ Max Units Delay from PWM Switching to Output Regulation 850 μs Ultrasonic Power Save Zero-Crossing Detector Threshold Ultrasonic Power Save Frequency LX - PGND 0 25 mV kHz Power Good Power Good Signal Threshold High Power Good Threshold Power Good Signal Threshold Low VDD=3V PGOOD Delay Time(1) VDD=5V Noise Immunity Delay Time Leakage Power Good On-Resistance 10 2 5 1 20 µs µA Ω 86 90 1 ms 93 116 120 124 %VOUT Fault Protection Output Under-Voltage Fault Output Over-Voltage Fault Smart PowerSave Protection Threshold OV, UV Fault Noise Immunity Delay Over-Temperature Shutdown OT Latched FB with Respect to REF, 8 Consecutive Clocks FB with Respect to REF FB with Respect to REF -30 +16 +7 -25 +20 +10 5 150 -20 +24 +13 % % % μs °C Enable/Power Save Output Enabled Output Disabled EN/PSV Input Bias Current Power Save Enabled Forced Continuous Conduction Mode EN/PSV floating EN/PSV = VDD or 0V 60 39 41 44 0.5 1 0.4 8.0 V V μA %VDD %VDD © 2010 Semtech Corporation 4 SC174 Electrical Characteristics (continued) Unless specified: VIN =5V, TA=+25°C for Typ, -40°C to +85°C for Min and Max, TJ < 125°C Parameter Gate Drivers BST Switch On resistance Symbol Conditions Min Typ Max Units 25 45 Ω Internal Power MOSFETs Valley Current Limit, VDD=5V Current Limit Valley Current Limit, VDD=3V LX Leakage Current Switch Resistance Low Side Non-overlap time (1) Note: (1) Typical value from EVB, not ATE tested. 40 30 65 ns VIN=5.5V, LX=0V, High Side High Side 3.5 4 1 50 10 75 mΩ µA 4.5 A © 2010 Semtech Corporation 5 SC174 Pin Descriptions (MLPD-10) Pin # 1 2 3 4 5 6 Pin Name BST VIN LX PGND PGOOD FB Pin Function Bootstrap pin. A capacitor is connected between BST to LX to develop the floating voltage for the high-side gate drive. Power input supply voltage. Switching (Phase) node. Power ground. Open-drain Power Good indicator. High impedance indicates power is good. An external pull-up resistor is required. Feedback input for switching regulator. Connect to an external resistor divider from the output to program the output voltage. Tri-state pin. Enable input for switching regulator. Pull EN/PSV high to enable the part with power save mode enabled. Connect EN/PSV to AGND to disable the switching regulator. Leave EN/PSV floating to enable the IC in forced continuous conduction mode. On-time set input. Set the on-time by a series resistor to AGND. Analog Ground. Input power for internal control circuit. Needs 1mF decoupling capacitor from this pin to AGND. Thermal pad for heatsinking purposes. Connect to ground plane using multiple vias. Not connected internally. 7 EN/PSV 8 9 10 TON AGND VDD PAD © 2010 Semtech Corporation 6 SC174 Block Diagram 10 VDD 9 AGND Reference 7 5 PGOOD Control 1 EN VDD BST VIN 2 Soft Start FB On-Time Generator Gate Drive Control VDD LX 3 6 8 TON PGND Zero-Cross Valley Current Limit R 4 © 2010 Semtech Corporation 7 SC174 Typical Characteristics Efficiency vs Output Current 100.0 95.0 3.340 Output Voltage vs Output Current Vin=5V, Vo=3.3V, LOUT: DS86C-B992AS-2R0N, COUT=22 Fx2 RTON = 80.6kOhm Red: PSAVE Mode Blue: Forced Continuous PWM Mode Efficiency (%) 90.0 85.0 80.0 75.0 70.0 65.0 0 1 Vin=5V, Vo=3.3V, LOUT: DS86C-B992AS-2R0N, COUT=22 Fx2 RTON = 80.6kOhm Red: PSAVE Mode Blue: Forced Continuous PWM Mode Output Voltage (V) 4 3.331 3.323 3.314 3.306 3.297 Output Current (A) 2 3 0 1 Output Current (A) 2 3 4 Efficiency vs Output Current 95.0 90.0 1.215 Output Voltage vs Output Current Vin=5V, Vo=1.2V, LOUT: DS86C-B992AS-2R0N COUT=22 F RTON = 54.9kOhm Red: PSAVE Mode Blue: Forced Continuous PWM Mode Efficiency (%) 85.0 80.0 75.0 70.0 65.0 60.0 55.0 0 1 Vin=5V, Vo=1.2V, LOUT: DS86C-B992AS-2R0N COUT=22 F RTON = 54.9kOhm Red: PSAVE Mode Blue: Forced Continuous PWM Mode Output Voltage (V) 3 4 1.210 1.205 1.200 Output Current (A) 2 1.195 0 1 Output Current (A) 2 3 4 Efficiency vs Output Current 95.0 90.0 1.210 Output Voltage vs Output Current Vo=1.205V, RTON = 54.9kOhm LOUT: DS86C-B992AS-2R0N COUT:22 F Forced Continuous Mode Black: Vin = 5V Red: Vin = 4V Blue: Vin = 3.5V Output Voltage (V) 3 4 85.0 Efficiency (%) 1.208 80.0 75.0 70.0 65.0 60.0 55.0 50.0 0 Vo=1.2V, RTON = 54.9kOhm LOUT: DS86C-B992AS-2R0N COUT:22 F Forced Continuous Mode Red:Vin = 3.5V Green: Vin = 4.0V Blue: Vin = 5.0V 1.205 1.203 1 © 2010 Semtech Corporation Output Current (A) 2 1.200 0 1 Output Current (A) 2 3 4 8 SC174 Typical Characteristics OCP Valley Threshold vs Temperature 5.5 5.3 OCP Valley Threshold (A) Start up waveform ( VIN=5V, VOUT=1.2V, IOUT=4A, Channel 1: 500mV/Div, Channel 4: 2A/Div, Time: 0.5ms/Div ) V DD =5.0V 5.0 4.8 4.5 4.3 4.0 3.8 3.5 -50 -25 0 25 50 75 100 125 Temperature (°C) V DD =3.0V Load Transient Test (VIN=5V, VOUT=1.2V, IOUT= 0A to 4A, Forced Continuous Conduction Mode, LOUT=1.3 H, COUT=3x22 F, Channel 1: 50mV/Div, Channel 2:5V/Div,Channel 4: 2A/Div, Time:10 s/Div) IVIN Input Current In Shutdown vs Temperature 3.5 IVIN Input Current In Shutdown ( A) 3.0 2.5 2.0 1.5 1.0 0.5 0.0 -50 -25 0 25 50 75 100 125 Temperature (°C) VIN =5V Blue: VLX=GND Black: VLX=VIN Load Transient Test (VIN=5V, VOUT=1.2V, IOUT= 4A to 0A, Forced Continuous Conduction Mode, LOUT=1.3 H, COUT=3x22 F, Channel 1: 50mV/Div, Channel 2:5V/Div,Channel 4: 2A/Div, Time:10 s/Div) IBST Leakage Current vs Temperature 0.5 0.0 -0.5 -1.0 -1.5 VIN =5V IBST Leakage Current ( A) -2.0 -2.5 -3.0 -50 -25 0 VBST=VIN 25 50 75 100 125 Temperature (°C) © 2010 Semtech Corporation 9 SC174 Typical Characteristics FB Voltage vs Temperature 0.755 Load Transient Test (VIN=5V, VOUT=1.2V, IOUT= 0A to 4A, PSAVE Mode Enabled, LOUT=1.3 H,COUT=3x22 F,Channel 1: 50mV/Div, Channel 2:5V/Div,Channel 4:2A/Div,Time:20 s/Div) 0.753 FB Voltage (V) 0.750 Black:VDD=5.0V Red: VDD=3.0V 0.748 0.745 -40 -15 10 35 60 85 110 135 Temperature (°C) 58 54 On-State Resistance (m ) Low Side Switch On-State Resistance vs T emperature Load Transient Test (VIN=5V, VOUT=1.2V, IOUT= 4A to 0A, PSAVE Mode Enabled, LOUT=1.3 H, COUT=3x22 F, Channel 1: 50mV/Div,Channel 2:5V/Div,Channel 4:2A/Div, Time:20 s/Div) 50 46 42 38 34 30 -40 -15 10 35 60 85 110 135 Temperature (°C) Blue: VDD=3.0V Black: VDD=5.0V 64 60 On-State Resistance (m ) High Side Switch On-State Resistance vs T emperature 56 52 48 44 40 -40 -15 10 35 60 85 110 135 Temperature (°C) Blue: VDD=3.0V Black: VDD=5.0V © 2010 Semtech Corporation 10 SC174 Applications Information SC174 Synchronous Buck Converter The SC174 is a step down synchronous buck dc-dc regulator. The SC174 is capable of 4A operation at very high efficiency in a tiny 3x3-10 pin package. The programmable operating frequency range of 200KHz – 1MHz enables the user to optimize the solution for minimum board space and optimum efficiency. The buck regulator employs pseudo-fixed frequency adaptive on-time control. This control scheme allows fast transient response thereby lowering the size of the power components used in the system. Input Voltage Range The SC174 can operate with an input voltage ranging from 3V to 5.5V. Psuedo-fixed Frequency Adaptive On-time Control The PWM control method used by the SC174 is pseudofixed frequency, adaptive on-time, as shown in Figure 1. The ripple voltage generated at the output capacitor ESR is used as a PWM ramp signal. This ripple is used to trigger the on-time of the controller. VIN CIN Q1 L VLX Q2 ESR TON VFB VLX FB threshold VOUT The adaptive on-time is determined by an internal oneshot timer. When the one-shot is triggered by the output ripple, the device sends a single on-time pulse to the high-side MOSFET. The pulse period is determined by VOUT and VIN; the period is proportional to output voltage and inversely proportional to input voltage. With this adaptive on-time arrangement, the device automatically anticipates the on-time needed to regulate VOUT for the present VIN condition and at the selected frequency. The advantages of adaptive on-time control are: • • • • • Predictable operating frequency compared to other variable frequency methods. Reduced component count by eliminating the error amplifier and compensation components. Reduced component count by removing the need to sense and control inductor current. Fast transient response — the response time is controlled by a fast comparator instead of a typically slow error amplifier. Reduced output capacitance due to fast transient response One-Shot Timer and Operating Frequency The one-shot timer operates as shown in Figure 2. The FB Comparator output goes high when VFB is less than the internal 750mV reference. This feeds into the gate drive and turns on the high-side MOSFET, and also starts the one-shot timer. The one-shot timer uses an internal comparator, timing capacitor, and a low pass filter (LPF) which regenerates VOUT from LX. One comparator input is connected to the filtered LX voltage, the other input is connected to the capacitor. When the on-time begins, the internal capacitor charges from zero volts through a current which is proportional to VIN. When the capacitor voltage reaches VOUT, the on-time is completed and the high-side MOSFET turns off. This method automatically produces an on-time that is proportional to VOUT and inversely proportional to VIN. Under steady-state operation conditions, the switching frequency can be determined from the on-time by the following equation. COUT + FB Figure 1 — PWM Control Method, VOUT Ripple © 2010 Semtech Corporation fSW = VOUT TON × VIN 11 SC174 Applications Information (continued) FB REF + Enable and Power-save Inputs The EN/PSV inputs are used to enable or disable the switching regulator. When EN/PSV is low (grounded), the switching regulator is off and in its lowest power state. When off, the output power switches are tri-stated. When EN/PSV is allowed to float, the pin voltage will float to 41% of the voltage at VDD. The switching regulator turns on with power-save disabled and all switching is in forced continuous mode. When EN/PSV is high (above 60% of the voltage at VDD), the switching regulator turns on with ultrasonic powersave enabled. The SC174 ultrasonic power-save operation maintains a minimum switching frequency of 25kHz, for applications with stringent audio requirements. Forced Continuous Mode Operation The SC174 operates in Forced Continuous Mode (FCM) by floating the EN/PSV pin (see Figure 4). In this mode one of the power MOSFETs is always on, with no intentional dead time other than to avoid cross-conduction. This feature results in uniform frequency across the full load range with the trade-off being poor efficiency at light loads due to the high-frequency switching of the MOSFETs. PWM S R Q Hi-Side and Lo-Side Gate Drivers VIN Q1 VLX Q2 LPF L ESR COUT + VOUT VOUT VIN On-Shot Timing Generator Time = K x VOUT/VIN FB RTON Figure 2 — On-Time Generation The SC174 uses an external resistor to set the on-time which indirectly sets the frequency. The on-time can be programmed to provide operating frequency from 200kHz to 1MHz using a resistor between the TON pin and ground. The resistor value is selected by the following equation. R TON 1 = 25pF ⋅ fSW VOUT Voltage Selection The switcher output voltage is regulated by comparing VOUT as seen through a resistor divider at the FB pin to the internal 750mV reference voltage, see Figure 3. VOUT R1 R2 To FB pin Figure 3 — Output Voltage Selection Note that this control method regulates the valley of the output ripple voltage, not the DC value. The DC output voltage VOUT is offset by the output ripple according to the following equation.  R V VOUT = 0.75V ⋅  1 + 1  + RIPPLE  R2  2   Figure 4 — Forced Continuous Mode Operation © 2010 Semtech Corporation 12 SC174 Applications Information (continued) Ultrasonic Power Save Operation The SC174 provides ultrasonic power save operation at light loads, with the minimum operating frequency fixed at 25kHz. This is accomplished using an internal timer that monitors the time between consecutive high-side gate pulses. If the time exceeds 40µs, DL drives high to turn the low-side MOSFET on. This draws current from VOUT through the inductor, forcing both VOUT and VFB to fall. When VFB drops to the 750mV threshold, the next DH on-time is triggered. After the on-time is completed the high-side MOSFET is turned off and the low-side MOSFET turns on. The low-side MOSFET remains on until the inductor current ramps down to zero, at which point the low-side MOSFET is turned off. Because the on-times are forced to occur at intervals no greater than 40µs, the frequency will not fall below ~25kHz. Figure 5 shows ultrasonic power save operation. power save enabled, this can force VOUT to slowly rise and reach the over-voltage threshold, resulting in a hard shutdown. Smart power save prevents this condition. When the FB voltage exceeds 10% above nominal (exceeds 825mV), the device immediately disables powersave, and DL drives high to turn on the low-side MOSFET. This draws current from VOUT through the inductor and causes VOUT to fall. When VFB drops back to the 750mV trip point, a normal TON switching cycle begins. This method prevents a hard OVP shutdown and also cycles energy from VOUT back to VIN. Figure 6 shows typical waveforms for the Smart Power Save feature. Figure 6 — Smart Power Save Current Limit Protection The device features fixed current limiting, which is accomplished by using the RDS(ON) of the lower MOSFET for current sensing. While the low-side MOSFET is on, the inductor current flows through it and creates a voltage across the RDS(ON). During this time, the voltage across the MOSFET is negative with respect to ground. During this time, If this MOSFET voltage drop exceeds the internal reference voltage, the current limit will activate. The current limit then keeps the low-side MOSFET on and will not allow another high side on-time, until the current in the low-side MOSFET reduces enough to drop below the internal reference voltage once more. This method regulates the inductor valley current at the level shown by ILIM in Figure 7. Figure 5 — Ultrasonic Power Save Operation Smart Power Save Protection Active loads may leak current from a higher voltage into the switcher output. Under light load conditions with © 2010 Semtech Corporation 13 SC174 Applications Information (continued) Inductor Current IPEAK ILOAD ILIM During soft start the regulator turns off the low-side MOSFET on any cycle if the inductor current falls to zero. This prevents negative inductor current, allowing the device to start into a pre-biased output. This soft start operation is implemented even if FCM is selected. FCM operation is allowed only after PGOOD is high. Power Good Output Time Figure 7 — Valley Current Limit Setting the valley current limit to a value of ILIM results in a peak inductor current of ILIM plus the peak-to-peak ripple current. In this situation, the average (load) current through the inductor will be ILIM plus one half the peakto-peak ripple current. Soft start of PWM Regulator Soft start is achieved in the PWM regulator by using an internal voltage ramp as the reference for the FB comparator. The voltage ramp is generated using an internal charge pump which drives the reference from zero to 750mV in ~1.8mV increments, using an internal ~500kHz oscillator. When the ramp voltage reaches 750mV, the ramp is ignored and the FB comparator switches over to a fixed 750mV threshold. During soft start the output voltage tracks the internal ramp, which limits the start-up inrush current and provides a controlled soft start profile for a wide range of applications. Typical soft start ramp time is 0.85ms. The power good (PGOOD) output is an open-drain output which requires a pull-up resistor. When the output voltage is 10% below the nominal voltage, PGOOD is pulled low. It is held low until the output voltage returns to the nominal voltage. PGOOD is held low during soft start and activated approximately 1ms after VOUT reaches regulation. The total PGOOD delay is typically 2ms. PGOOD will transition low if the VFB pin exceeds +20% of nominal, which is also the over-voltage shutdown threshold (900mV). PGOOD also pulls low if the EN/PSV pin is low when VDD is present. Output Over-Voltage Protection Over-Voltage Protection (OVP) becomes active as soon as the device is enabled. The threshold is set at 750mV + 20% (900mV). When VFB exceeds the OVP threshold, DL latches high and the low-side MOSFET is turned on. DL remains high and the controller remains off, until the EN/ PSV input is toggled or VDD is cycled. There is a 5μs delay built into the OVP detector to prevent false transitions. PGOOD is also low after an OVP event. Output Under-Voltage Protection When VFB falls to 75% of its nominal voltage (falls to 562.5mV) for eight consecutive clock cycles, the switcher is shut off and the DH and DL drives are pulled low to turn off the MOSFETs. The controller stays off until EN/PSV is toggled or VDD is cycled. VDD UVLO, and POR Under-Voltage Lock-Out (UVLO) circuitry inhibits switching and tri-states the power FETs until VDD rises above 2.9V. An internal Power-On Reset (POR) occurs when VDD exceeds 2.9V, which resets the fault latch and soft start counter to begin the soft start cycle. The SC174 then begins a soft start cycle. The PWM will shut off if VDD falls © 2010 Semtech Corporation 14 SC174 Applications Information (continued) below 2.7V. Design Procedure When designing a switch mode supply the input voltage range, load current, switching frequency, and inductor ripple current must be specified. The maximum input voltage (VINMAX) is the highest specified input voltage. The minimum input voltage ( VINMIN) is determined by the lowest input voltage after evaluating the voltage drops due to connectors, fuses, switches, and PCB traces. The following parameters define the design. R TON = setting the frequency) using the following equation. 1 25pF ⋅ fSW Substituting RTON results in the following solution. RTON=50kW, we use RTON=49.9kW in real application. Inductor Selection In order to determine the inductance, the ripple current must first be defined. Low inductor values result in smaller size but create higher ripple current which can reduce efficiency. Higher inductor values will reduce the ripple current/voltage and for a given DC resistance are more efficient. However, larger inductance translates directly into larger packages and higher cost. Cost, size, output ripple, and efficiency are all used in the selection process. The ripple current will also set the boundary for powersave operation. The switching will typically enter powersave mode when the load current decreases to 1/2 of the ripple current. For example, if ripple current is 4A then Power-save operation will typically start for loads less than 2A. If ripple current is set at 40% of maximum load current, then power-save will start for loads less than 20% of maximum current. The inductor value is typically selected to provide a ripple current that is between 25% to 50% of the maximum load current. This provides an optimal trade-off between cost, efficiency, and transient performance. During the DH on-time, voltage across the inductor is (VIN - VOUT). The equation for determining inductance is shown next. TON = VOUT VINMAX ⋅ fSW • • • • Nominal output voltage (VOUT) Static or DC output tolerance Transient response Maximum load current (IOUT) There are two values of load current to evaluate — continuous load current and peak load current. Continuous load current relates to thermal stresses which drive the selection of the inductor and input capacitors. Peak load current determines instantaneous component stresses and filtering requirements such as inductor saturation, output capacitors, and design of the current limit circuit. The following values are used in this design. VIN = 5V + 10% VOUT = 1.0V + 4% fSW = 800kHz Load = 4A maximum • • • • Frequency Selection Selection of the switching frequency requires making a trade-off between the size and cost of the external filter components (inductor and output capacitor) and the power conversion efficiency. The desired switching frequency is 800kHz which results from using components selected for optimum size and cost . A resistor (RTON) is used to program the on-time (indirectly © 2010 Semtech Corporation L= (VIN - VOUT) × TON IRIPPLE 15 SC174 Applications Information (continued) Example In this example, the inductor ripple current is set equal to 50% of the maximum load current. Therefore ripple current will be 50% x 4A or 2A. To find the minimum inductance needed, use the VIN and TON values that correspond to VINMAX. TON_VINMAX = L= The maximum ripple current of 0.511A creates a ripple voltage across the ESR. The maximum ESR value allowed is shown by the following equations. ESR MAX = IRIPPLEMAX V RIPPLE = 40 mV 0.51A ESRMAX = 80 mΩ The output capacitance is chosen to meet transient requirements. A worst-case load release, from maximum load to no load at the exact moment when inductor current is at the peak, determines the required capacitance. If the load release is instantaneous (load changes from maximum to zero in < 1µs), the output capacitor must absorb all the inductor’s stored energy. This will cause a peak voltage on the capacitor according to the following equation. COUTMIN = L × (IOUT + 1 × IRIPPLEMAX )2 2 (VPEAK )2 - (VOUT )2 25pF × R TON × VOUT = 227ns VINMAX (5.5V - 1V) • 227ns = 0.51mH 2A A larger value of 2µH is selected. This will decrease the maximum IRIPPLE to 0.511A. Note that the inductor must be rated for the maximum DC load current plus 1/2 of the ripple current. The ripple current under minimum VIN conditions is also checked using the following equations. TON_VINMIN = 25pF × RTON × VOUT = 277ns VINMIN Assuming a peak voltage VPEAK of 1.050V (50mV rise upon load release), and a 4A load release, the required capacitance is shown by the next equation. 2mH × (4A + 1 × 0.511A) 2 2 = 353 mF (1.05V) 2 - (1.0V) 2 IRIPPLE = IRIPPLE_VINMIN = (VIN - VOUT ) × TON L COUTMIN = (4.5V - 1V) × 277ns = 0.485A 2mH Capacitor Selection The output capacitors are chosen based on required ESR and capacitance. The maximum ESR requirement is controlled by the output ripple requirement and the DC tolerance. The output voltage has a DC value that is equal to the valley of the output ripple plus 1/2 of the peakto-peak ripple. Change in the output ripple voltage will lead to a change in DC voltage at the output. The design goal is for the output voltage regulation to be ±4% under static conditions. The internal 750mV reference tolerance is 1%. Assuming a 1% tolerance from the FB resistor divider, this allows 2% tolerance due to VOUT ripple. Since this 2% error comes from 1/2 of the ripple voltage, the allowable ripple is 4%, or 40mV for a 1V output. © 2010 Semtech Corporation If the load release is relatively slow, the output capacitance can be reduced. At heavy loads during normal switching, when the FB pin is above the 750mV reference, the DL output is high and the low-side MOSFET is on. During this time, the voltage across the inductor is approximately -VOUT. This causes a down-slope or falling di/dt in the inductor. If the load di/dt is not much faster than the -di/dt in the inductor, then the inductor current will tend to track the falling load current. This will reduce the excess inductive energy that must be absorbed by the output capacitor, therefore a smaller capacitance can be used. The following can be used to calculate the needed capacitance for a given dILOAD/dt. Peak inductor current is shown by the next equation. 16 SC174 Applications Information (continued) 1 ILPK = 4A + × 0.511A = 4.26A 2 increase the ESR of the output capacitors. It is also imperative to provide a proper PCB layout as discussed in the Layout Guidelines section. Another way to eliminate doubling-pulsing is to add a small (~ 10pF) capacitor across the upper feedback resistor, as shown in Figure 8. This capacitor should be left unpopulated unless it can be confirmed that doublepulsing exists. Adding the CTOP capacitor will couple more ripple into FB to help eliminate the problem. An optional connection on the PCB should be available for this capacitor. CTOP Rate of change of load current is dI LOAD 0.6A = dt 1m s IMAX = maximum load release = 4A COUT I I L × LPK - MAX × dt VOUT dILOAD = ILPK × 2 × ( VPK - VOUT ) 2mH × 4.26A 4A × 1ms 1V 0.6A 2 × (1.05V - 1V) VOUT R1 R2 To FB pin COUT = 4.26A × C OUT = 79mF Figure 8 — Capacitor Coupling to FB Pin ESR loop instability is caused by insufficient ESR. The details of this stability issue are discussed in the ESR Requirements section. The best method for checking stability is to apply a zero-to-full load transient and observe the output voltage ripple envelope for overshoot and ringing. Ringing for more than one cycle after the initial step is an indication that the ESR should be increased. One simple way to solve this problem is to add trace resistance in the high current output path. A side effect of adding trace resistance is a decrease in load regulation. Note that COUT is much smaller in this example, 79µF compared to 353µF based upon a worst-case load release. To meet the two design criteria of minimum 79µF and maximum 15mΩ ESR, select two capacitors rated at 47µF and 15mΩ ESR or less. It is recommended that an additional small capacitor be placed in parallel with COUT in order to filter high frequency switching noise. Stability Considerations Unstable operation is possible with adaptive on-time controllers, and usually takes the form of double-pulsing or ESR loop instability. Double-pulsing occurs due to switching noise seen at the FB input or because the FB ripple voltage is too low. This causes the FB comparator to trigger prematurely after the minimum off-time has expired. In extreme cases the noise can cause three or more successive on-times. Double-pulsing will result in higher ripple voltage at the output, but in most applications it will not affect operation. This form of instability can usually be avoided by providing the FB pin with a smooth, clean ripple signal that is at least 10mVp-p, which may dictate the need to © 2010 Semtech Corporation ESR Requirements A minimum ESR is required for two reasons. One reason is to generate enough output ripple voltage to provide 10mVp-p at the FB pin (after the resistor divider) to avoid double-pulsing. The second reason is to prevent instability due to insufficient ESR. The on-time control regulates the valley of the output ripple voltage. This ripple voltage is the sum of the two voltages. One is the ripple generated by the ESR, the other is the ripple due to capacitive charging 17 SC174 Applications Information (continued) and discharging during the switching cycle. For most applications, the total output ripple voltage is dominated by the output capacitors, typically SP or POSCAP devices. For stability the ESR zero of the output capacitor should be lower than approximately one-third the switching frequency. The formula for minimum ESR is shown by the following equation. ESR MIN = 3 2 × π × COUT × fSW Output Voltage Dropout The output voltage adjustable range for continuousconduction operation is limited by the fixed 320ns (typical) minimum off-time. When working with low input voltages, the duty-factor limit must be calculated using worst-case values for on and off times. The duty-factor limitation is shown by the next equation. DUTY TON(MIN) TON(MIN) TOFF(MAX ) Using Ceramic Output Capacitors When applications use ceramic output capacitors, the ESR is normally too small to meet the previously stated ESR criteria. In these applications it is necessary to add a small signal injection network as shown in Figure 9. In this network RL and CL filter the LX switching waveform to generate an in-phase ripple voltage comparable to the ripple seen on higher ESR capacitors. CC is a coupling capacitor used to AC couple the generated ripple onto the FB pin. Capacitor CFF is required for min COUT applications. This capacitor introduces a lead/lag into the control with the maximum phase placed at 1/2 fSW for added stability. VIN Q1 VLX RL Q2 CL CC L CFF The inductor resistance and MOSFET on-state voltage drops must be included when performing worst-case dropout duty-factor calculations. System DC Accuracy — VOUT Controller Three factors affect VOUT accuracy: the trip point of the FB error comparator, the ripple voltage variation with line and load, and the external resistor tolerance. The error comparator offset is trimmed so that under static conditions it trips when the feedback pin is 750mV, +1%. The on-time pulse from the SC174 in the design example is calculated to give a pseudo-fixed frequency of 800kHz. Some frequency variation with line and load is expected. This variation changes the output ripple voltage. Because adaptive on-time converters regulate to the valley of the output ripple, ½ of the output ripple appears as a DC regulation error. For example, if the output ripple is 50mV with VIN = 5 volts, then the measured DC output will be 25mV above the comparator trip point. If the ripple increases to 30mV with VIN = 5.5V, then the measured DC output will be 15mV above the comparator trip. The best way to minimize this effect is to minimize the output ripple. To compensate for valley regulation, it may be desirable to use passive droop. Take the feedback directly from the output side of the inductor and place a small amount of trace resistance between the inductor and output capacitor. This trace resistance should be optimized so that at full load the output droops to near the lower regulation limit. Passive droop minimizes the required output capacitance because the voltage excursions due to load steps are reduced as seen at the load. 18 R1 COUT R2 Figure 9 — Signal Injection Circuit The values of RL, CL, CC and CFF are dependent on the conditions of the specific application such as VIN, VOUT, fSW and IOUT. For switching frequencies ranging from 600kHz to 800kHz, calculations plus experimental test results show that the following combination of RL=2.5kW, CL=10nF, CC=68pF and CFF=39pF can be used for many output voltages and loads. © 2010 Semtech Corporation SC174 Applications Information (continued) The use of 1% feedback resistors may result in up to an additional 1% error. If tighter DC accuracy is required, resistors with lower tolerances should be used. The output inductor value may change with current. This will change the output ripple and therefore will have a minor effect on the DC output voltage. The output ESR also affects the output ripple and thus has a minor effect on the DC output voltage. Switching Frequency Variation The switching frequency will vary depending on line and load conditions. The line variations are a result of fixed propagation delays in the on-time one-shot, as well as unavoidable delays in the power FET switching. As VIN increases, these factors make the actual DH on-time slightly longer than the ideal on-time. The net effect is that frequency tends to fall slightly with increasing input voltage. The switching frequency also varies with load current as a result of the power losses in the MOSFETs and the inductor. For a conventional PWM constant-frequency converter, as load increases the duty cycle also increases slightly to compensate for IR and switching losses in the MOSFETs and inductor. A adaptive on-time converter must also compensate for the same losses by increasing the effective duty cycle (more time is spent drawing energy from VIN as losses increase). The on-time is essentially constant for a given VOUT and VIN combination, to offset the losses the off-time will tend to reduce slightly as load increases. The net effect is that switching frequency increases slightly with increasing load. © 2010 Semtech Corporation 19 SC174 Layout Guideline VIN+ C11 VINC5 VO+ C1 11 22 33 L1 C4 C3 C2 R3 C7 R6 R7 44 SC174 U1 BST VIN LX PGND PGOOD VDD AGND TON EN/PSV PAD R1 C10 10 10 99 8 77 66 R2 EN/PSV C6 0 0 55 FB VO- 0 0 R4 C9 C8 0 Schematic for layout illustration Since the SC174 has integrated switches, special consideration should be given to board layout. Let us use the schematic shown above as an example. The board level layout is illustrated in the following four layers. As shown on the top layer layout, U1 is the switching regulator SC174. C1 and C11 serve as the decoupling capacitor for the buck converter power train. C11, with a value between 1nF and 10nF, is the high frequency filtering capacitor. It is recommended to put C1 and C11 as close as possible to the SC174 to get the best decoupling performance, with C11 closest. L1 is the output filtering inductor. C2, C3 and C4 are the output filtering capacitors. C5 is the boostrap capacitor. Pin 10 (VDD) is the input bias power for the internal circuits. It is recommended to get the power from VIN through an RC filtering network consisted of R1, C6 and C10. The value of R1 can be between 3.01W and 10W and the capacitance of C10 should be above 1mF. C6, with a value of 1nF, is the high frequency filtering capacitor. The locations of C6 and C10 should be as close as possible to pins 9 and 10, © 2010 Semtech Corporation with C6 closest, to get the best possible filtering result. R2 is the on-time programming resistor. R2 should be located as close as possible to pin 8 and it should return to analog ground. The EN/PSV pin is a tri-state pin. Pull EN/PSV high to enable the part with power save mode enabled. Connect EN/PSV to AGND to disable the switching regulator. Leave EN/PSV floating to enable the IC in forced continuous conduction mode. Since there are two integrated MOSFETs inside the SC174 that will dissipate a lot of power, to help spread the heat out of the IC more efficiently, there is a thermal pad underneath the SC174 serving as a heat sink. To enlarge the heat sinking area, a large copper plane under the thermal pad as shown on the top layer is recommended. On inner layer 2, a large analog ground plane (AGND) on the right hand side is connected to the thermal pad underneath the SC174 using vias. Thus the heat generated inside the SC174 can be spread through the vias to the inner layers to expand the heat sinking area. 20 SC174 On the bottom layer, the resistor network composed of R3 and R4 determines the output voltage. C7 is the feed forward capacitor which helps to stabilize the circuit. R6 in series with C9 is connected to the LX pin (through the via) to the power ground. C8 is the coupling capacitor which injects the ramp signal generated on C9 to the FB pin of the SC174. R7 is the pull up resistor for the PGOOD pin. © 2010 Semtech Corporation 21 SC174 VIN+ Top Layer C5 U1 C6 C10 R1 AGND C11 L1 C1 PGND C2 C3 C4 AGND R2 EN/PSV VO+ VO VIN VIN+ Inner Layer 1 LX 1 2 3 PGND 4 5 AGND 10 9 8 7 6 VO+ VO VIN © 2010 Semtech Corporation 22 SC174 VIN+ Inner Layer 2 LX 1 2 3 PGND AGND 10 9 8 7 6 4 5 VO+ VO VIN Bottom Layer 1 LX R7 PGND R6 AGND 10 9 8 7 6 2 3 4 5 C9 C8 R4 C7 R3 VO+ VO VIN © 2010 Semtech Corporation 23 SC174 Typical Application Circuits VIN+ R1 5.11Ohm VINC1 10uF/6.3V 1uF/6.3V C4 L1 2.0uH R6 9.09k BST VIN LX SC174 VDD AGND TON EN/PSV FB R3 54.9k 10uF/6.3V C5 0 VOUT+ 0 0 R5 2.5k R4 100k PGND PGOOD Enable FB VOUT- C6 22uF/6.3v 38p C10 0 FB C18 68pF R2 15k C19 10n Application Circuit: Buck Converter with 1.2V out and 0 to 4A load current (Vin=3V to 5.5V) VIN+ R1 5.11Ohm VINC1 10uF/6.3V 1uF/6.3V C4 L1 2.0uH R6 51.1k SC174 BST VIN LX VDD 10uF/6.3V C5 R3 80.6k 0 VOUT+ 0 AGND TON EN/PSV FB 0 R5 3.48k R4 100k PGND PGOOD Enable FB VOUT- C2 C6 22uF/6.3v 22uF/6.3v 38p C10 0 FB C18 68pF R2 15k C19 10n Application Circuit: Buck Converter with 3.3V out and 0 to 4A load current (Vin=5V) © 2010 Semtech Corporation 24 SC174 Outline Drawing - MLPD-10 3x3 A D B DIMENSIONS INCHES MILLIMETERS DIM MIN NOM MAX MIN NOM MAX A A1 A2 b D D1 E E1 e L N aaa bbb .031 .039 .000 .002 (.008) .008 .010 .012 .114 .118 .122 .087 .089 .091 .114 .118 .122 .057 .059 .061 .020 BSC .018 .020 .022 10 .003 .004 0.80 1.00 0.00 0.05 (0.20) 0.20 0.25 0.30 2.90 3.00 3.10 2.20 2.25 2.30 2.90 3.00 3.10 1.45 1.50 1.55 0.50 BSC 0.45 0.50 0.55 10 0.08 0.10 E PIN 1 INDICATOR (LASER MARK) A aaa C SEATING PLANE C A1 D1 1 2 A2 LxN E/2 E1 N e bxN bbb CAB D/2 NOTES: 1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES). 2. COPLANARITY APPLIES TO THE EXPOSED PAD AS WELL AS TERMINALS. © 2010 Semtech Corporation 25 SC174 Land Pattern - MLPD-10 3x3 K DIM K (C) H (C) H G G Y P DIM Z Z Y X X P C G H K P X Y Z (.114) (2.90) C DIMENSIONS G INCHES .083MILLIMETERS 2.10 1.40 H (.112) .055 (2.85) 2.20 K .079 .087 2.00 1.50 0.50 P .059 .020 2.25 0.30 X .089 .012 .020 0.50 .031 0.80 Y 0.30 .012 3.70 Z .033 .146 0.85 .146 3.70 DIMENSIONS INCHES MILLIMETERS NOTES: NOTES: 1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES). 2. 1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES). 2. THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY. CONSULT YOURMANUFACTURING GROUP TO ENSURE YOUR YOUR MANUFACTURING GROUP TO ENSURE CONSULT YOUR COMPANY'S MANUFACTURING GUIDELINES MET. MET. COMPANY'S MANUFACTURING GUIDELINES ARE ARE THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY. 3. 3.THERMAL VIAS IN THE LAND PATTERN OF THE EXPOSED PAD PAD THERMAL VIAS IN THE LAND PATTERN OF THE EXPOSED SHALL BE CONNECTED TO A A SYSTEM GROUND PLANE. SHALL BE CONNECTED TOSYSTEM GROUND PLANE. FAILURE TO DO SO MAY COMPROMISE THE THERMAL AND/OR FAILURE TO DO SO MAY COMPROMISE THE THERMAL AND/OR FUNCTIONAL PERFORMANCE OF THE DEVICE. FUNCTIONAL PERFORMANCE OF THE DEVICE. © 2010 Semtech Corporation 26 SC174 © Semtech 2010 All rights reserved. Reproduction in whole or in part is prohibited without the prior written consent of the copyright owner. The information presented in this document does not form part of any quotation or contract, is believed to be accurate and reliable and may be changed without notice. No liability will be accepted by the publisher for any consequence of its use. Publication thereof does not convey nor imply any license under patent or other industrial or intellectual property rights. Semtech assumes no responsibility or liability whatsoever for any failure or unexpected operation resulting from misuse, neglect improper installation, repair or improper handling or unusual physical or electrical stress including, but not limited to, exposure to parameters beyond the specified maximum ratings or operation outside the specified range. SEMTECH PRODUCTS ARE NOT DESIGNED, INTENDED, AUTHORIZED OR WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS, DEVICES OR SYSTEMS OR OTHER CRITICAL APPLICATIONS. INCLUSION OF SEMTECH PRODUCTS IN SUCH APPLICATIONS IS UNDERSTOOD TO BE UNDERTAKEN SOLELY AT THE CUSTOMER’S OWN RISK. Should a customer purchase or use Semtech products for any such unauthorized application, the customer shall indemnify and hold Semtech and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs damages and attorney fees which could arise. Contact Information Semtech Corporation Power Mangement Products Division 200 Flynn Road, Camarillo, CA 93012 Phone: (805) 498-2111 Fax: (805) 498-3804 www.semtech.com © 2010 Semtech Corporation 27