SC2441EVB

Very Low Input Voltage 2-Phase Synchronous Step-down Controllers with Step-up Converter POWER MANAGEMENT Description The SC2441 is a high-frequency triple output switching regulator controller. It consists of a dual out-of-phase synchronous step-down PWM controller with high-current output gate drives and a 1.7A integrated step-up switching regulator. The dual-phase step-down controller of the SC2441 can be configured to provide two individually controlled and regulated outputs or a single output with shared current in each phase. The buck controller can operate from an input voltage of at least 4.72V or they can run off a supply generated locally with the integrated boost regulator. This makes the SC2441 ideally suited for applications where a low-voltage input (3.3V, 2.5V, or 1.8V) is to be stepped down for lower voltage logic yet the input is too low to drive power MOSFET’s efficiently. The boost regulator can be used to provide a third auxiliary output while generating the bias for the buck controllers. Both the step-down controllers and the step-up regulator employ fixed frequency peak current-mode control for fast transient response. The master oscillator frequency can be programmed by the user. Individual soft-start and overload shutdown timer are employed in each step-down controller for hiccup overload protection. In single-output configuration, the channel 1 timer controls the soft-start and overload shutdown functions of both controllers. SC2441 Features 2-Phase Synchronous step-down controllers 2-Phase Synchronous Continuous Conduction Mode For High Efficiency Step-down Converters Out of Phase Operation For Low Input Current Ripples Operates Up To 1MHz Per Channel Configurable Dual Outputs Or 2-Phase Single Output Operation with Peak Current Mode Control Excellent Current Sharing Between Phases Wide Input Voltage Range: 1.8V to 15V Duty Cycle Up to 90% 0.5V Feedback Voltages For Low-Voltage Outputs Precision 50mV Current-Limit Threshold Patented Combi-sense Technique for High SNR of Current-Sensing Individual Soft-Start, Overload Shutdown and Enable Step-up Regulator Wide Input Voltage Range: 1.8V to 15V Operates At Twice The Individual Channel Frequency Of The Buck Controllers 0.23V VCESAT Switch at 1A Fixed Frequency with Current-Mode Control Common Features External Synchronization Industrial Temperature Range Applications Low Voltage Distributed DC-DC Converters Telecommunication Power Supplies Servers and base stations VIN VINGND Typical Application Circuit R12 D1 R14 + C38 D2 VIN C36 + C1 C2 R28 7 8 FB3 COMP3 PVIN BST2 GDH2 IN PH3 BST1 GDH1 L3 1 D3 VIN C3 C18 VIN Q1 R3 28 18 27 25 23 R4 + C4 C17 VO1 C8 C25 R9 Q2 L2 C23 R6 R13 RCS+2 C21 R11 L1 C20 R5 C19 R10 D7 Q6 19 VO2 C24 + C15 VO1GND + R7 20 SC2441 GDL2 GDL1 PGND 2 22 24 26 4 5 11 12 C31 GND 6 3 17 SY NC SS1/EN1 SS2/EN2 Rosc VCC 9 10 21 R20 R8 Q7 D10 VO2GND RCS+1 VPN2 CS2+ CS2FB2 COMP2 VPN1 CS1+ CS1FB1 COMP1 16 15 RCS-1 C27 R18 R17 C30 14 13 RCS-2 C28 R19 R16 C32 Figure 1 Revision: January 10, 2005 C33 U1 1 US patent 6,441,597 www.semtech.com SC2441 POWER MANAGEMENT Absolute Maximum Rating Exceeding the specifications below may result in permanent damage to the device, or device malfunction. Operation outside of the parameters specified in the Electrical Characteristics section is not implied. Parameter Input Voltages Supply Voltage For Step-Dow n Controller High-Side Driver Supply Voltages FB1, FB2 Voltage COMP1, COMP2 Voltages CS1(+), CS1(-), CS2(+) and CS2(-) Voltages SY NC/SHDN Voltage ROSC Voltage SS1/EN1 AND SS2/EN2 Voltages Peak Gate Drive Current Peak VPN1 and VPN2 Output Currents FB3 Voltage COMP3 Voltage PH3 Voltage Ambient Temperature Thermal Resistance Junction to Case Thermal Resistance Junction to Ambient Storage Temperature Range Lead Temperature (Soldering) 10 sec Symbol VIN , VPVIN VCC VBST1, VBST2 VFB1, VFB2 VCOMP1, VCOMP2 VCS1(+), VCS1(-), VCS2(+), VCS2(-) VS/S VROSC VSS1, VSS2 IGDH1, IGDH2, IGDL1, IGDL2 IVPN1, IVPN2 VFB3 VCOMP3 VPH3 TA θJC θJA TSTG TLEAD Maximum Ratings -0.3 to 20 -0.3 to 20 -0.3 to 20 -0.3 to 20 -0.3 to 4.5 -0.3 to 20 -0.3 to 20 -0.3 to 5 -0.3 to 6 3 100 4 -0.3 to 2 -0.3 to 35 -40 to 85 13 84 -60 to 150 260 Units V V V V V V V V V A mA V V V °C °C/W °C/W °C °C Electrical Characteristics Unless specified: VIN = 2V, VCC = VBST1 = VBST1 =8V, SYNC/SHDN =2V, ROSC = 51.1kΩ, -40°C < TA = TJ < 85°C Parameter Undervoltage Lockout VCC Start Threshold VCC UVLO Threshold VCC Operating Current Channel 1 and 2 Error Amplifiers Feedback Voltage Feedback Pin Input Bias Current Amplifier Transconductance Open Loop Voltage Gain Amplifier Unity Gain Bandwidth Amplifier Output Sink Current Amplifier Output Source Current COMP Threshold for PWM Operation FB2 Voltage For 2-Phase Single Output Operation  2005 Semtech Corp. Symbol VCCTH VCCTL ICC Conditions VCC Increasing VCC Decreasing VCC = 8V, VS/S = 2V VCC < VCCTL, VS/S = 2V VCC = 8V, VS/S = 0V VIN = 3V VCCTL < VCC < 10V -40°C < TA < 85°C Min Typ 4.65 Max 4.72 21 0.25 13 Units V V mA 4.34 4.45 14 0.15 10 VFB1, VFB2 IFB1, IFB2 GM1, GM2 ao1, ao2 0.487 0.496 -160 400 75 0.507 -400 V nA µΩ −1 dB MHz (Note 1) VFB1, 2 = 1V, VCOMP1,2 = 2.5V VFB1, 2 = 0V, VCOMP1,2 = 2.5V VCS1(+) = VCS1(-) = 0 VCS2(+) = VCS2(-) = 0 -40°C < TA < 85°C 20 10 1.3 1.55 2 5 32 17 1.7 40 30 2.2 µA µA V V www.semtech.com SC2441 POWER MANAGEMENT Electrical Characteristics (Cont.) Unless specified: VIN = 2V, VCC = VBST1 = VBST1 =8V, SYNC/SHDN =2V, ROSC = 51.1kΩ, -40°C < TA = TJ < 85°C Parameter Oscillator Step-dow n Channel Sw itching Frequency Maximum Duty Cycle Minimum Duty Cycle SY NC/SHDN Synchronizing Frequency SY NC/SHDN Input High Voltage SY NC/SHDN Input Low Voltage SY NC/SHDN Input Current Shutdow n Delay Symbol fOSC1, fOSC2 DMAX1, DMAX2 DMIN1, DMIN2 VS/SH VS/SL IS/S Conditions ROSC = 51.1kΩ ROSC = 51.1kΩ ROSC = 51.1kΩ ROSC = 51.1kΩ (Note 1) Min 470 88 Typ 510 90 Max 550 Units KHz % 0 1.2 1.5 0.5 2 % MHz V V µA µs VS/S = 0.2V VS/S = 2V (Note 1) 50 85 1 100 Current-Sense Amplifiers and Current-Limit Comparators Current Limit Threshold Current Limit Threshold Positive Current-Sense Input Bias Current Negative Current-Sense Input Bias Current Minimum PWM On-time Gate Drivers High-Side Gate Drive Peak Source Current High-Side Gate Drive Peak Sink Current Low -Side Gate Drive Peak Source Current Low -Side Gate Drive Peak Sink Current Gate Drive Rise Time Gate Drive Fall Time Low -side Gate Drive to High-side Gate Drive Non-overlapping Delay High-side Gate Drive to Low -side Gate Drive Non-overlapping Delay Soft-Start, Overload Shutoff and Enable Soft-Start Charging Current Soft-Start Voltage to Enable Overload Shutoff Overload Shutoff FB Threshold Soft-Start Discharge Current Soft-Start Voltage to Recover From Overload Shutoff ISS1, ISS2 VSSEN1, VSSEN2 VFBOL1, VFBOL2 ISS1(DIS), ISS2(DIS) VSSRCV1, VSSRCV2 VSS1 = VSS2 = 1.5V VSS1 and VSS2 Increasing VSS1, 2 = 3.8V FB1 and FB2 Decreasing VFB1 = VFB2 = 0.3V VSS1 = VSS2 = 3.8V VSS1 and VSS2 Decreasing 0.29 0.348 2.3 3.25 0.36 1.4 0.47 0.63 0.372 µA V V µA V (Note 1) (Note 1) (Note 1) (Note 1) CL = 3300pF CL = 3300pF CL = 0 CL = 0 2 2 2 2 30 30 74 62 A A A A ns ns ns ns VILIM1, VILIM2 VILIM1, VILIM2 ICS1(+), ICS2(+) ICS1(-), ICS2(-) VCC = 8V VCS1(-) = VCS2(-) = 0V VCC = 8V VCS1(-) = VCS2(-) = 5V VCS1(+) = VCS1(-) = 0 VCS2(-) = VCS2(-) = 0 VCS1(+) = VCS1(-) = 0 VCS2(+) = VCS2(-) = 0 TA = 25°C, (Note 1) 40 40 48 46.5 -0.37 -0.32 180 56 56 -1 -1 mV mV µA µA ns  2005 Semtech Corp. 3 www.semtech.com SC2441 POWER MANAGEMENT Electrical Characteristics (Cont.) Unless specified: VIN = 2V, VCC = VBST1 = VBST1 =8V, SYNC/SHDN =2V, ROSC = 51.1kΩ, -40°C < TA = TJ < 85°C Parameter Gate Drive Disable SS/EN Voltage Gate Drive Enable SS/EN Voltage Virtual Phase Nodes Output High Voltage Output Low Voltage Output High Voltage Output Low Voltage Boost Converter VIN Start Threshold VIN UVLO Threshold Feedback Pin Bias Current Feedback Voltage Feedback Amplifier Transconductance Feedback Amplifier Open-Loop Gain Boost Converter Switching Frequency Maximum Switch Duty Cycle Boost Converter Switch Saturation Voltage Boost Switch Leakage Current Boost Switch Current Limit Symbol Conditions Min 1.2 Typ Max 0.5 Units V V VVPN1,2H VVPN1,2L VVPN1,2H VVPN1,2L VINTH VINTL IFB3 V FB 3 GM3 a o3 fOSC3 DMAX3 VCESAT ILEAKAGE ILIMIT IVPN1=0, IVPN2=0 IVPN1=0, IVPN2=0 IVPN1= IVPN2= -12mA IVPN1= IVPN2= 12mA VIN Increasing VIN Decreasing 1.55V < VIN < 16.5V -40°C < TA < 85°C PVIN 0.05 20 PVIN 0.22 200 1.74 1.45 1.59 40 1.225 1.250 180 50 250 1.275 1.8 V mV V mV V V nA V µΩ-1 dB 1.1 86 0.23 0.35 5 MHz % V µA A ROSC = 51.1kΩ ISW = 1A, TA = 25°C V S W = 12V 0.94 82 1.7 2 Note 1: Guaranteed by design not tested in production.  2005 Semtech Corp. 4 www.semtech.com SC2441 POWER MANAGEMENT Pin Configurations (TOP VIEW) IN VPN2 SS1/EN1 CS1+ CS1SYNC/SHDN FB3 COMP3 GND ROSC FB1 COMP1 COMP2 FB2 Ordering Information D evice SC 2441ITSTRT(1)(2) S C 2441E V B 28 27 26 25 24 23 22 21 20 19 18 17 16 15 PVIN PH3 VPN1 BST1 PGND GDH1 GDL1 VCC GDL2 GDH2 BST2 SS2/EN2 CS2+ CS2- P ackag e TSSOP-28 Temp. R ange( TA) -40 - 85°C Evaluati on Board 1 2 3 4 5 6 7 8 9 10 11 12 13 14 Notes: (1) Only available in tape and reel packaging. A reel contains 2500 devices for the TSSOP-28 package. (2) Lead free product. (28-Pin TSSOP)  2005 Semtech Corp. 5 www.semtech.com SC2441 POWER MANAGEMENT Pin Descriptions Pin 1 2 Pin N ame IN VPN2 Pin Function Power Supply Voltage for the Analog Secti on of the Boost C onverter. The Vi rtual Phase (Unloaded) Node of the Second Step-down C onverter. Used for "C ombi " current sense only. Thi s pi n i s left open when sensi ng current wi th a sense resi stor at the converter output. An external capaci tor ti ed to thi s pi n sets (i ) the soft-start ti me (i i ) output overload latch off ti me for step-down converter 1. Pulli ng thi s pi n below 0.5V shuts off the gate dri vers for the fi rst controller. The Non-i nverti ng Input of the C urrent-sense Ampli fi er/C omparator for the Step-down C ontroller 1. The Inverti ng Input of the C urrent-sense Ampli fi er/C omparator for the Step-down C ontroller 1. Normally ti ed to the output of the converter. Synchroni zati on and Shutdown Input. For normal operati on, ti e thi s pi n to a voltage above 1.5V. To shut-off both step-down controllers and the boost regulator, force thi s pi n to a voltage less than 0.5V. The master osci llator can be synchroni zed by dri vi ng thi s pi n wi th an external clock (external fCLK > frequency set wi th ROSC ). The boost converter runs at the external clock frequency whereas the step-down controllers operate at half the clock frequency. The Inverti ng Input of the Error Ampli fi er for the Boost C onverter. FB3 i s ti ed to an external resi sti ve di vi der for output3 voltage setti ng. The Error Ampli fi er Output of the Boost C onverter. Thi s pi n i s used for loop compensati on. Pulli ng thi s pi n below 0.4V di sables the step-up converter. Analog Si gnal Ground. An external resi stor connected from thi s pi n to GND sets the osci llator frequency. The Inverti ng Input of the Error Ampli fi er for the Step-down C ontroller 1. Ti e to an external resi sti ve di vi der between OUTPUT1 and the ground for output voltage sensi ng. The Error Ampli fi er Output for Step-down C ontroller 1. Thi s pi n i s used for loop compensati on. The Error Ampli fi er Output for Step-down C ontroller 2. Thi s pi n i s used for loop compensati on. The Inverti ng Input of the Error Ampli fi er for the Step-down C ontroller 2. Ti e to an external resi sti ve di vi der between output2 and the ground for output voltage sensi ng. Ti e to IN or VC C for two-phase si ngle output appli cati ons The Inverti ng Input of the C urrent-sense Ampli fi er/C omparator for the Step-down C ontroller 2. Normally ti ed to the output of the converter. The Non-i nverti ng Input of the C urrent-sense Ampli fi er/C omparator for the Step-down C ontroller 2 An external capaci tor ti ed to thi s pi n sets (i ) the soft-start ti me (i i ) output overload latch off ti me for step-down converter 2. Pulli ng thi s pi n below 0.5V shuts off the gate dri vers for the second controller. Leave open for two-phase si ngle output appli cati ons. Bootstrapped Supply for the Hi gh-si de Gate D ri ve 2. C onnect to a bootstrap capaci tor and an external di ode as descri bed i n appli cati on i nformati on. Gate D ri ve Output for the Hi gh-si de N-channel MOSFET of Output 2. Gate dri ve voltage swi ngs from ground to VBST2. Gate D ri ve Output for the Low-si de N-channel MOSFET of Output 2. Gate dri ve voltage swi ngs from ground to VC C . 3 SS1/EN1 4 5 C S 1+ C S 1- 6 SYNC /SHD N 7 8 9 10 11 12 13 FB 3 C OMP3 GND ROSC FB 1 C OMP1 C OMP2 14 FB 2 15 16 C S 2C S 2+ 17 SS2/EN2 18 19 20 BST2 GD H2 GD L2  2005 Semtech Corp. 6 www.semtech.com SC2441 POWER MANAGEMENT Pin Descriptions 21 VCC Supply Voltage for Both Step-dow n Controllers and the Low -side Gate Drivers. The boost converter output is tied to VCC if VIN in not high enough to fully enhance the pow er MOSFET's and the boost converter provides an auxiliary supply voltage for the step-dow n controllers. Tie VCC to VIN if the boost converter is not needed. Gate Drive Output for the Low -side N-channel MOSFET of Output 1. Gate drive voltage sw ings from ground to VCC. Gate Drive Output for the High-side N-channel MOSFET of Output 1. Gate drive voltage sw ings from ground to VBST1. Ground Supply of the High-side and the Low -side Gate Drivers of Both Step-dow n Controllers. It is also the emitter of the boost sw itch. Bootstrapped Supply for the High-side Gate Drive 1. Connect to a bootstrap capacitor and an external diode as described in application information. The Virtual Phase (Unloaded) Node of the First Step-dow n Converter. Used for "Combi" current sense only. This pin is left open w hen sensing current w ith a sense resistor at the converter output. Boost Sw itch Collector. Connect to a boost inductor and a rectifying diode. Pow er Supply Voltage for the Boost Sw itch and the Virtual Phase Node Drivers. 22 23 24 25 GDL1 GDH1 PGND BST1 26 27 28 VPN1 PH3 PVIN  2005 Semtech Corp. 7 www.semtech.com SC2441 POWER MANAGEMENT Block Diagram V IN UVLO 1.60/1.74V IN 1 P VIN CLK2 OSCILLATOR FREQUENCY CLK1 DIVIDER S LOPE C OMP S HDN S HDN S LOPE2 28 0.5V 0.36V VCC 21 UVLO 4.45/4.65V BST1 25 GDH1 23 R S Q Non-Overlapping Conduction Control FAULT V PN1 26 GDL1 22 1.25V S YNC/SHDN 6 ROSC 10 COMP1 12 FB1 11 0.5V CLK REFERENCE - + EA1 S LOPE1 + P WM1 CS1+ 4 CS15 + - ISEN1 + + Σ + Soft-Start And Overload Hiccup Control 1 OL DSBL P GND 24 S S1/EN1 3 B A Y S EL 50mV 1.25V - ILIM1 I + S EL ANALOG SWITCH CLK2 BST2 18 GDH2 COMP2 13 FB2 14 0.5V - 19 EA2 + S LOPE2 R S Q Non-Overlapping Conduction V CC Control FAULT P VIN V PN2 2 GDL2 20 + P WM2 CS2+ 16 CS215 + - ISEN2 + + Σ + Soft-Start And Overload Hiccup Control 2 OL DSBL S S2/EN2 17 GND 9 50mV - ILIM2 I Step-down Controllers Functional Diagram Figure 2  2005 Semtech Corp. 8 www.semtech.com SC2441 POWER MANAGEMENT Block Diagram COMP3 8 FB3 7 1.25V CLK PH3 27 + EA3 - R S Q + PWM3 I LIM3 SLOPE COMP + - 14mV 7mΩ + Σ + + I SEN3 - 24 Step-up Converter Functional Diagram Figure 3 PGND FB 0.36V SS/EN + 2 µΑ S Q R 0.47V/3.25V OL DSBL FAULT 0.6V/0.9V 3 µΑ Details of the Soft-start and Overload Hiccup Control Circuit Figure 4  2005 Semtech Corp. 9 www.semtech.com SC2441 POWER MANAGEMENT Typical Performance Characteristics FB1 AND FB2 VOLTAGES vs TEMPERATURE 0.5 V IN = 2V 0.498 VOLTAGE (V) V CC = 5V 1.3 VIN START AND UVLO THRESHOLD VOLTAGES vs TEMPERATURE 2 FB3 VOLTAGE vs TEMPERATURE 1.28 VOLTAGE (V ) VOLTAGES (V) 1.9 0.496 1.26 1.8 VIN START 1.7 VIN UVLO 0.494 1.24 0.492 1.22 1.6 0.49 -50 -25 0 25 50 75 100 TEMPERATURE (ºC) 1.2 -50 -25 0 25 50 75 100 TEMPERATURE (ºC) 1.5 -50 -25 0 25 50 75 100 TEMPERATURE (ºC) VCC START AND UVLO THRESHOLD VOLTAGES vs TEMPERATURE 5 520 515 4.8 FREQUENCY (KHz ) VCC START VOLTAGES (V) 4.6 VCC UVLO 4.4 510 505 500 495 490 485 4 -50 -25 0 25 50 75 100 TEMPERATURE (ºC) 480 -50 STEP-DOWN CHANNEL SWITCHING FREQUENCY vs TEMPERATURE ROSC = 51.1KΩ STEP-UP CONVERTER SWITCHING FREQUENCY vs TEMPERATURE 1050 1040 1030 FREQUENCY (KHz ) 1020 1010 1000 990 980 970 960 950 ROSC = 51.1KΩ 4.2 -25 0 25 50 75 100 -50 -25 TEMPERATURE (ºC) 0 25 50 TEMPERATURE (ºC) 75 100 MAXIMUM DUTY- CYCLE OF STEP-DOWN CHANNEL vs TEMPERATURE 95 ROSC = 51.1KΩ 93 DUTY CYCLE (%) DUTY CYCLE (%) STEP-UP SWITCH MAXIMUM DUTY- CYCLE vs TEMPERATURE 90 ROSC = 51.1KΩ 88 VOLTAGES (mV) 48 50 STEP-DOWN CONTROLLER CURRENT-LIMIT THRESHOLD vs TEMPERATURE CS(-) = 0 91 86 46 CS(-) = 5V 89 84 44 87 82 42 VCC = 8V 85 -50 -25 0 25 50 75 100 TEMPERATURE (ºC) 80 -50 -25 0 25 50 75 100 TEMPERATURE (ºC) 40 -50 -25 0 25 50 75 100 TEMPERATURE (ºC)  2005 Semtech Corp. 10 www.semtech.com SC2441 POWER MANAGEMENT Operation Overview The SC2441 is a constant frequency triple-output switching regulator especially designed for operating from input voltages as low as 1.8V. It consists of two currentmode step-down switch-mode PWM controllers driving all N-channel MOSFET’s and a 1.7A step-up current-mode controller with integrated 1.7A power switch. A low voltage input (3.3V, 2.5V or 1.8V) can be stepped up to 5V-10V locally using the boost regulator to provide sufficient gate drives for the step-down converters. The boost converter can also be used to generate a third output. The two step-down channels of the SC2441 operate at 180 degrees out of phase from each other. Since input currents are interleaved in a two-phase converter, input ripple current is lower and smaller input capacitor can be used for filtering. The step-down controllers of the SC2441 operate in synchronous continuous-conduction mode. They can be configured either as two independent step-down controllers producing two separate outputs or as a dualphase single-output controller by tying the FB2 pin to VCC. In single output operation, the channel-one error amplifier controls both channels and the channel-two error amplifier is disabled. Soft-start and overload hiccup of both channels is also synchronized to channel one. Frequency Setting and Synchronization The step-up regulator in the SC2441 runs at twice the frequency of step-down controllers. Each step-down controller runs at one-half of the oscillator frequency and is 180 degrees out of phase from the other step-down controller. The switching frequency of the step-up regulator is the oscillator frequency and can be set with an external resistor from the ROSC pin to the ground. The boost regulator and the step-down controllers are capable of operating up to 2 MHz and 1 MHz respectively. It is necessary to consider the operating duty-ratio range before deciding the switching frequency. See Applications Information section for more details. When synchronized externally, the applied clock frequency (hence switching frequency of the step-up converter) should be twice the individual phase frequency of the step-down controllers. The synchronizing clock frequency  2005 Semtech Corp. should also be between 1-1.33 times the set free-running frequency. If not synchronized, the SYNC/SHDN pin should be tied to the input. Pulling the SYNC/SHDN pin below 0.5V shuts off the SC2441 after 85µs time delay. Control Loop The step-down controllers and the boost regulator in the SC2441 use peak current-mode control for fast transient response, ease of compensation and current sharing in single output operation. The low-side MOSFET of each step-down channel is turned off at the falling-edge of the phase clock. After a brief non-overlapping conduction interval of 74ns, the high-side MOSFET is turned on. The phase inductor current ramps up. When the sensed inductor current reaches the threshold determined by the error amplifier output and ramp compensation, the high-side MOSFET is turned off. After a non-overlapping delay of 62ns, the low-side MOSFET is turned on. The supply voltages for the high-side gate drivers are obtained from two diode-capacitor bootstrap circuits. If the bootstrap capacitor is charged from VCC, then the high-side gate drive voltage will swing from approximately 2VCC to the ground. The outputs of the low-side gate drivers swing from VCC to the ground. All three converters in the SC2441 have internal rampcompensation to prevent sub-harmonic oscillation when operating above 50% duty cycle. The internal compensating ramp is designed for an inductor ripplecurrent of between ¼ to 1/3 of the maximum inductor current and the peak-to-peak current-sense voltage (CSPCSN of the step-down controllers) of between ¼ to 1/3 of the current-limit threshold (50mV). The current-limits of all three converters are unaffected by the compensation ramps. Current-Sensing Since the inductor current ramp is used as the modulating ramp in current-mode control, the inductor current needs to be sensed. There are two current sensing methods for the step-down controllers. Since the maximum currentsense voltage (CSP-CSN) is only 50mV, a precision sense resistor in series with the inductor can be used at the output without resulting in excessive power dissipation. 11 www.semtech.com SC2441 POWER MANAGEMENT Operation (Cont.) Although accurate and far easier to lay out than highside resistor sensing, a pair of precision sense resistors adds cost to the converter. The SC2441 has provision to reconstruct a differential voltage proportional to the inductor current at the output of the converter. The voltage to current ratio or the equivalent sense resistance Req is a combination of high-side and low-side MOSFET RDS(ON) ’s and the inductor series resistance (hence the name “combi-sense”). The SC2441 provides the virtual phase voltages VPN1 and VPN2 (these are unloaded versions of their respective phase voltages) for current sensing. This method does not require precision sense resistor. It is cheaper to implement but is less accurate than resistor current sensing. Since the sensed voltage is developed at the output of the step-down converter, it is less prone to switching transient spikes. This method will be described in more details in the Applications Information section. Boost switch current is sensed with an integrated sense resistor with a current-limit of 1.7A. Error Amplifiers All error amplifiers in the SC2441 are transconductance amplifiers. Converters are compensated with series RC network from the COMP pins to the ground. An additional small parallel capacitor may be required for stability. In closed loop operation, the step-down error amplifiers output range from 1.7V to 3.5V. There is no control (highside) gate drive until the COMP voltage exceeds 1.6V. Both non-inverting inputs of the feedback amplifiers are tied to an internal 0.5V voltage reference. The error amplifier of the step-up converter has 1.25V as its reference voltage. Its output voltage ranges from 0.8V to 1.35V in closed-loop operation. Current-Limit The maximum current sense voltage of +50mV is the cycle-by-cycle peak current limit when the load is drawing current from the converter. Soft-Start and Overload Protection The undervoltage lockout circuit discharges the SS/EN capacitors. After VCC r ises above 4.65V, the SS/EN capacitors are slowly charged by internal 2.3µA current sources. With internal PNP transistors, the SS/EN voltages clamp the error amplifier outputs. When the error amplifier output rises to 1.7V, the high-side MOSFET starts to switch. As the SS/EN capacitor continues to charge, the COMP voltage follows. The converter gradually delivers increasing power to the output. The inductor current follows the COMP voltage envelope until the output goes into regulation. The SS/EN clamp on COMP is released. After the SS/EN capacitor is charged above 3.25V (high enough for the error amplifier to provide full load current), the overload detection circuit is activated. If the output voltage falls below 70% of its set value, an overload latch will be set and both the top and the bottom MOSFET’s will be turned off. The SS/EN capacitor is slowly discharged with an internal 1.4 µ A current sink. The overload latch is reset when the SS/EN capacitor is discharged below 0.47V. The SS/EN capacitor is then recharged with the 2.3µA current source and the converter undergoes soft-start. If overload persists, the step-down converters will undergo repetitive shutdown and restart (hiccup). Soft-start process should be slow enough to allow the output to reach 70% of its final value before the SS/ EN capacitor is charged above 3.25V (see Figure 4). If the output is short-circuited, the inductor current will not increase indefinitely between the time the inductor current reaching its current limit and shutdown. This is due to cycle skipping reduces the actual operating frequency. The SS/EN pin can also be used as the enable input for that channel. Both the high-side and the low-side MOSFET’s will be turned off if the SS/EN pin is pulled below 0.5V.  2005 Semtech Corp. 12 www.semtech.com SC2441 POWER MANAGEMENT Applications Information The SC2441 consists of two current-mode synchronous buck controllers and an auxiliary boost converter. The SC2441 can be used to generate 1) two independent step-down outputs or 2) dual phase single output with current sharing and 3) a step-up output The application information using SC2441 for the control of step-down and step-up converters are described below. Step-down Converter Specifications of a step-down converter are given by the followings Input voltage range: Vin ∈ [ Vin,min , Vin,max ] Input voltage ripple (peak-to-peak): ∆Vin Output voltage: Vo Output voltage accuracy: ε Output voltage ripple (peak-to-peak): ∆Vo Nominal output (load) current: Io Maximum output current limit: Io,max Output (load) current transient slew rate: dIo (A/s) Circuit efficiency: η. Based on these converter specifications, selection criteria and design procedures for the following components are described. 1) output inductor (L) type and value, 2) output capacitor (Co) type and value, 3) input capacitor (Cin) type and value, 4) power switch MOSFET’s, 5) current sensing and limiting circuitry, 6) voltage sensing circuitry, 7) loop compensation circuitry. To illustrate the design process, the following example is used: Vin=3.3V, Vo=1.2V, Io=4A, fs=500kHz. Operating Frequency (fs) The switching frequency in the SC2441 is userprogrammable. The advantages of constant frequency operation are simple passive component selection and fast transient response with simple frequency compensation. Before setting the operating frequency, the following tradeoffs should be considered. 1) passive component sizes 2) converter efficiency 3) EMI 4) Minimum switch on time and 5) Maximum duty ratio For a given output power, the sizes of the passive components are inversely proportional to the switching frequency, whereas MOSFET’s/Diodes switching losses are proportional to the operating frequency. Other issues such as heat dissipation, packaging and the cost issues are also to be considered. The frequency bands for signal transmission should be avoided because of EM interference. Minimum Switch On Time Limitation In both step-down controllers, the falling edge of the clock turns on the top MOSFET. The inductor current ramps up so does the sensed voltage. After the sensed voltage crosses a threshold determined by the error amplifier output, the top MOSFET is turned off. The propagation delay time from the turn-on of the controlling FET to its turn-off is the minimum switch on time. The SC2441 has a minimum on time of about 180ns at room temperature. This is the shortest on interval of the controlling FET. The controller either does not turn on the top MOSFET at all or turns it on for at least 180ns. For a synchronous step-down converter, the operating duty cycle is V /V . So the required on time for the top o IN MOSFET is V /(V fS). If the frequency is set such that o IN the required pulse width is less than 180ns, then the converter will start skipping cycles. Due to minimum on time limitation, simultaneously operating at very high switching frequency and very short duty cycle is not practical. If the input voltage is 3.3V and the operating frequency is 1MHz, the lowest output voltage will be 0.6V. There will not be enough modulation headroom if the on time is simply made equal to the minimum on time of the SC2441. For ease of control, we recommend the required pulse width to be at least 1.5 times the minimum on time. Maximum Duty-cycle Consideration When operating at 500KHz, the maximum top MOSFET on duty-cycle is 90%. The top MOSFET therefore turns off for at least 200ns every cycle regardless of the switching frequency. This places an upper bound on the voltage 13 www.semtech.com  2005 Semtech Corp. SC2441 POWER MANAGEMENT Applications Information (Cont.) conversion ratio at a given switching frequency. If the desired output voltage requires high operating duty-cycle, then operating frequency will have to be lowered to allow modulating headroom. Setting the Step-down Channel Frequency The switching frequency of both step-down controllers is set with an external resistor from Pin 10 to the ground. The set frequency is inversely proportional to the resistor value (Figure 5). L= Vo (1 − D) . δIo fs The peak current in the inductor becomes (1+δ/2)*Io and the RMS current is IL,rms = Io 1 + δ2 . 12 800 700 600 fs (kHz) 500 400 300 200 100 0 0 50 100 150 200 250 Rosc (k Ohm) The followings are to be considered when choosing inductors. a) Inductor core material: For high efficiency applications above 350KHz, ferrite, Kool-Mu and polypermalloy materials should be used. Low-cost powdered iron cores can be used for cost sensitive-applications below 350KHz but with attendant higher core losses. b) Select inductance value: Sometimes the calculated inductance value is not available off-the-shelf. The designer can choose the adjacent (larger) standard inductance value. The inductance varies with temperature and DC current. It is a good engineering practice to re-evaluate the resultant current ripple at the rated DC output current. c) Current rating: The saturation current of the inductor should be at least 1.5 times of the peak inductor current under all conditions. Output Capacitor (Co) and Vout Ripple Figure 5. Step-down Channel Free-running frequency vs. ROSC. Inductor (L) and Ripple Current Both step-down controllers in the SC2441 operate in synchronous continuous-conduction mode (CCM) regardless of the output load. The output inductor selection/design is based on the output DC and transient requirements. Both output current and voltage ripples are reduced with larger inductors but it takes longer to change the inductor current during load transients. Conversely smaller inductors results in lower DC copper losses but the AC core losses (flux swing) and the winding AC resistance losses are higher. A compromise is to choose the inductance such that peak-to-peak inductor ripple-current is 20% to 30% of the rated output load current. Assuming that the inductor current ripple (peak-to-peak) is δ*Io, the inductance will then be  2005 Semtech Corp. The output capacitor provides output current filtering in steady state and serves as a reservoir during load transient. The output capacitor can be modeled as an ideal capacitor in series with its parasitic ESR (Resr) and ESL (Lesl) (Figure 6). Co Lesl Resr Figure 6. Co equivalent circuit If the current through the branch is ib(t), the voltage across the terminals will then be 14 www.semtech.com SC2441 POWER MANAGEMENT Applications Information (Cont.) di ( t ) 1 v o ( t ) = Vo + ib ( t )dt + L esl b + R esr ib ( t ). Co 0 dt ∫ t δIo 23 . This basic equation illustrates the effects of ESR, ESL and Co on the output voltage. The first term is the DC voltage across Co at time t=0. The second term is the ripple-voltage caused by the inductor ripple-current. The third term is the voltage ripple due to ESL and the fourth term is the voltage ripple due to ESR. The total output voltage ripple is then a vector sum of the last three terms. Since the inductor current is a triangular waveform with peak-to-peak value δ*Io, the ripple-voltage caused by inductor current ripples is ∆v C ≈ δIo . 8Co fs δIo D Usually it is necessary to have several capacitors of the same type in parallel to satisfy the ESR requirement. The voltage ripple cause by the capacitor charge/discharge should be an order of magnitude smaller than the voltage ripple caused by the ESR. To guarantee this, the capacitance should satisfy Co > 10 . 2πfsR esr The ripple-voltage due to ESL is ∆v ESL = L esl fs In many application circuits, several low ESR ceramic capacitors are added in parallel with the aluminum capacitors to further reduce ESR and improve high frequency decoupling. Since the capacitances and the ESR’s of ceramic and aluminum capacitors are different, the following remarks are made to clarify some practical issues. Remark 1: High frequency ceramic capacitors may not carry most of the ripple current. It also depends on the capacitor value. Only when the capacitor value is set properly, the effect of ceramic capacitor low ESR starts to be significant. For example, if a 10 µ F, 4m Ω c eramic capacitor is connected in parallel with 2x1500µF, 90mΩ electrolytic capacitors, the ripple current in the ceramic capacitor is only about 42% of the current in the electrolytic capacitors at the ripple frequency. If a 100µF, 2mΩ ceramic capacitor is used, the ripple current in the ceramic capacitor will be about 4.2 times of that in the electrolytic capacitors. When two 100µF, 2mΩ ceramic capacitors are used, the current ratio increases to 8.3. In this case most of the ripple current flows in the ceramic decoupling capacitor. The ESR of the ceramic capacitors will then determine the output ripple-voltage. Remark 2: The total equivalent capacitance of the filter bank is not simply the sum of all the paralleled capacitors. The total equivalent ESR is not simply the parallel combination of all the individual ESR’s either. Instead they should be calculated using the following formulae. C eq (ω) := (R1a + R1b )2 ω2C1a C1b + (C1a + C1b )2 (R1a C1a + R1b C1b )ω2 C1a C1b + (C1a + C1b ) 2 2 2 2 and the ESR ripple-voltage is ∆v ESR = R esr δIo . Aluminum capacitors (e.g. electrolytic, solid OS-CON, POSCAP, tantalum) have high capacitances and low ESL’s. The ESR has the dominant effect on the output ripple voltage. It is therefore very important to minimize the ESR. When determining the ESR value, both the steady state ripple-voltage and the dynamic load transient need to be considered. To keep the steady state output ripple-voltage < ∆Vo, the ESR should satisfy R esr1 < ∆Vo . δIo To limit the dynamic output voltage overshoot/undershoot within α (say 3%) of the steady state output voltage) under 0 to full load current swing, the ESR value should be R esr 2 αVo < . Io The required ESR value of the output capacitors should be Resr = min{Resr1,Resr2 }. In the aluminum capacitor selection, the working voltage rating is normally suggested to be greater than 1.5Vo. The allowable current ripple (RMS) should be greater than  2005 Semtech Corp. 15 www.semtech.com SC2441 POWER MANAGEMENT Applications Information (Cont.) R eq (ω) := R1aR1b (R1a + R1b )ω2C1a C1b + (R1b C1b + R1a C1a ) (R1a + R1b )2 ω2 C1a C1b + (C1a + C1b )2 2 2 2 2 2 2 In Figure 8 the DC input voltage source has an internal impedance Rin and the input capacitor Cin has an ESR denoted as Resr. MOSFET and input capacitor current waveforms, ESR voltage ripple and input voltage ripple are shown in Figure 9. where R 1a a nd C 1a a re the ESR and capacitance of electrolytic capacitors, and R1b and C1b are the ESR and capacitance of the ceramic capacitors respectively (Figure 7). C1a C1b Ceq R1a R1b Req Figure 7. Equivalent RC branch. Req and Ceq are both functions of frequency. For rigorous design, the equivalent ESR should be evaluated at the ripple frequency for voltage ripple calculation when both ceramic and electrolytic capacitors are used. If R1a = R1b = R1 and C1a = C1b = C1, then Req and Ceq will be frequencyindependent and Req = 1/2 R1 and Ceq = 2C1. Input Capacitor (Cin) The input supply to the converter usually comes from a pre-regulator. Since the input supply is not ideal, input capacitors are needed to filter the current pulses at the switching frequency. A simple buck converter is shown in Figure 8. Figure 9. Typical waveforms at the input of a buck converter. It can be seen that the current in the input capacitor pulses with high di/dt. Capacitors with low ESL should be used. It is also important to place the input capacitor close to the MOSFET’s on the PC board to reduce trace inductances around the pulse current loop. The RMS value of the capacitor current is approximately ICin = Io D[(1 + δ2 D D )(1 − )2 + 2 (1 − D) ]. η 12 η The power losses at the input capacitors is then PCin = ICin2Resr. For reliable operation, the maximum power dissipation in the capacitors should not result in more than 10oC of temperature rise. Many manufacturers specify the maximum allowable ripple current (ARMS) rating of the capacitor at a given ripple frequency and ambient temperature. The input capacitance should be high enough to handle the ripple current. For higher power applications, multiple capacitors are placed in parallel to increase the ripple current handling capability. Sometimes meeting tight input voltage ripple specifications may require the use of larger input capacitance. At full load, the peak-to-peak input voltage ripple due to the ESR is 16 www.semtech.com Figure 8. Buck converter input model  2005 Semtech Corp. SC2441 POWER MANAGEMENT Applications Information (Cont.) δ ∆v ESR = R esr (1 + )Io . 2 If D1>0.5 and D2 > 0.5, then ICin ≈ (D1 + D 2 − 1)(Io1 + Io 2 )2 + (1 − D 2 )Io1 + (1 − D1 )Io2 . 2 2 The peak-to-peak input voltage ripple due to the capacitor is ∆v C ≈ DIo . Cin fs Power MOSFET Selection and Gate Drive Main considerations in selecting the MOSFET’s are power dissipation, cost and packaging. Switching losses and conduction losses of the MOSFET’s are directly related to the total gate charge (Cg) and channel on-resistance (Rds(on)). In order to judge the performance of MOSFET’s, the product of the total gate charge and on-resistance is used as a figure of merit (FOM). Transistors with the same FOM follow the same curve in Figure 10. From these two expressions, CIN can be found to meet the input voltage ripple specification. In a multi-phase converter, channel interleaving can be used to reduce ripple. The two step-down channels of the SC2441 operate at 180 degrees from each other. If both step-down channels in the SC2441 are connected in parallel, both the input and the output RMS currents will be reduced. Ripple cancellation effect of interleaving allows the use of smaller input capacitors. When converter outputs are connected in parallel and interleaved, smaller inductors and capacitors can be used for each channel. The total output ripple-voltage remains unchanged. Smaller inductors speeds up output load transient. When two channels with a common input are interleaved, the total DC input current is simply the sum of the individual DC input currents. The combined input current waveform depends on duty ratio and the output current waveform. Assuming that the output current ripple is small, the following formula can be used to estimate the RMS value of the ripple current in the input capacitor. Let the duty ratios and output currents of Channel 1 and Channel 2 be D1, D2 and Io1, Io2 respectively. If D1> Z2. To simplify design, one usually assumes that C3