SC416

SC416 POWER MANAGEMENT Features VIN Range 3-25V Outputs Adjustable from 0.75 to 5.25V, or Preset Output Voltages: VOUT1 = 1.8 or 1.5V VOUT2 = 1.25 or 1.05V Proportional or Coincident Tracked Start-up and Shutdown of Both Outputs Adjustable Soft-start Rates for Each Output Regulated Shutdown for Each Output Low Shutdown Power Constant On-Time for Fast Dynamic Response Adjustable Switching Frequency Separated Frequencies for Minimal Switching Interaction: VOUT1 = up to 600kHz VOUT2 = up to 720kHz Power Save or Continuous Operation at Light Load Over-Voltage and Under-Voltage Fault Protection Cycle-by-Cycle Valley Current Limit DC Current Sense Using Low-Side RDSON Sensing, or RSENSE In Source of Low-Side MOSFET for Greater Accuracy Separate Power Good Outputs Separate Enable/Power Save Inputs 3.1A Non-Overlapping Gate Drive SmartDriveTM for High-side MOSFET MLP 4x4 24 Pin Lead-free Package Industrial Temperature Range WEEE and RoHS Compliant Dual Synchronous Buck Controller with Tracking Start-up/Shutdown Description The SC416 is a versatile, constant on-time, pseudofixed-frequency, dual synchronous buck PWM controller intended for notebook computers and other battery operated portable devices. The SC416 contains all the features needed to provide cost-effective control of two independent switchmode power supplies. The SC416 provides adjustable voltage tracking during start-up and shut-down, making it an effective solution for a variety of applications where the voltage differential between supply rails must meet defined limits during all operating conditions including start-up and shutdown. The SC416 supports proportional tracking which gives equal start-up and shut-down times for both outputs, and can also provide coincident tracking where the two output voltages are equal during the lower voltage’s start-up and shut-down ramp. The two DC outputs are adjustable from 0.75V to 5.25V. Additional features for each output include cycle-bycycle current limit, voltage soft-start, under-voltage and over-voltage protection, programmable over-current protection, regulated shutdown, selectable power save and non-overlapping gate drive. The SC416 provides two enable/power save inputs, two soft-start inputs, two power good outputs and an on-time adjust input. The constant on-time topology provides fast dynamic response. The excellent transient response means that SC416 based solutions require less output capacitance than competing fixed frequency converters. Switching frequency is constant until a step in load or line voltage occurs, at which time the pulse density and frequency moves to counter the change in output voltage. After the transient event, the controller frequency returns to steady state operation. At light loads with power save enabled, the SC416 reduces switching frequency for improved efficiency. • • • • Applications Notebook and Sub-Notebook Graphics Voltage Controllers Tablet PCs Embedded Applications May 17, 2007 © 2007 Semtech Corporation 1 SC416 Typical Application Circuit PGD1 VIN +5V Q1 CIN1 10K PGD2 10K +5V Q3 CIN2 VIN 20 RLIM1 D2 VOUT1 L1 1 +5V + COUT1 D1 Q2 2 3 4 5 19 24 23 22 21 1UF PGD1 PGD2 DH1 ILIM1 ILIM2 DH2 PAD LX2 BST2 RLIM2 25 18 17 16 15 14 13 1UF D3 L2 VOUT2 LX1 BST1 VDD1 DL1 EN1 +5V Q4 COUT2 + D4 U1 SC416 VDD2 DL2 EN2 VOUT1 FB1 TON RTN SS1 VOUT2 1UF 10NF EN1 6 SS2 1UF FB2 10NF 10 11 12 7 8 9 EN2 VOUT1 R1 R3 R2 RTON R4 VOUT2 1UF 1UF 330PF VIN 2 SC416 Pin Configuration PGD1 PGD2 ILIM1 ILIM2 DH1 DH2 Ordering Information Device SC416MLTRT(1)(2) SC416EVB 18 Package MLPQ-24 4X4 Evaluation Board 24 19 LX1 BST1 1 LX2 BST2 TOP VIEW VDD1 VDD2 SC416 DL1 EN1 SS1 6 7 12 DL2 GND (PAD) 13 Notes: (1) Available in tape and reel only. A reel contains 3,000 devices. (2) Available in lead-free package only. Device is WEEE and RoHS compliant. EN2 SS2 VOUT1 RTN TON FB2 MLPQ24: 4x4 24 LEAD Marking Information SC416 yyww xxxxx xxxxx nnnn = Part Number (example: SC416 yyww = Date Code (example: 0652) xxxxx = Semtech Lot No. (example: 09010 xxxxx 01-10) VOUT2 FB1 3 SC416 Absolute Maximum Ratings DHx, BSTx to GND (DC) ……….……………….……. -0.3 to +30V DHx, BSTx to GND (transient - 100nsec max) …………-2.0 to +33V LXx, TON to GND (DC) ……………………………… -0.3 to +25V LXx, TON to GND (transient - 100nsec max) ………… -2.0 to +28V BST1 to LX1, BST2 to LX2 (DC) ……………………… -0.3 to +6.0V BST1 to LX1, BST2 to LX2 (transient - 100nsec max) …..-0.3 to +7.5V DLx to GND (DC) ……………………………………-0.3 to +6.0V GND to RTN ……………………………………… -0.3 to +0.3V VDDx to RTN ……………………………………… -0.3 to +6.0V ENx, FBx, ILIMx, PGDx, SSx, VOUTx to RTN…… -0.3 to VDDx +0.3V Thermal Information Junction to Ambient(1) ……………………………… 29°C/W Storage Temperature Range ……………… -60 to +150°C Operating Junction Temperature Range … -40 to +125°C Lead Temperature (Soldering) 10 sec …….…………… 260°C Peak IR Reflow Temperature(10-40s)…………………. ESD Protection Level(2) ………………………………… 260°C 2kV Exceeding the above specifications may result in permanent damage to the device or device malfunction. Operation outside of the parameters specified in the Electrical Characteristics section is not recommended. NOTES(1) Calculated from package in still air, mounted to 3” x 4.5”, 4 layer FR4 PCB with thermal vias under the exposed pad per JESD51 standards. (2) Tested according to JEDEC standard JESD22-A114-B. Electrical Characteristics Test Conditions: VIN = 15V, VOUT = 1.5V, TA = 25 oC, 0.1% resistor dividers; RTON = 1Meg; VDD1/2 = 5.0V; GND connects to PAD pin. 25°C Parameter Input Supplies VIN Input Voltage VDDx Input Voltage VDD1 + VDD2 Shutdown Current VDD1 + VDD2 Operating Current EN1, EN2 = 0V EN1, EN2 = 5V (Powersave Mode) FB1, FB2 > REF 3.0 4.5 7 800 25 5.5 -40° to 85°C Max Min Max Units Conditions Min Typ V V 10 1200 μA μA Regulation FB1, FB2 On-Time Threshold VOUTx Output Voltage Range 0 to 85°C -40 to 85°C External Resistors FB1 = 5V; 0 to 85°C FB1 = 5V; -40 to 85°C VOUT1 On-Time Threshold FB1 = RTN; 0 to 85°C FB1 = RTN; -40 to 85°C 1.5 1.5 1.4812 1.4737 1.5188 1.5263 V 1.8 0.75 0.7425 0.7388 0.75 1.7775 1.7685 0.7575 0.7612 5.25 1.8225 1.8315 V V V 4 SC416 Electrical Characteristics (continued) 25°C Parameter Regulation (Continued) FB2 = 5V; 0 to 85°C FB2 = 5V; -40 to 85°C VOUT2 On-Time Threshold FB2 = RTN; 0 to 85°C FB2 = RTN; -40 to 85°C VOUTx Line Regulation Error VOUTx Load Regulation Error 1.05 1.05 0.04 0.3 1.0368 1.0316 1.0631 1.0684 V %/V % 1.25 1.25 1.2343 1.2281 1.2656 1.2718 V -40° to 85°C Max Min Max Units Conditions Min Typ Timing VOUTx On-Time(1) VOUTx Set to 1.5V VOUT1; RTON = 1Meg VOUT2; RTON = 1Meg VOUT1; RTON = 499K VOUT2; RTON = 499K DH1, DH2 DL1, DL2 400 330 200 167 50 330 320 260 480 400 nsec Minimum On-Time Minimum Off-Time ns ns Soft-Start/Shutdown Soft-Start SSx Current Source Shutdown SSx Current Sink Soft-Start Ramp Time SSx Shutdown Discharge Resistance VOUTx Shutdown Discharge Resistance CSSx = 4.7nF ENx = RTN, VOUTx < 300mV ENx = RTN, VOUTx < 300mV 5 5 700 16 3.5 3.5 6.5 6.5 μA μA μsec Ω 16 Ω Analog Inputs/Outputs VOUT1 Input Resistance VOUT2 Input Resistance FBx Input Bias Current EN1 = VDD1 EN2 = VDD2 120 90 -1 +1 kΩ kΩ μA Current Sense ILIMx Source Current ILIMx Comparator Offset Current Limit (Negative) Zero Crossing Detector Threshold ILIMx - GND LXx - GND LXx - GND 80 10 9 -10 60 -7 11 +10 100 +7 μA mV mV mV Power Good PGDx Threshold PGDx Threshold Delay Time (2) 1% Hysteresis Typical, with Respect to Regulation Point -12 5 -9 -15 % μs 5 SC416 Electrical Characteristics (continued) 25°C Parameter PGDx Leakage -40° to 85°C Max Min Max 1 Conditions Min Typ Units μA Fault Protection VDD1 Under-Voltage Lockout VOUTx Under-Voltage Fault VOUTx Under-Voltage Fault Delay (2) VOUTx Over-Voltage Fault VOUTx Over-Voltage Fault Delay (2) Thermal Shutdown (2) Latching, >10°C Hysteresis VDD1 Falling Edge VOUTx Falling Edge 4.0 -30 8 +20 5 160 +17 +23 3.7 -35 4.35 -25 V % clks % μs °C Inputs/Outputs EN1, EN2 Input Low Voltage ENx Input Forced Continuous Mode Operation ENx Input High Voltage VOUTx Disabled ENx = Open (float) VOUTx Enabled, Power Save Enabled R Pull Up to VDDx ENx Input Resistance R Pull Down to RTN FBx Input Low Voltage FBx Input High Voltage Power Good Output Low Voltage RPGDx = 10kΩ to VDDx 1 0.3 VDDx -0.3 0.4 MΩ V V V 1.5 2.0 1.2 V V 3.1 V MΩ Gate Drivers Shoot-Thru Protection Delay(2) DLx Pull-Down Resistance DLx Sink Current (2) DLx Pull-Up Resistance DLx Source Current (2) DHx Pull-Down Resistance DHx or DLx Rising DL Low VDLx = 2.5V DLx High VDLx = 2.5V DHx Low, BSTx - LXx = 5V DHx High, BSTx - LXx = 5V VDHx = 2.5V 30 0.8 3.1 2 1.3 2 4 4 1.6 ns Ω A Ω A Ω DHx Pull-Up Resistance(3) DHx Sink/Source Current (2) 2 1.3 4 Ω A Notes: 1) RTON = 1Meg. 2) Guaranteed by design. 3) Semtech’s SmartDriver™ FET drive first pulls DH high with a pull-up resistance of 10Ω (typical) until LX = 1.5V (typical). At this point, an additional pull-up device is activated, reducing the resistance to 2Ω (typical). This negates the need for an external gate or boost resistor. 6 SC416 Pin Descriptions Pin # 1 2 3 4 5 Pin Name LX1 BST1 VDD1 DL1 EN1 Pin Function Switching (phase) node for VOUT1 Boost capacitor connection for VOUT1 high-side gate drive 5V power input for VOUT1 analog circuits and gate drive outputs. Under-voltage lockout for the 5V supply is sensed on VDD1 only. Gate drive output for the VOUT1 low-side external MOSFET Enable input for VOUT1: ground to disable the VOUT1 switcher, leave open to enable VOUT1 switcher with power save disabled, connect to VDD1 to enable VOUT1 in power save mode Soft-start and ramped shutdown input for VOUT1. For independent startup connect a capacitor to RTN. For tracked startup, connect to a capacitor or to a resistor divider connected to the other output; see the Voltage Tracking section. Connect to the output capacitor of VOUT1 - used for On-Time generation and for VOUT1 regulation when FB1 is connected to VDD1 or RTN Feedback input for VOUT1 - connect to an external resistor divider to adjust VOUT1, or connect to RTN or VDD1 to select internal feedback. Analog return (ground) for both VOUT1 and VOUT2 On-time adjust input - connect a resistor from VIN to TON to program the on-time - the on-time one-shot for VOUT1 is internally set at 20% greater than for VOUT2 to prevent frequency interaction between the two converters Feedback input for VOUT2 - connect to an external resistor divider to adjust VOUT2 or connect to RTN or VDD2 to select internal feedback Connect to the output capacitor of VOUT2 - used for On-Time generation and for VOUT2 regulation when FB2 is connected to VDD2 or RTN Soft-start and ramped shutdown input for VOUT2. For independent startup connect a capacitor to RTN. For tracked startup, connect to a capacitor or to a resistor divider connected to the other output; see the Voltage Tracking section. Enable input for VOUT2 - ground to disable the VOUT2 switcher - leave open to enable VOUT2 switcher with power save disabled - connect to VDD2 to enable VOUT2 in power save mode Gate drive output for the VOUT2 low-side external MOSFET 5V power input for VOUT2 analog circuits and gate drive outputs. VDD2 must connect to the same supply as VDD1. Boost capacitor connection for VOUT2 high-side gate drive Switching (phase) node for VOUT2 Gate drive output for the VOUT2 high-side external MOSFET Current limit input for VOUT2 - connect through a resistor to the drain of the VOUT2 low-side MOSFET Open-drain Power Good output for VOUT2 Open-drain Power Good output for VOUT1 Current limit input for VOUT1 - connect through a resistor to the drain of the VOUT1 low-side MOSFET Gate drive output for the VOUT1 high-side external MOSFET Power ground for VOUT1 and VOUT2 gate drivers and thermal pad for heatsinking 6 SS1 7 8 9 10 VOUT1 FB1 RTN TON 11 12 FB2 VOUT2 13 SS2 14 15 16 17 18 19 20 21 22 23 24 T EN2 DL2 VDD2 BST2 LX2 DH2 ILIM2 PGD2 PGD1 ILIM1 DH1 PAD 7 SC416 Block Diagram VDD1 VDD1 REF FB1 Comparator S Q BST1 DH1 DRV VIN FB1 Select FB1 EN1 SS1 VOUT1 SS1 Control VDD1 TON1 TON1 One-Shot UV- OV Monitor VDD1 Valley ILIM/ ZCD Detect REF LX1 R QB DRV LX1 DL1 PGD1 ILIM1 TON TON Reference VDD2 VDD2 REF FB2 Comparator BST2 DH2 S Q DRV LX2 DL2 DRV VIN FB2 EN2 SS2 VOUT2 FB2 Select SS2 Control TON2 One-Shot UV- OV Monitor TON2 R QB VDD2 PGD2 Valley ILIM/ ZCD Detect ILIM2 LX2 PAD RTN GND SC416 Block Diagram 8 SC416 Applications Information SC416 Synchronous Buck Controller The SC416 is a dual synchronous controller which simplifies the task of designing a dual-output power supply with synchronized or tracking ramps for startup and shutdown. VIN and +5V Bias Supplies The SC416 requires an external +5V bias supply in addition to the VIN supply. If stand-alone capability is required, the +5V supply can be generated with an external linear regulator. Pseudo-Fixed Frequency Constant On-Time PWM Controller The PWM control method for each output is a constanton-time, pseudo-fixed frequency PWM controller, see Figure 1. The ripple voltage seen across the output capacitor’s ESR provides the PWM ramp signal. The on-time is determined by an internal one-shot whose period is proportional to output voltage and inversely proportional to input voltage. A separate one-shot sets the minimum off-time (typically 330ns). The two converters are designed to operate at different frequencies to reduce interaction; side2 frequency is set typically 20% higher than side1. VIN TON VLX CIN VLX L Q2 ESR R1 FB + COUT R2 On-Time One-Shot (TON) Each internal on-time one-shot comparator has two inputs. One input looks at the output voltage via the VOUT pin, while the other input samples the input voltage via the TON pin and converts it to a proportional current. This current charges an internal on-time capacitor. The TON on-time is the time required for this capacitor to charge from zero volts to VOUT, thereby making the on-time directly proportional to output voltage and inversely proportional to input voltage. This implementation results in a fairly constant switching frequency with no clock generator. The nominal frequency is set through an external resistor RTON connected between VIN and the TON pin. To minimize interaction between the two converters, side2 is set to operate at a slightly higher frequency. The general equations for the side1 and side2 on-times are: TON1 = 3.30 × (RTON + 37) × (VOUT/VIN) + 35 TON2 = 2.75 × (RTON + 37) × (VOUT/VIN) + 35 (TON in nsec, RTON in kΩ) Switch-Mode Operation The switch-mode operation is explained below, and is identical for both sides except for the TON timing. The output voltage is sensed at the FB pin and is compared to the internal 750mV reference. (The output voltage can also be sensed at the VOUT pin which uses an internal resistor divider, see VOUT Voltage Selection.) When the sensed voltage drops below 750mV, this triggers a single TON pulse, which is fed to the DH high-side driver. The DH pulse-width follows TON according to the TON equation, and after that time DH drives low to shut off the high-side MOSFET. After DH drives low, the DL output drives high to energize the low-side MOSFET. Once high, DL has a minimum pulse width of typically 330nsec which is the minimum off-time. At the end of the minimum off-time, DL continues to stay high until Q1 VFB FB Threshold 750mV VOUT Figure 1 9 SC416 Applications Information (continued) one of the following occurs: The FB comparator input drops to the 750mV reference, as sensed through the FB pin or the VOUT pin The Zero Cross detector trips, if psave is active The Negative Current Limit detector trips If DL drives low because FB has dropped to 750mV, then another DH on-time is started. This is normal operation at heavy load (fully synchronous operation with either DH or DL high except during transitions). The Zero Cross detector monitors the voltage across the low-side MOSFET during the DL high time and detects when it reaches zero. If DL drives low because of the Zero Cross detector, and psave is active, then both DH and DL will remain low until FB drops to 750mV, at which point the next DH on-time will begin. If a Zero Cross is detected on eight consecutive cycles, then for each subsequent switching cycle DL will shut off when the Zero Cross detector trips; see the Psave Operation section. When this occurs, both DH and DL will stay low until FB drops to 750mV, which will begin the next DH on-time. This is normal operation at light load, see the Psave Operation section. The Negative Current Limit detector trips when the drain voltage at the low-side MOSFET reaches +80mV, indicating that a large negative current flows through the inductor from VOUT. When this occurs, DL drives low. Both DH and DL will then stay low until FB drops to 750mV, which will begin the next DH on-time. Tripping Negative Current Limit is rare. To help reduce noise interaction between sides, the rising edge of each DH driver is inhibited momentarily if the other side is switching. For example, if side1 is performing a DH or DL transition (up or down), then side2’s DH driver is held off for roughly 30nsec to allow side1 to finish switching. VOUT Voltage Selection Output voltage is regulated by comparing VOUT as seen through a resistor divider to the internal 750mV reference, see Figure 2. Each output can be adjusted to a voltage between 0.75 – 5.25V. The output voltage is set by the equation: VOUT = 0.75 • (1 + R1/R2) VOUT R1 R2 To FB pin Figure 2 Note: the parallel resistance of R1 and R2 should not be less than 2kΩ. Using a smaller resistance can cause the IC to default to the internal preset output voltages shown below. There are fixed output voltages accessible through each FB pin. If the FB pin is connected to either RTN or +5V, then the IC will ignore the FB pin and instead regulate the output voltage using the VOUT pin which is connected to internal resistors. The voltage selections available are as shown: FB Internal Voltage Selection FBx = RTN FBx = +5V Side1 1.5V 1.8V Side2 1.05V 1.25V Note that each FB input operates independently. 10 SC416 Applications Information (continued) Enable/Psave Inputs Each converter has a separate Enable pin. Each EN input operates as follows: EN = GND. This turns the converter off. EN = open (float). This turns the converter on with psave mode disabled (continuous conduction mode). In this case, the EN pin will float to approximately 2V due to an internal 1.5Meg/1Meg resistor divider from +5V to ground. EN = high (3.1V min). This turns the converter on with psave mode enabled. At light loads, the converter will operate in psave mode. immediately exits Psave. Once Psave is exited, it requires 8 switching cycles at light load to re-enter psave. Since the controller counts zero crossings, the converter can sink current as long as the current does not cross zero on eight consecutive cycles. This allows the output voltage to recover quickly in response to negative load steps. When operating in Psave mode at light loads, the LX waveform will not have the typical square wave shape seen when operating in continuous conduction mode. Shortly after DL drives low, and both MOSFETs are off, the LX voltage will show ringing. This ringing is caused by the LC circuit formed by the inductor and device capacitance of the MOSFETs and low-side diode. When the low-side MOSFET turns off the inductor current falls toward zero. When it reaches zero, the inductance and MOSFET capacitances will tend to ring freely. This is normal Psave operation as shown in Figure 3. VIN Note that the two EN pins are separate, so each output can be disabled or operated with or without psave independently. For tracking operation during startup, the two EN pins are typically tied together; see the Voltage Tracking section. If both EN1 and EN2 are grounded, the device is placed into the lowest-power state, drawing typically 7μA from the +5V supply. Psave Operation Each output provides automatic psave operation at light loads if the ENx pin is set high. The internal Zero-Cross comparator looks for inductor current (via the voltage across the lower MOSFET) to fall to zero on eight consecutive switching cycles. Once observed, the controller enters psave mode and turns off the low-side MOSFET on each subsequent cycle when the current crosses zero. To add hysteresis, the on-time is also increased by 25% in Psave, for that converter only; it does not affect the other converter’s on-time. The efficiency improvement at light loads more than offsets the disadvantage of slightly higher output ripple. If the inductor current does not cross zero on any switching cycle, the controller CQ1 DH Q1 L VOUT DL Q2 COUT CQ2 DL VIN LX VOUT Typical ringing at LX Figure 3 11 SC416 Applications Information (continued) Smart Psave Protection In some applications, circuits connected to VOUT can leak current from a higher voltage and thereby cause VOUT to slowly rise and reach the OVP threshold, causing a hard shutdown; the SC416 uses Smart Psave to prevent this. When the output voltage exceeds 8% above nominal (810mV at FB), that converter then exits Psave (if already active), and DL drives high to energize the low-side MOSFET. This will draw current from VOUT via the inductor causing VOUT to fall. When FB drops to the 750mV trip point, a normal TON switching cycle begins. This method cycles energy from VOUT back to VIN and prevents a hard OVP shutdown, and also minimizes operating power by avoiding continuous conduction-mode operation. If a light load is present, DH/DL switching continues for 8 consecutive cycles and then the IC returns to Psave mode to reduce operating power. Current Limit For current limiting, the RDSON of the lower MOSFET can be used as a current sensing element, or a sense resistor at the lower MOSFET source can be used if greater accuracy is needed. RDSON sensing is more efficient and less expensive. In both cases, the RILIM resistor sets the overcurrent threshold. RILIM connects from the ILIM pin to either the lower MOSFET drain (for RDSON sensing) or the high side of the current-sense resistor. RILIM connects to a 10μA current source from the ILIM pin which turns on when the low-side MOSFET is on (DL is high). If the voltage drop across the sense resistor or low-side MOSFET exceeds the voltage across RILIM, then the voltage at the ILIM pin will be negative or below GND, and current limit will activate. The high-side MOSFET is not allowed to turn on until the voltage drop across the sense resistor or MOSFET falls below the voltage across the RILIM resistor (ILIM pin reaches GND). If the overload at the output continues, the DH pulses will get farther apart, and the output voltage will fall. Eventually the output will fall enough to cause FB to drop to 525mV, activating the under-voltage protection and shutting down the converter. The current sensing scheme actually regulates the inductor valley current, (see Figure 4). This means that if the current limit is set to 10A, the peak current through the inductor would be 10A plus the peak ripple current, and the average current through the inductor would be 10A plus ½ the peak-to-peak ripple current. INDUCTOR CURRENT I PEAK I LOAD I LIMIT TIME Valley Current Limit Figure 4 The RDSON sensing circuit is shown in Figure 5 with RILIM = R1 and RDSON of Q2. +5V VIN D1 Q1 + CIN VOUT L BST DH LX ILIM VDD DL PAD C2 R1 Q2 D2 + C3 Figure 5 The resistor sensing circuit with RILIM = R1 and RSENSE = R4 is shown in Figure 6. 12 SC416 Applications Information (continued) Output Over-Voltage Protection (OVP) When FB exceeds 20% of nominal (900mV), DL latches high and the low-side MOSFET is turned on. DL stays high and the output stays off until the EN/PSV input is toggled or VDD is recycled. There is a 5μs delay built into the OVP detector to prevent false transitions. PGD is also held low after an OVP. Output Under-Voltage Protection (UVP) When FB falls 30% below nominal (to 525mV) for eight consecutive clock cycles, the output is shut off; the DL/ DH drives are pulled low to tri-state the MOSFETs, and the converter stays off until its EN/PSV input is toggled or the +5V supply at VDD1 supply is recycled. The other output remains on during a UVP event. POR and UVLO Under-voltage lockout circuitry (UVLO) inhibits switching and tri-states all DH/DL drivers until the +5V supply at VDD1 rises above 4.4V. An internal power-on reset (POR) occurs when VDD1 exceeds 4.4V, which resets the fault latches and quickly discharges the soft-start capacitors to prepare the PWM for startup switching. At this time the SC416 will exit UVLO and begin the startup cycle. Startup Sequence The startup sequence for each output relies on an external ramp at the SS pin. During startup, the FB comparator uses the SS ramp voltage as the reference until SS reaches 750mV, at which point the FB comparator switches over to the internal 750mV reference. The external ramp is typically created by connecting a capacitor to the SS pin, which will be charged by a constant current. Or, the SS pin can be connected to an external ramp provided by the other output’s ramp-up as seen through a resistor divider. See the Voltage Tracking section. Before starting, with EN low, the SS pin is internally tied to GND through 4kΩ. When EN is released, SS is briefly pulled to GND through 16 ohms to discharge the SS capacitance. Then the resistances are removed, startup begins, and a 5μA source current flows out of the SS pin. +5V VIN D1 Q1 C2 BST DH LX ILIM VDDP DL PAD L1 + CIN VOUT Q2 D2 + C3 R1 R4 Figure 6 The following over-current equation can be used for both RDSON or resistive sensing. For RDSON sensing, the MOSFET RDSON rating is used for the value of RSENSE. ILOC(Valley) = 10μA × RILIM / RSENSE Power Good Output Each output provides a power good (PGD) output, which is an open-drain output requiring a pull-up resistor. When the output voltage as sensed at FB is -9% from the 750mV reference (682mV), PGD is pulled low. It is held low until the output voltage returns above -9% of nominal; the falling edge of VOUT does not latch PGD. PGD is held low during start-up and will not be allowed to transition high until soft-start is completed, when SS reaches 750mV. There is a 5μsec delay built into the PGD circuit to prevent false transitions. PGD also transitions low if the FB pin exceeds +20% of nominal (900mV), which is also the over-voltage shutdown point. 13 SC416 Applications Information (continued) If an external capacitor is connected to SS, with no other components, then the 5uA current into the capacitor creates a linear voltage ramp. The internal FB comparator tracks this SS ramp, which forces VOUT to track the SS ramp. The time in msec needed for the SS ramp to reach the 750mV reference is: TSTART = Css × 150 (TSTART in μsec, Css in nF) At the end of this time, the SS pin has reached 750mV and the output voltage is at its nominal value. The FB comparator then switches over to the internal 750mV REF, and the SS pin is thereafter ignored. The 5μA current source remains on, so the Css capacitor continues to charge up to +5V. The SS pin can be connected to a capacitor to provide a programmable ramp, or connected to a resistor divider from the other output to provide a synchronized startup, see the Voltage Tracking section. The startup waveforms when using a capacitor for the voltage ramp are shown in Figure 7. TSTART Shutdown Sequence Each output has a controlled shutdown. The ramp-down sequence is optimized for use with an external SS capacitor. When the EN pin is set low, psave mode is disabled and the shutdown sequence begins. The SS pin stops sourcing 5uA, and an internal 4K pulldown resistor is enabled along with a 5uA current sink. The 4K resistor helps discharge of the SS capacitor (if used), which was previously charged to +5V by the 5uA current source. The SS capacitor discharges quickly; when the SS voltage reaches 810mV, the 4K resistor is released, leaving the 5μA current sink on (current into the SS pin), which discharges the SS capacitor from 810mV to 750mV at a slower rate. When the SS voltage reaches 750mV, the ramp-down at VOUT begins. With SS at 750mV, the FB comparator ignores the internal 750mV reference and instead uses the SS ramp-down to regulate VOUT. As the SS pin ramps down from 750mV due to the current sink, VOUT follows the SS pin in a linear ramp-down. Note that psave is disabled during the shutdown cycle to allow VOUT to actively track the SS ramp-down, even with no load. The ramp-down continues until VOUT falls to 300mV as sensed at the VOUT pin, not the FB pin. At this point the switching is disabled, the DH and DL drivers are set low (both MOSFETs off), and an internal 16 ohm pulldown connects to the VOUT pin to finish soft-discharge of the output. 5V VOUT 750mV SS SS linear ramp: 5uA current source into CSS EN EN SS CSS SC416 Figure 7 14 SC416 Applications Information (continued) The general shutdown waveforms are shown in Figure 8. TSD_DELAY TSD_RAMP Voltage Tracking The SC416 provides voltage tracking during both startup and shutdown. The use of the SS pins determines the type of tracking. The two most common tracking schemes are proportional and coincident tracking. Proportional Tracking (FB tracking) Proportional tracking causes the output waveforms to have the same startup and shutdown ramp time as shown in Figure 9. TSD_RAMP TSTART VOUT1 VOUT2 VOUT VOUT linear down-ramp Soft-discharge into 16 ohms 300mV: switching stops SS Rapid CSS discharge into 4Kohm 810mV: SS linear down-ramp begins +5V Linear down-ramp EN EN SS CSS SC416 750mV: VOUT down-ramp begins 300mV +5V Rapid discharge of SS capacitor Figure 8 SS1 SS2 750mV Note that due to the discharging of CSS from +5V to 750mV, there is a small delay between EN going low and VOUT beginning to ramp down. The delay is given by the equation: TSD_DELAY = Css × 19 (TSD_DELAY in μsec, Css in nF) EN1 EN2 EN1 EN2 SC416 SS1 SS2 CSS Figure 9 For proportional tracking, the two SS pins are tied together to share the same voltage ramp. The two EN pins must also be tied together. Since the SS pins are tied together, the startup and shutdown ramps for each side will track the same signal. Note that during startup and shutdown, the FB pins for each side will track the same SS signal; this method can also be called FB tracking. Startup for proportional tracking (FB tracking) is as follows. When both EN pins are set high simultaneously, each SS pin sources 5uA. By connecting the SS pins together, the startup ramp rates for both FB1 and FB2 are forced to be the same. Since the total current source is After this delay, the time for the output to ramp down to the switch-off point of 300mV is given by: TSD_RAMP = Css × 150 • (1 – 300mV/VOUT) (TSD_ RAMP in μsec, Css in nF) Once the output reaches 300mV, the shutdown sequence is complete. DH and DH stop switching, the internal 16 ohm discharge pulldown at VOUT is energized, and that side will go to the lowest-power state. 15 SC416 Applications Information (continued) 10μA instead of 5μA, the CSS capacitor must be twice as large as in the independent case. It is acceptable to use only one SS capacitor for both SS pins, and the startup time follows the equation: TSTART = Css × 75 (TSTART in μsec, Css in nF) Once the SS pins reach 750mV, each FB comparator switches over to the internal 750mV reference, and the SS pin is ignored. CSS will continue to charge to +5V, just as in the independent case. Shutdown for proportional tracking is similar to the independent case. Both EN pins are simultaneously pulled to GND to start the shutdown sequence. When the EN pins are set low, the two SS 5μA current sources are disabled, and the internal 4k pull-down of each SS pin is energized. This begins a quick discharge of CSS from the initial +5V starting point. When CSS discharges down to 810mV, each 4k resistor is removed and each SS 5μA current sink is energized, to cause a more gradual ramp down to 750mV. When SS reaches 750mV, each FB comparator ignores the internal 750mV reference and uses the falling SS ramp as the reference. This causes both FB pins to fall at a constant linear rate. This in turn causes both VOUTs to fall linearly. This continues for each VOUT until it reaches 300mV, at which point all switching for that output will stop. Note that the output ramp-down is controlled only until the point where the output voltage (as sensed at the VOUT pin, not the FB pin) reaches 300mV, and then switching stops. Also note that the two output voltages may not reach 300mV at the same time. With proportional tracking, the lower voltage will reach 300mV first; at that point, the lower voltage will enter soft-discharge into 16 ohms, but the other output will continue on the SS ramp-down until it also reaches 300mV. For the majority of applications this is acceptable since 300mV is typically too small to cause a voltage-differential problem for the load. Coincident Tracking (VOUT tracking) Coincident tracking has the lower output actively matching the higher output during startup and shutdown. During startup, the two voltages track identically until the lower VOUT reaches its nominal point and then remains at that point. The higher voltage continues on a linear ramp until reaching its final value. During shutdown, the higher voltage initially starts to fall while the lower voltage stays in regulation. When the higher voltage has dropped to the same level as the lower voltage, the lower voltage joins the ramp-down, and both voltages fall at the same rate. When the outputs reach 300mV, switching stops and the outputs will softdischarge into their VOUT pins. The waveforms and schematic for coincident tracking are shown in Figure 10. TSTART TSD_RAMP VOUT1 VOUT2 300mV +5V Rapid discharge of SS capacitor SS1 750mV EN1 EN2 EN1 EN2 CSS SS1 VOUT1 R3 R4 SS2 VOUT2 R1 R2 SC416 FB2 Figure 10 Note that both EN pins are tied together. For coincident tracking, the higher output voltage must act as the control, and the lower output must be the follower. Either side can act as controller or follower. In this case, VOUT1 is the higher output voltage. When the EN pins are set 16 SC416 Applications Information (continued) high, both outputs start switching. SS1 is connected to a capacitor and therefore VOUT1 will follow the normal (independent) startup and shutdown operation. SS2 does not have the typical capacitor connection; instead SS2 connects to VOUT1 through resistor divider R3/R4, which forces the SS2 voltage to be proportional to VOUT1. As VOUT1 rises during startup, SS2 also rises, and the FB2 comparator will force VOUT2 to track SS2 until SS2 reaches 750mV. Note that FB2 sets the output voltage for VOUT2 through R1/R2. By setting the same ratio for R3/R4 as R1/R2, VOUT2 is forced to track VOUT1. This tracking continues until FB2 reaches 750mV, at which point VOUT2 ignores the SS2 ramp and switches over to the internal 750mV reference. SS2 will continue to rise above 750mV since it is connected to VOUT2 which continues to rise to its nominal value. Note that during this time the SS2 pin will source 5uA, as if it were charging a capacitor. This current will flow through into R3/R4 and create an offset voltage at SS2. The full equation for SS2 voltage is: VSS2 = VOUT1 × (R4 / (R3 + R4) + 5μA × (R3||R4) Since the ratio of R5/R6 is dictated by the output voltage, the way to minimize the offset is to reduce R3 and R4. It is recommended to set the R3/R4 parallel combination to 1kΩ max, which limits the SS2 offset to 5mV. For shutdown, both EN pins are pulled low to start the shutdown sequence. The higher output falls first (VOUT1 in this case) as described in the independent case. The SS2 pin also falls as VOUT1 falls. When SS2 has reached 750mV, the ramp-down of VOUT2 begins. The FB2 comparator switches over to the SS2 pin and tracks SS2 as it ramps down. SS2’s ramp-down is in turn controlled by the ramp-down of VOUT1. Note that during the initial time when both EN pins pulled low, the internal 4kΩ pulldown on SS2 is energized, along with the 5uA current sink. The connecting of the 4k pulldown affects the SS2 voltage, causing it to be significantly lower than predicted by the R3/R4 divider. If the SS2 voltage suddenly shoots below 750mV, this will cause the FB2 comparator to ignore the internal 750mV reference and switch over to the SS2 voltage. Also, the 4k resistor is released, which causes the SS2 voltage to go back above 750mV, since SS2 will now track VOUT1 through R3/R4. This can create an unwanted overshoot on VOUT2. This is easily prevented by placing a 10nF capacitor across R4, to prevent sudden changing of the SS2 voltage. Modified Coincident Tracking (VOUT Proportional Tracking) A modification of coincident tracking is to set R3/R4 (SS2 resistor divider) to a ratio other than R1/R2 (FB1 resistor divider). VOUT2 will still track SS2 during startup and shutdown, but VOUT2 can be made to ramp either quicker or slower than VOUT1, based on the R3/R4 ratio, see Figure 11. The term for the SS2 offset is (5μA × R3||R4). When tracking VOUT1 during startup, SS2 will be slightly higher than the ideal value due to this offset term. This offset at SS2 is tracked by FB2. Because of the FB2 resistor divider, this causes a consequently higher offset at VOUT2. The net effect is that VOUT2 will be slightly higher than VOUT1 during startup tracking. The net offset is given by the following equation, where R5/R6 are the top and bottom resistor values used for the FB2 resistor divider: VOUT1 – VOUT2 = 5μA × (R3||R4) × (1 + R5/R6) 17 SC416 Applications Information (continued) TSTART TSD_RAMP VOUT1 VOUT2 R3/R4 > R1/R2 R3/R4 < R1/R2 +5V Rapid discharge of SS capacitor 300mV 750mV SS1 SmartDriveTM Each side uses Semtech’s proprietary SmartDrive to reduce switching noise. The DH drivers will turn on the high-side MOSFET at a lower rate initially, allowing a softer, smooth turn-off of the low-side diode. Once the diode is off, the SmartDrive circuit automatically drives the highside MOSFET on at a rapid rate. This technique reduces switching less while maintaining high efficiency, and also avoids the need for snubbers or series resistors in the gate drive. EN1 EN2 EN1 EN2 CSS SS1 VOUT1 R3 R4 SS2 VOUT2 R1 R2 SC416 FB2 Figure 11 Note that R3/R4 must never exceed the FB resistor ratio for VOUT1. If this were the case, SS2 cannot exceed 750mV, and the FB2 comparator would never switch over to the internal 750mV reference: VOUT2 would not complete its startup cycle and would not reach the nominal regulation point. To prevent this it is recommended to set R3/R4 such that SS2 is greater than 850mV when VOUT1 is at the nominal point. MOSFET Gate Drivers The DH and DL drivers are optimized to drive moderate high-side and larger low-side power MOSFETs. An adaptive dead-time circuit monitors the DL output and prevents the high-side MOSFET from turning on until DL is fully off; another circuit monitors the DH output and prevents the low-side MOSFET from turning on until DH is fully off. Note: be sure there is low resistance and low inductance between the DH and DL outputs to the gate of each MOSFET. Figure 12 Design Procedure Prior to designing a switch mode supply, the input voltage, load current, and switching frequency must be specified. The maximum input voltage (VINMAX) is determined by the highest AC adaptor voltage, and the minimum input voltage (VINMIN) is determined by the lowest supply voltage after accounting for voltage drops due to connectors, fuses, and switches. In general, four parameters are needed to define the design: 1. 2. 3. 4. Nominal output voltage (VOUT) Static or DC output tolerance Transient response Maximum load current (IOUT) 18 SC416 Applications Information (continued) There are two values of load current to consider: continuous load current and peak load current. Continuous load current is concerned with thermal stresses which drive the selection of input capacitors, MOSFETs and diodes. Peak load current determines instantaneous component stresses and filtering requirements such as inductor saturation, output capacitors and design of the current limit circuit. Design example: Side1 will be used. VIN = 10V min, 20V max VOUT1 = 1.8V +/- 4% Load = 10A maximum Note that side2 will run typically 20% faster than side1, in this case 320kHz. During the DH on-time, voltage across the inductor is (VIN - VOUT). To determine the inductance, the ripple current must be defined. Smaller ripple current will give smaller output ripple and but will lead to larger inductors. The ripple current will also set the boundary for Psave operation. The switcher will typically enter Psave operation when the load current decreases to ½ of the ripple current; (i.e. if ripple current is 4A then Psave operation will typically start for loads less than 2A. If ripple current is set at 40% of maximum load current, then Psave will occur for loads less than 20% of maximum current). The equation for inductance is: L = (VIN - VOUT) × TON / IRIPPLE Inductor Selection Low inductor values result in smaller size but create higher ripple current. Higher inductor values will reduce the ripple current but are larger and more costly. Because wire resistance varies widely for different inductors and because magnetic core losses vary widely with operating conditions, it is often difficult to choose which inductor will optimize efficiency. The general rule is that higher inductor values have better efficiency at light loads due to lower core losses and lower peak currents, but at high load the smaller inductors are better because of lower resistance. The inductor selection is generally based on the ripple current which is typically set between 20% to 50% of the maximum load current. Cost, size, output ripple and efficiency all play a part in the selection process. The first step is to select the switching frequency. In this case VOUT1 will be used at a nominal 270kHz. For 15V input and 1.8V output, the typical on-time should be: TONtyp = VOUT/VIN/Frequency TONtyp = 444nsec. Use the maximum value for VIN, and for TON use the value associated with maximum VIN, and that side’s TON using the RTON value selected. For selecting the inductor, we start with the highest VOUT setting and a maximum ripple current of 4A. TON1 = 343 nsec at 20VIN, 1.8VOUT L = (20V - 1.8V) × 343 nsec / 4A = 1.56μH We will use 1.5μH which will slightly increase the maximum IRIPPLE to 4.2A. Note: the inductor must be rated for the maximum DC load current plus ½ of the ripple current. The minimum ripple current is also calculated. This occurs when VIN is at the minimum value of 10V. TONVINMIN = 3.3 × (RTON + 37) × (VOUT/VIN) + 35 TONVINMIN = 651nsec IRIPPLE = (VIN - VOUT) × TON / L IRIPPLE_VINMIN = (10 – 1.8) × 651nsec / 1.5μH = 3.55A The timing resistor RTON is selected to provide TONtyp: RTON = (TONtyp – 35) × (VIN / (3.3 × VOUT) – 37 RTON = 976K. We will use RTON = 1Meg. 19 SC416 Applications Information (continued) Capacitor Selection The output capacitors are chosen based on required ESR and capacitance. The ESR requirement is driven by the output ripple requirement and the DC tolerance. The output voltage has a DC value that is equal to the valley of the output ripple, plus ½ of the peak-to-peak ripple. Changing the ripple voltage will lead to a change in DC output voltage. The design goal is +/-4% output regulation. The internal 750mV reference tolerance is 1%, and assuming 1% tolerance for the FB resistor divider, this allows 2% tolerance due to VOUT ripple. Since this 2% error comes from ½ of the ripple voltage, the allowable ripple is 4%, or 72mV for a 1.8V output. Although this is acceptable from a regulation standpoint, 72mV ripple is high for a 1.8V output and therefore more realistic ripple value of 36mV will be used (2% of VOUT). The maximum ripple current of 4.2A creates a ripple voltage across the ESR. The maximum ESR value allowed would create 36mV ripple: ESRMAX = VRIPPLE/IRIPPLEMAX = 36mV / 4.2A ESRMAX = 8.6 mΩ With a peak voltage VPEAK of 1.98V (180mV or 10% rise above 1.8V upon load release), the required capacitance is: COUTMIN = 1.5μH × (10 + 1/2 × 4.2)2 ___________________ (1.982 - 1.82) COUTMIN = 323μF The previous requirements (323μF, 6.4mΩ) will be met using a single 330μF 6mΩ device. Note that output voltage ripple is often higher than expected due to the ESL (inductance) of the capacitor. See the FB/VOUT Ripple section. If the load release is relatively slow, the output capacitance can be reduced. At heavy loads during normal switching, when the FB pin is above the 750mV reference, the DL output is high and the low-side MOSFET is on. During this time, the voltage across the inductor is approximately -VOUT. This causes a down-slope or falling di/dt in the inductor. If the load di/dt is not much faster than the di/dt in the inductor, then the inductor current can track the changing load current, and there will be relatively less overshoot from a load release. The following can be used to calculate the needed capacitance for a given dILOAD/dt. Peak inductor current: ILPEAK = ILOADMAX + 1/2 × IRIPPLEMAX ILPEAK = 10 + 1/2 × 4.2 = 12.1A Rate of change of Load current = dILOAD/dt IMAX = maximum DC load current = 10A While the ESR is chosen to meet ripple requirements, the output capacitance (μF) is typically chosen based on transient requirements. A worst-case load release, from maximum load to no load at the exact moment when inductor current is at the peak, defines the required capacitance. If the load release is instantaneous (load changes from maximum to zero in a very small time), the output capacitor must absorb all the inductor’s stored energy. This will cause a peak voltage on the capacitor according to the equation: COUTMIN = L × (IOUT + 1/2 × IRIPPLEMAX) ___________________ (VPEAK2 - VOUT2) 2 COUT = ILPEAK × (L ×ILPEAK / VOUT - IMAX/(dILOAD /dt)) ___________________________________ 2 × (VPEAK - VOUT) Example: Load dI/dt = 2.5A/μsec 20 SC416 Applications Information (continued) This would cause the output current to move from 10A to zero in 4μsec. COUT = 12.1 × (1.5μH × 12.1 / 1.8 - 10 / (2.5A/1μsec)) ____________________________________ 2 × (1.98 - 1.8) COUT = 204μF DH VIN L mimics the ESR ramp. This virtual ESR ramp is created by integrating the voltage across the inductor, and coupling the signal into the FB pin as shown in Figure 13. Note that 204μF is less than the 323uF needed to meet the harder (instantaneous) transient load release. DL RL CC CL Stability Considerations Unstable operation shows up in two related but distinctly different ways: fast-feedback loop instability due to insufficient ESR and double-pulsing. Loop instability can cause oscillations at the output as a response to line or load transients. These oscillations can trip the over-voltage protection latch or cause the output voltage to fall below the tolerance limit. The best way for checking stability is to apply a zero-to-full load transient and observe the output voltage ripple envelope for overshoot and ringing. Over one cycle of ringing after the initial step is a sign that the ESR should be increased. SC416 ESR Requirements The on-time control used in the SC416 regulates the valley of the output ripple voltage. This ripple voltage consists of a term generated by the ESR of the output capacitor and a term based on the capacitance charging and discharging during the switching cycle. A minimum ESR is required to generate the required ripple voltage for regulation. For most applications the minimum ESR ripple voltage is dominated by PCB layout and the properties of the output capacitors, typically SP or POSCAP devices. For stability the ESR zero of the output capacitor should be lower than one-third the switching frequency. The formula for minimum ESR is: ESRMIN = 3 / (2 × π × COUT × FREQ) R1 COUT R2 To FB Figure 13 Double-pulsing Double-pulsing occurs because the ripple waveform seen at the FB pin is either too small, or because the FB and VOUT ripple waveforms are very noisy and prone to cause premature triggering of the FB comparator. Both are discussed below. Increasing FB Ripple If the ripple waveform at FB is too small, the FB waveform will be susceptible to switching noise. Note that under normal conditions the FB voltage is within 10-20mV of the 750mV trip point. Noise can couple into the FB point from either side1 or side1, or from an external circuit. This causes the FB comparator to trigger too quickly after the 330ns minimum off-time has expired. Double-pulsing will result in higher ripple voltage at the output but in most cases is harmless. A way to remedy this is to couple more ripple into FB from VOUT. Note that the feedback resistor divider attenuates the FB ripple. This can be compensated by placing a small capacitor in parallel with the top resistor, which effectively increases the ripple that appears at FB. For applications using ceramic output capacitors, the ESR is generally too small to meet the above criteria. In these cases it is possible to create a ripple voltage ramp that 21 SC416 Applications Information (continued) The schematic with added capacitor C is shown below. Inductor Ripple Current VOUT ripple from ESR FB ripple 750 mV VOUT VOUT C R1 COUT R2 To FB LX R1 To FB LOAD ESR COUT Figure 14 R2 This capacitor should be left out until confirmation that double-pulsing exists. It is best to leave a spot on the PCB in case it is needed. Figure15 In many applications, the output capacitor also has some series inductance (ESL), and this can have a large effect on ripple. The ripple current creates voltage across the ESL; this is a square wave similar to the LX waveform. The result is shown below. Note the fast rising and falling edges created by the ESL. FB/VOUT Ripple Because the constant on-time control method triggers a DH pulse whenever the FB waveform reaches the 750mV trip point, it is important that the VOUT and FB ripple waveforms are well shaped. This waveform will depend on the output capacitors. The idealized case for the output filter has an inductor and a capacitor COUT with series ESR. The voltage ripple due to charging and discharging COUT is typically much smaller than the ripple voltage due to ESR, so the overall ripple waveform is generally determined by ESR only. The result is a well-defined sawtooth, see Figure 15. 22 SC416 Applications Information (continued) Ripple Current LX VOUT ripple from ESR VOUT ripple from ESL VOUT R1 ESL ESR R2 COUT LOAD To FB Total VOUT ripple FB ripple VOUT R1 To FB R2 COUT ESL ESR CB 750 mV LX LOAD Figure 17 Figure 16 Note also that the effect of ESL becomes increasingly more noticeable with lower VOUT and higher VIN. The square wave due to ESL is basically the same square wave seen at the LX node, scaled down by the ratio of the output inductor and the ESL. As VOUT gets lower, the output inductor gets smaller, and this directly increases the ESL square wave. Moreover, lower output voltages have tighter ripple requirements, so there is generally less room available for pk-pk ripple. In addition to the ESL, most applications also have a small capacitor in parallel with COUT; this is typically a small ceramic capacitor intended to absorb high frequency noise not filtered by the output capacitor, as shown by CB in Figure 17. This capacitor CB can have a large effect on the ripple waveform. The switch transitions are fast, typically 1030nsec. At this high speed, the output capacitor (COUT) impedance is dominated by ESL, which is in parallel with CB. The effective circuit is a parallel L-C filter. The choice of CB can have significant effect on the ripple waveform. This parallel LC circuit can lead to ringing at VOUT. Since the FB waveform goes below the 750mV threshold soon after the DH pulse is finished, there is the potential for double-pulsing; the ringing can cause the FB waveform to go below the trip point too early. 23 SC416 Applications Information (continued) Figure 18 shows examples of this. A second solution is to add a small RC filter in series with the FB resistors, shown by RF/CF in Figure 20. This RC filter is intended to remove the high-frequency noise but still allow the ripple to reach the FB pin. Recommended values are 10 ohms and 10nF. VOUT RF R1 FB trip point False trigger on FB Actual ripple with CBP Idealized ESL ripple CF R2 VOUT ripple To FB Idealized ESL ripple FB ripple FB trip point False trigger on FB Figure 20 Figure 18 There are two ways to deal with this issue. One is to use a larger ceramic capacitor, typically 2.2–10μF, which significantly smooths the ripple waveform as shown in Figure 19. Dropout Performance The VOUT adjust range for continuous-conduction operation is limited by the fixed 330nsec (typical) Minimum Off-time One-shot. When working with low input voltages, the duty-factor limit must be calculated using worstcase values for on and off times. The IC duty-factor limitation is given by: DUTY = TONMIN / (TONMIN + TOFFMAX) Large CB (~2.2 – 10 uF) Actual ripple with CBP Ripple due to ESL Be sure to include inductor resistance and MOSFET onstate voltage drops when performing worst-case dropout duty-factor calculations. SC416 System DC Accuracy (VOUT Controller) Three factors affect VOUT accuracy: the trip point of the FB error comparator, the switching frequency variation with line and load, and the external resistor tolerance. The error comparator is trimmed to trip when the FB pin is 750mV, +/-1% over the range of 0 to 85°C. FB trip point Figure 19 24 SC416 Applications Information (continued) The on-time pulse is programmed using the RTON resistor to give a desired frequency. However, some frequency variation with line and load is expected. This variation changes the output ripple voltage. Because constant ontime converters regulate to the valley of the output ripple, ½ of the output ripple appears as a DC regulation error. For example, If the output ripple is 50mV with VIN = 6 volts, then the measured DC output will be 25mV above the comparator trip point. If the ripple increases to 80mV with VIN = 25 volts, then the measured DC output will be 40mV above the comparator trip. The best way to minimize this effect is to minimize the output ripple. To compensate for valley regulation it is often desirable to use passive droop. Take the feedback directly from the output side of the inductor, placing a small amount of trace resistance between the inductor and output capacitor. This trace resistance should be optimized so that at full load the output droops to near the lower regulation limit. Passive droop minimizes the required output capacitance because the voltage excursions due to load steps are reduced. The use of 1% feedback resistors contributes typically 1% error. If tighter DC accuracy is required use 0.1% resistors. Note that for the internal preset voltages, the FB accuracy of 1.25% for 0 to 85°C already includes the tolerance of the internal resistors. The output inductor value may change with current. This will change the output ripple and thus the DC output voltage. The output ESR also affects the ripple and thus the DC output voltage. Switching Frequency Variation The switching frequency will vary somewhat due to line and load conditions. The line variations are a result of a fixed offset in the on-time one-shot, as well as unavoidable delays in the external MOSFET switching. As input voltage increases, these factors make the actual DH ontime slightly longer than the idealized on-time. The net effect is that frequency tends to fall slightly with increasing input voltage. The frequency variation with load is due to losses in the power train from IR drop and switching losses. For a conventional PWM constant-frequency topology, as load increases the duty cycle also increases slightly to compensate for IR and switching losses in the MOSFETs and inductor. A constant on-time topology must also overcome the same losses by increasing the effective duty cycle (more time is spent drawing energy from VIN as losses increase). Since the on-time is constant for a given VOUT/VIN combination, the way to increase duty cycle is to gradually shorten the off-time. The net effect is that switching frequency increases slightly with increasing load. Layout Guidelines As with any switch-mode converter, and especially a dual-channel converter, a good pcb layout is essential for optimum performance. The following guidelines should be used for PCB layout. Placement Note that the pins on the IC are arranged in four groups, i.e. Side1 Power, Side2 Power, Side1 Analog, and Side2 Analog, as shown in Figure 21. Side 1 Power/Gate Drive LX1 BST1 VDD1 1 PGD1 PGD2 ILIM1 ILIM2 24 19 18 Side 2 Power/Gate Drive LX2 BST2 VDD2 DH1 SC416 DL1 EN1 SS1 6 7 12 GND (PAD) 13 DL2 EN2 SS2 RTN VOUT1 TON FB1 FB2 VOUT2 DH2 Side 1 Analog Side 2 Analog Figure 21 25 SC416 Applications Information (continued) For placement, power devices for side1 should be grouped together near the gate drive pins for side1 (pins 23-24 and 1-8). Power devices for side2 should be grouped together near the gate-drive pins for side2 (pins 15-20). The feedback and VOUT sense components should be located near the FBx/VOUTx pins. This includes the feedback resistors and capacitors if used. Ground Connections When doing placement, be aware that there are four grounds to consider on the pcb. ‘ 1. 2. 3. 4. Power ground for Side1 Power ground for Side2 Analog ground for Side1 Analog ground for Side2 drive to the low-side MOSFETs. The DL gate-drive current peaks can be 2 amps or more, with fast switching. As such the ground connection between the low-side MOSFETs and the ground PAD should be as short and wide as practical. Note that the ground PAD, which is the return path for the high-noise DL drive current, is not accessible on the top layer of the pcb, due to the other pins. The ground PAD connection to the MOSFETs must therefore be done on an inner or bottom layer. For this reason, it is best to place the low-side MOSFET on the opposite side of the pcb, to allow a wide and direct connection to the ground PAD on the bottom layer. Otherwise an inner layer must be used for the ground PAD connection; if needed, this should be done with many vias to minimize the high-frequency impedance. This applies to both side1 and side2 MOSFETs. The remaining power devices should then be placed with their ground pins near each other, and near the IC. That is, the ground connections between the IC, the lowside MOSFET, the low-side diode (if used), the input capacitors, and the output capacitor, should be short. The other non-ground power connections (from input cap to high-side MOSFET, from MOSFETs to inductor, and from inductor to output capacitor) should be short and wide as well, to minimize the loop length and area. Use short, wide traces from the DL/DH pins to the MOSFETs to reduce parasitic impedance; the low-side MOSFET is most critical. Maintain a length to width ratio of