TS12012ITD1022

TS12011/TS12012 A 0.8V/1.5µA Nanopower Op Amp, Comparator, and Reference FEATURES DESCRIPTION            NanoWatt Analog™ Op Amp, Comparator, and 0.58V Reference in Single 4 mm2 Package Ultra Low Total Supply Current: 1.6µA (max) Supply Voltage Range: 0.8V to 2.5V Internal 0.58V Reference Op Amp and Comparator Input Ranges are Rail-to-Rail Unity-gain Stable Op Amp with AVOL = 104dB Op Amp Output: Rail-to-Rail and PhaseReversal-Free Internal ±7.5mV Comparator Hysteresis 20µs Comparator Propagation Delay Resettable Latched Comparator TS12011: Push-pull Rail-to-Rail Output TS12012: Open-drain Output APPLICATIONS Battery-powered Systems Single-Cell and +1.8V, +2.5V Powered Systems Low-Frequency, Local-Area Alarms/Detectors Smoke Detectors and Safety Sensors Infrared Receivers for Remote Controls Instruments, Terminals, and Bar-Code Readers Smart-Card Readers The TS12011/TS12012 combine a 0.58V reference, a 20µs comparator, and a unity-gain stable op amp in a single IC. All three devices operate from a single 0.8V to 2.5V power supply and consume less than 1.6µA total supply current. Supply current for all three functions over 0.8V to 2.5V supply range is guaranteed 1.6µA max. Super-flexible for crafting voltage detectors, timers, and wake-up circuits, these bundled functions exhibit low shoot-through currents and graceful power-down modes. Both the comparator and the op amp feature rail-to-rail input stages. The latching comparator exhibits ±7.5mV of internal hysteresis for clean, chatter-free output switching. When compared against similar products, the TS12011/TS12012 offer a factor-of-20 lower power consumption and at least a 55% reduction in pcb area. The TS12011’s comparator has a push-pull output stage with break-before-make switches for low shootthrough currents. The TS12012’s comparator has an open-drain output having no parasitic diode to VDD, for interfacing to wired-OR or mixed-voltage logic. The TS12011 and the TS12012 are fully specified over the -40°C to +85°C temperature range and each is available in a low-profile, 10-pin 2x2mm TDFN package with an exposed back-side paddle. TYPICAL APPLICATION CIRCUIT Part Number TS12011 TS12012 Comparator Output Stage Push-pull Open-Drain Page 1 © 2014 Silicon Laboratories, Inc. All rights reserved. TS12011/TS12012 ABSOLUTE MAXIMUM RATINGS Supply Voltage (VDD to VSS) ................................................. +2.75 V Input Voltage AMPIN+, AMPIN-…………………….….VSS – 0.3V to VDD + 0.3V COMPIN+, COMPIN-…..........................VSS – 0.3V to VDD + 0.3V LHDET………………………………..…….….. VSS - 0.3V to +5.5V Output Voltage AMPOUT, REFOUT……….………….....VSS – 0.3V to VDD + 0.3V COMPOUT (TS12011)………….........…VSS - 0.3V to VDD + 0.3V COMPOUT (TS12012)……...…..………….…VSS - 0.3V to +5.5V Differential Input Voltage (AMPIN, COMPIN)........................ ±2.75V Output Current AMPOUT, COMPOUT…………………...............................50mA Short-Circuit Duration (REFOUT, AMPOUT, COMPOUT)………………...….Continuous Continuous Power Dissipation (TA = +70°C) 10-Pin TDFN (Derate at 13.48mW/°C above +70°C) ......... 1078mW Operating Temperature Range ................................. -40°C to +85°C Junction Temperature……………………………………..……+150°C Storage Temperature Range .................................. -65°C to +150°C Lead Temperature (Soldering, 10s)...................................... +300°C Electrical and thermal stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only and functional operation of the device at these or any other condition beyond those indicated in the operational sections of the specifications is not implied. Exposure to any absolute maximum rating conditions for extended periods may affect device reliability and lifetime. PACKAGE/ORDERING INFORMATION ORDER NUMBER PART CARRIER QUANTITY MARKING ORDER NUMBER PART CARRIER QUANTITY MARKING TS12011ITD1022 Tape & Reel TS12012ITD1022 Tape & Reel ----- Tape & Reel 3000 ----- AAL TS12011ITD1022T AAM Tape & Reel 3000 TS12012ITD1022T Lead-free Program: Silicon Labs supplies only lead-free packaging. Consult Silicon Labs for products specified with wider operating temperature ranges. Page 2 TS12011/12 Rev. 1.0 TS12011/TS12012 ELECTRICAL CHARACTERISTICS VDD = 0.8V; VSS = 0V; VCOMPIN+/- = 0V; VAMPIN+/- = 0V; VAMPOUT = (VDD + VSS)/2; VCOMPOUT = HiZ; TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C. See note 1. PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS Supply Voltage VDD 0.8 2.5 V 1.1 1.6 TA = +25°C µA Supply Current IDD REFOUT = open -40°C ≤ TA ≤ 85°C 2 REFERENCE SECTION 555 577 600 TA = +25°C Reference Output mV VDD = 0.8V or 2.5V VREFOUT Voltage -40°C ≤ TA ≤ 85°C 552 602 Reference Load 0.5 % IOUT = ±100nA Regulation AMPLIFIER SECTION 3.5 mV TA = +25°C Input Offset Voltage VOS VAMPIN+/- = VDD or VAMPIN+/- = VSS -40°C ≤ TA ≤ 85°C 7 Input Bias Current Input Offset Current Input Common-Mode Range Large-Signal Voltage Gain Gain-Bandwidth Product Phase Margin Slew Rate Common-Mode Rejection Ratio Power-Supply Rejection Ratio Output High Voltage Output Low Voltage Output Source Current Output Sink Current Output Load Capacitive Drive IIN+, INIOS VAMPIN+, VAMPIN- = (VDD – VSS)/2 IVR Guaranteed by Input Offset Voltage Test VSS AVOL RL = 100K to VDD/2; VSS + 50mV < VOUT < VDD - 50mV 90 0.01 20 nA 5 nA VDD V 104 dB GBWP RL = 100kΩ//20pF 15 kHz φM SR RL = 100kΩ//20pF RL = 100kΩ//20pF 70 6 deg V/ms CMRR 0V ≤ VIN(CM) ≤ 2.1V; VDD = 2.5V 50 75 dB PSRR 0.65V ≤ (VDD - VSS) ≤ 2.5V 50 75 dB VOH VOL RL = 100kΩ to VSS RL = 100kΩ to VDD ISC+ VAMPOUT = VSS 0.28 mA ISC- VAMPOUT = VDD 4.5 mA VDD – 50mV VSS + 50mV 50 COUT V V pF VHB IIN+, INIOS IVR COMPARATOR SECTION TA = +25°C VAMPIN+/- = VDD; VAMPIN+/- = VSS; See Note 2 -40°C ≤ TA ≤ 85°C See Note 3 VCOMPIN+, VCOMPIN- = VDD or VSS VCOMPIN+, VCOMPIN- = VDD or VSS Guaranteed by Input Offset Voltage Test VSS CMRR 0V ≤ VIN(CM) ≤ 2.1V; VDD = 2.5V 50 60 dB PSRR 0.8V ≤ (VDD - VSS) ≤ 2.5V 50 70 dB 30 20 30 20 µs µs µs µs V V V mA mA mA nA Input Offset Voltage VOS Input Hysteresis Input Bias Current Input Offset Current Input Voltage Range Common-Mode Rejection Ratio Power-Supply Rejection Ratio Low-to-High Propagation Delay High-to-Low Propagation Delay Output High Voltage Output Low Voltage Output Low Voltage Output Short-Circuit Current VAMPIN+, VAMPIN- = (VDD – VSS)/2 tPD+ tPDVOH VOL VOL ISC Open Drain Leakage TS12011/12 Rev. 1.0 VOVERDRIVE = 10mV; See Note 4 TS12011 VOVERDRIVE = 100mV; See Note 4 VOVERDRIVE = 10mV; See Note 4 VOVERDRIVE = 100mV; See Note 4 TS12011; IOUT = -100μA TS12011 ; IOUT = 100μA TS12012 ; IOUT = 100μA Sourcing; VCOMPOUT = VSS TS12011 ; Sinking; VCOMPOUT = VDD TS12012 ; Sinking; VCOMPOUT = VDD TS12012 ; VCOMPOUT = 5V 4.5 8 mV ±7.5 0.2 mV nA nA V 20 5 VDD VDD – 0.1 VSS + 0.1 VSS + 0.11 0.1 0.5 1.4 20 Page 3 TS12011/TS12012 VDD = 0.8V, VSS = 0V, VCOMPIN+/- = 0V, VAMPIN+/- = 0V, VAMPOUT = (VDD + VSS)/2, VCOMPOUT = HiZ. TA = -40°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C. See note 1. PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS CONTROL PIN SECTION 0.1 Comparator Latched Output 0.8V ≤ VDD ≤ 1.1V V VIL LHDET Input Low Voltage Enabled 1.1V < VDD ≤ 2.5V 0.2 VDD - 0.1 Comparator Latched Output 0.8V ≤ VDD ≤ 1.1V VIH V LHDET Input High Voltage Disabled 1.1V < VDD ≤ 2.5V 1 LHDET Input Leakage VLHDET = VSS; VLHDET = 5.5V 100 nA Note 1: All devices are 100% production tested at TA = +25°C and are guaranteed by characterization for TA = TMIN to TMAX, as specified. Note 2: VOS is defined as the center of the hysteresis band at the input minus VIN(CM). Note 3: The hysteresis-related trip points are defined by the edges of the hysteresis band and measured with respect to the center of the hysteresis band. Note 4: The propagation delays are specified with an output load capacitance of CL = 15pF. VOVERDRIVE is defined above and is beyond the offset voltage and hysteresis of the comparator input. Page 4 TS12011/12 Rev. 1.0 TS12011/TS12012 TYPICAL PERFORMANCE CHARACTERISTICS VDD = 2.5V; VSS = 0V; VAMPOUT = HiZ; VCOMPOUT = HiZ, unless otherwise noted. Typical values are at TA = +25°C. Supply Current vs Supply Voltage and Temperature Reference Voltage vs Temperature 0.589 REFERENCE VOLTAGE - V SUPPLY CURRENT - µA 1.6 TA = +85ºC 1.4 TA = +25ºC 1.2 1 TA = -40ºC 0.587 0.585 0.583 0.581 0.8 0.8 1.23 1.65 2.08 2.5 -15 -40 SUPPLY VOLTAGE - V 60 85 Comparator Short-Circuit Current vs Supply Voltage 16 20 VAMPOUT = VSS SHORT-CIRCUIT CURRENT - mA SHORT-CIRCUIT CURRENT - mA 35 TEMPERATURE - ºC Op Amp Short-Circuit Current vs Supply Voltage 16 12 8 4 0 0.8 1.23 1.65 2.08 VCOMPOUT = VSS 12 8 4 0 2.5 0.8 1.23 1.65 2.08 SUPPLY VOLTAGE - V SUPPLY VOLTAGE - V Op Amp Short-Circuit Current vs Supply Voltage Comparator Short-Circuit Current vs Supply Voltage 2.5 18 45 VAMPOUT = VDD SHORT-CIRCUIT CURRENT - mA SHORT-CIRCUIT CURRENT - mA 10 38.5 32 25.5 19 0.8 1.23 1.65 2.08 SUPPLY VOLTAGE - V TS12011/12 Rev. 1.0 2.5 VCOMPOUT = VDD 12 6 0 0.8 1.23 1.65 2.08 2.5 SUPPLY VOLTAGE - V Page 5 TS12011/TS12012 TYPICAL PERFORMANCE CHARACTERISTICS VDD = 2.5V; VSS = 0V; VAMPOUT = HiZ; VCOMPOUT = HiZ, unless otherwise noted. Typical values are at TA = +25°C. Comparator Output Voltage Low vs Sink Current Comparator Output Voltage High vs Source Current 0.6 0.4 0.3 VOL - V VDD - VOH - V 0.4 0.2 0.2 0.1 0 0 0 1 2 3 4 0 3 SINK CURRENT - mA Op Amp Output Voltage High vs Source Current Op Amp Output Voltage Low vs Sink Current 0.35 0.5 0.28 0.4 VOL - V VDD - VOH - V 2 SOURCE CURRENT - mA 0.6 0.3 0.21 0.14 0.2 0.07 0.1 0 0 300 2 4 6 8 2 4 6 SOURCE CURRENT - mA SINK CURRENT - mA Op Amp Input Offset Voltage vs Supply Voltage Comparator Input Offset Voltage vs Supply Voltage 1 200 100 VINCM = VDD 0 -100 VINCM = VSS -200 VINCM = VSS 0.5 0 VINCM = VDD -0.5 -1 -300 0.8 1.23 1.65 2.08 SUPPLY VOLTAGE - V Page 6 0 INPUT OFFSET VOLTAGE - mV 0 INPUT OFFSET VOLTAGE - µV 1 2.5 0.8 1.23 1.65 2.08 2.5 SUPPLY VOLTAGE - V TS12011/12 Rev. 1.0 TS12011/TS12012 TYPICAL PERFORMANCE CHARACTERISTICS VDD = 2.5V; VSS = 0V; VAMPOUT = HiZ; VCOMPOUT = HiZ, unless otherwise noted. Typical values are at TA = +25°C. Op Amp Input Offset Voltage vs Input Common-Mode Voltage 0.8 INPUT OFFSET VOLTAGE - mV INPUT OFFSET VOLTAGE - mV 0.8 Op Amp Input Offset Voltage vs Input Common-Mode Voltage VDD = 0.8V 0.6 0.4 0.2 VDD = 2.5V 0.7 0.6 0.5 0.4 0 0.2 0.4 0.6 0.8 0 0.5 1 1.5 2 2.5 SUPPLY VOLTAGE - V TS12011 Op Amp Small-Signal Transient Response VDD = 2.5V, RLOAD = 100kΩ, CLOAD = 15pF TS12011 Op Amp Large Signal Transient Response VDD = 2.5V, RLOAD = 100kΩ, CLOAD = 15pF OUTPUT 1V/DIV OUTPUT 50mV/DIV INPUT 1V/DIV INPUT 50mV/DIV SUPPLY VOLTAGE - V 500µs/DIV TS12011 Comparator Propagation Delay (TPD+) VDD = 2.5V, VOVERDRIVE = 100mV, CLOAD = 15pF TS12011 Comparator Propagation Delay (TPD-) VDD = 2.5V, VOVERDRIVE = 100mV, CLOAD = 15pF OUTPUT 1V/DIV OUTPUT 1V/DIV INPUT 50mV/DIV INPUT 50mV/DIV 200µs/DIV 20µs/DIV TS12011/12 Rev. 1.0 20µs/DIV Page 7 TS12011/TS12012 TYPICAL PERFORMANCE CHARACTERISTICS VDD = 2.5V; VSS = 0V; VAMPOUT = HiZ; VCOMPOUT = HiZ, unless otherwise noted. Typical values are at TA = +25°C. Gain and Phase vs Frequency 100 50 GAIN - dB PHASE 40 50 30 0 20 -50 GAIN 10 -100 14kHz VDD = 0.8V TA = +25ºC RL = 100kΩ CL = 20pF AVCL = 1000V/V 0 -10 -200 -20 100 1k -150 PHASE - Degrees 70º 10k -250 100k FREQUENCY - Hz PIN FUNCTIONS Page 8 PIN TS12011 1 2 3 4 PIN TS12012 1 2 3 4 AMPOUT AMPINAMPIN+ VSS 5 5 LHDET 6 7 8 8 7 6 COMPIN+ REFOUT COMPIN- 9 9 COMPOUT 10 10 VDD EP EP ---- NAME FUNCTION Amplifier Output Amplifier Inverting Input Amplifier Non-inverting Input Negative Supply Voltage. Latch Enable Pin, active low. Tie to VDD for normal operation. Do not leave floating. See Latch Truth Tables below. Comparator Non-inverting Input 0.58V Reference Output Comparator Inverting Input Comparator Output. TS12011: push-pull TS12012: open-drain Positive Supply Voltage. Connect a 0.1µF bypass capacitor from this pin to analog VSS/GND. Exposed paddle is electrically connected to VSS/GND. TS12011/12 Rev. 1.0 TS12011/TS12012 BLOCK DIAGRAM THEORY OF OPERATION The TS12011 and TS12012 are multi-purpose CMOS building blocks intended for creating analog glue functions around battery-powered uC systems. There is an op amp for signal conditioning, a comparator for detection, and a reference to establish detection threshold levels. It’s possible to build a wide variety of timers, event detectors, regulators, and voltage monitors using these flexible uncommitted blocks. Optimized for low-voltage operation, these devices draw less than 1.6uA total from a 0.8V to 2.5V supply. The op amp and comparator blocks typically continue to function down to less than 0.5V (REFOUT will go into dropout, however). Comparator The comparator block is designed for high gain and chatter-free output switching in noisy environments. The comparator inputs have rail-to-rail VIN range, TS12011/12 Rev. 1.0 and exhibit +/-7.5mV of hysteresis. The only difference between the two device types is in the output stage of the comparator. The TS12011 has a push-pull output and latches in the high state. The TS12012 has an open-drain output, latches in the low state, and can tolerate pull-up voltages higher than the supply (up to 5.5V absolute max above VSS/GND). TS12011 push-pull output driver was designed to minimize supply-current surges while driving ±100µA loads with an output swing to within 100mV of the supply rails. The TS12011 and the TS12012 can sink 0.5mA and 1.4mA of current, respectively. The TS12011 can source 0.1mA of current. The non-traditional latch function works to detect and latch changes in the input state. If the LHDET control input is enabled, the output will latch high (low for the TS12012) whenever the differential input voltage is high enough to force a change in that direction. If the differential voltage is in the wrong direction to force a Page 9 TS12011/TS12012 change, the comparator stays active and waits for the crossing, at which point it will latch in its final state. An internal POR circuit ensures that the latch powers up in the “comparator active” state if LHDET is low when VDD is first applied. Latch Truth Table – TS12011 CMPIN+ CMPOUT to CMPINinitial LHDET difference state voltage HIGH X N/A LOW HIGH X LOW LOW negative LOW LOW positive CMPOUT Normal operation HIGH (latched) LOW (comparator active) HIGH (latched) X = Don’t Care Latch Truth Table – TS12012 CMPIN+ CMPOUT to CMPINinitial LHDET difference state voltage HIGH X N/A LOW LOW X LOW HIGH positive LOW HIGH negative CMPOUT Normal operation LOW (latched) HIGH (comparator active) LOW (latched) X = Don’t Care Reference The TS12011 and TS12012 on-board 0.58V ±4.5% reference voltage can source and sink 0.1µA and 0.1µA of current and can drive a capacitive load less than 50pF and greater than 50nF with a maximum capacitive load of 250nF. The higher the capacitive load, the lower the noise on the reference voltage and the longer the time needed for the reference voltage to respond and become available on the REFOUT pin. With a 250nF capacitive load, the reference voltage will settle to within specifications in approximately 20ms. Page 10 Op Amp The TS12011 and TS12012 have a unity-gain stable op-amp with a GBWP of 15kHz, a slew rate of 6V/ms, and can drive a capacitive load up to 50pF. The common mode input voltage range extends from VSS to VDD and the input bias current and offset current are less than 20nA and 2nA, respectively. Op-Amp Stability The TS12011 and TS12012 op-amp is able to drive up to 50pF of capacitive load and still maintain stability in a unity-gain configuration with a 15kHz GBWP and a phase margin of 70 degrees with a 100kΩ//20pF output load. Though the TS12011 and TS12012 address low frequency applications, it is essential to perform good layout techniques in order to minimize board leakage and stray capacitance, which is of a concern in low power, high impedance circuits. For instance, a 10MΩ resistor coupled with a 1pF stray capacitance can lead to a pole at approximately 15kHz, which is the GBWP of the device. If stray capacitance is unavoidable, a feedback capacitor can be placed in parallel with the feedback resistor. APPLICATIONS INFORMATION Comparator Hysteresis As a result of circuit noise or unintended parasitic feedback, many analog comparators often break into oscillation within their linear region of operation especially when the applied differential input voltage approaches 0V (zero volt). Externally-introduced hysteresis is a well-established technique for stabilizing analog comparator behavior and requires external components. As shown in Figure 1, adding comparator hysteresis creates two trip points: VTHR (for the rising input voltage) and VTHF (for the falling input voltage). The hysteresis band (VHB) is defined as the voltage difference between the two trip points. When a comparator’s input voltages are equal, hysteresis effectively forces one comparator input to move quickly past the other input, moving the input out of the region where oscillation occurs. Figure 1 illustrates the case in which an IN- input is a fixed voltage and an IN+ is varied. If the input signals were reversed, the figure would be the same with an inverted output. To save cost and external pcb area, an internal ±7.5mV hysteresis circuit was added to the TS12011 and TS12012. TS12011/12 Rev. 1.0 TS12011/TS12012 and if IR2 = 150nA is chosen, then the formulae above produce two resistor values: 3.87MΩ and 12.8MΩ - a 4.02MΩ standard value for R2 is selected. 2) Next, the desired hysteresis band (VHYSB) is set. In this example, VHYSB is set to 100mV. 3) Resistor R1 is calculated according to the following equation: R1 = R2 x (VHYSB/VDD) Figure 1. TS12011/TS12012 Threshold Hysteresis Band and substituting the values selected in 1) and 2) above yields: Adding Hysteresis to the TS12011 Push-pull Output Option R1 = 4.02MΩ x (100mV/2.5V) = 160.8kΩ. Additional hysteresis can be generated with three external resistors using positive feedback as shown in Figure 2. Unfortunately, this method also reduces the hysteresis response time. The procedure to calculate the resistor values for the TS12011 is as follows: The 160kΩ standard value for R1 is chosen. 4) The trip point for COMPIN+ rising (VTHR) is chosen such that VTHR > VREFOUT x (R1 + R2)/R2 (VTHF is the trip point for VCOMPIN+ falling). This is the threshold voltage at which the comparator switches its output from low to high as VCOMPIN+ rises above the trip point. In this example, VTHR is set to 2. 5) With the VTHR from Step 4 above, resistor R3 is then computed as follows: R3 = 1/[VTHR/(VREFOUT x R1) - (1/R1) - (1/R2)] R3 = 1/[2V/(0.58V x 160kΩ) - (1/160kΩ) (1/4.02MΩ)] = 66.43kΩ Figure 2. Using Three Resistors Introduces Additional Hysteresis in the TS12011 1) Setting R2. As the leakage current at the IN pin is less than 20nA, the current through R2 should be at least 150nA to minimize offset voltage errors caused by the input leakage current. The current through R2 at the trip point is (VREFOUT - VCOMPOUT)/R2. In this example, a 69.8kΩ, 1% standard value resistor is selected for R3. 6) The last step is to verify the trip voltages and hysteresis band using the standard resistance values: For VCOMPIN+ rising: In solving for R2, there are two formulas – one each for the two possible output states: VTHR = VREFOUT x R1 [(1/R1) + (1/R2) + (1/R3)] = 1.93V R2 = VREFOUT/IR2 or R2 = (VDD - VREFOUT)/IR2 From the results of the two formulae, the smaller of the two resulting resistor values is chosen. For example, when using the TS12011 (VREFOUT = 0.58V) at a VDD = 2.5V TS12011/12 Rev. 1.0 For VCOMPIN+ falling: VTHF = VTHR - (R1 x VDD/R2) = 1.83V and Hysteresis Band = VTHR – VTHF = 100mV Page 11 TS12011/TS12012 Adding Hysteresis to the TS12012 Open-Drain Option The TS12012 has open-drain output and requires an external pull-up resistor to VDD as shown in Figure 3. R3 = 1/[VTHR/(VREFOUT x R1) - (1/R1) - (1/R2)] 6) As before, the last step is to verify the trip voltages and hysteresis band with the standard resistor values used in the circuit: For VCOMPIN+ rising: VTHR = VREFOUT x R1 x (1/R1+1/R2+1/R3) For VCOMPIN+ falling: VTHF = VREFOUT x R1 x(1/R1+1/R3+1/(R2+R4)) -(R1/(R2+R4)) x VDD and Hysteresis Band is given by VTHR – VTHF Figure 3. Using Four Resistors Introduces Additional Hysteresis in the TS12012 Additional hysteresis can be generated using positive feedback; however, the formulae differ slightly from those of the push-pull option TS12011. The procedure to calculate the resistor values for the TS12012 is as follows: 1) As in the previous section, resistor R2 is chosen according to the formulae: PC Board Layout and Power-Supply Bypassing While power-supply bypass capacitors are not typically required, it is good engineering practice to use 0.1uF bypass capacitors close to the device’s power supply pins when the power supply impedance is high, the power supply leads are long, or there is excessive noise on the power supply traces. To reduce stray capacitance, it is also good engineering practice to make signal trace lengths as short as possible. Also recommended are a ground plane and surface mount resistors and capacitors. R2 = VREFOUT/150nA Input Noise or where the smaller of the two resulting resistor values is the best starting value. Radiated noise is common in low power circuits that require high impedance circuits. To minimize this effect, all traces between the inputs of the comparator or op-amp and passive component networks should be made as short as possible. 2) As before, the desired hysteresis band (VHYSB) is set to 100mV. Pilot Light Flame Detector with Low-Battery Lockout Circuit 3) Next, resistor R1 is then computed according to the following equation: The TS12011 can be used to create a pilot flame detector with low-battery lockout circuit as shown in Figure 4. The circuit is able to detect when the thermocouple does not detect the pilot flame and when the battery in the circuit drops to 1.39V. This circuit makes use of the op-amp, comparator, and 0.58V reference in the TS12011. In this example, a type R thermocouple is used. It generates a voltage range from 9mV to 17mV that corresponds to a temperature range of 900ºC to 1500ºC, which is typical of a methane pilot flame. If the pilot flame is removed, the temperature drops; hence, the output voltage generated by the thermocouple is drops to a minimum voltage of 0.1mV that is applied to the non- R2 = (VDD- VREFOUT)/150nA - R4 R1 = (R2 + R4) x (VHYSB/VDD) 4) The trip point for VCOMPIN+ rising (VTHR) is chosen (again, remember that VTHF is the trip point for VCOMPIN+ falling). This is the threshold voltage at which the comparator switches its output from low to high as VCOMPIN+ rises above the trip point. 5) With the VTHR from Step 4 above, resistor R3 is computed as follows: Page 12 TS12011/12 Rev. 1.0 TS12011/TS12012 inverting input of the op-amp. This switches the output voltage of the op-amp to a LOW state and in turn, switches Q1 off. If, however, the battery voltage drops from 1.5V to 1.39V, the comparator output will switch from an output HIGH to a LOW. This will turn off Q2 and the output of the op-amp will turn Q1 off. The complete circuit consumes approximately 95µA of supply current at VDD = 1.5V. Figure 4. Pilot Light Flame Detector with Low-Battery Lockout Circuit TS12011/12 Rev. 1.0 Page 13 TS12011/TS12012 Figure 5. Sawtooth/Triangle Generator with Stable Frequency and Amplitude Page 14 TS12011/12 Rev. 1.0 TS12011/TS12012 Figure 6. Low-power One-shot and Latch Circuits TS12011/12 Rev. 1.0 Page 15 TS12011/TS12012 Figure 7. Adjustable Buffered Reference Generators Page 16 TS12011/12 Rev. 1.0 TS12011/TS12012 PACKAGE OUTLINE DRAWING 10-Pin TDFN22 Package Outline Drawing (N.B., Drawings are not to scale) Patent Notice Silicon Labs invests in research and development to help our customers differentiate in the market with innovative low-power, small size, analog-intensive mixed-signal solutions. Silicon Labs' extensive patent portfolio is a testament to our unique approach and world-class engineering team. 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