L6565
QUASI-RESONANT SMPS CONTROLLER
QUASI-RESONANT (QR) ZERO-VOLTAGESWITCHING (ZVS) TOPOLOGY
■ LINE FEED FORWARD TO DELIVER
CONSTANT POWER vs. MAINS CHANGE
■ FREQUENCY FOLDBACK FOR OPTIMUM
STANDBY EFFICIENCY
■ PULSE-BY-PULSE & HICCUP-MODE OCP
■ ULTRA-LOW START-UP (< 70µA) AND
QUIESCENT CURRENT (< 3.5mA)
■ DISABLE FUNCTION (ON/OFF CONTROL)
■ 1% PRECISION (@ T j = 25°C) INTERNAL
REFERENCE VOLTAGE
■ ±400mA TOTEM POLE GATE DRIVER WITH
UVLO PULL-DOWN
■ BLUE ANGEL, ENERGY STAR, ENERGY
2000 COMPLIANT
APPLICATIONS
■ TV/MONITOR SMPS
■ AC-DC ADAPTERS/CHARGERS
■ DIGITAL CONSUMER
■ PRINTERS, FAX MACHINES,
PHOTOCOPIERS AND SCANNERS
■
DIP8(Minidip)
SO-8
ORDERING NUMBERS:
L6565N
L6565D
DESCRIPTION
The L6565 is a current-mode primary controller IC,
specifically designed to build offline Quasi-resonant
ZVS (Zero Voltage Switching at switch turn-on) flyback converters.
Quasi-resonant operation is achieved by means of a
transformer demagnetization sensing input that triggers MOSFET's turn-on.
BLOCK DIAGRAM
COMP
VFF
2
3
1
INV
-
LINE VOLTAGE
FEEDFORWARD
+
40K
2.5V
4
CS
2V
VOLTAGE
REGULATOR
-
+
-
8
VCC
5pF
+
Hiccup-mode
OCP
INTERNAL
SUPPLY
R
Q
S
Q
VCC
20V
R1
+
R2
7
GD
UVLO
DRIVER
-
VREF2
Blanking
START
ZERO CURRENT
DETECTOR
Starter
STOP
+
BLANKING
2.1V
1.6V
Hiccup-mode
OCP
5
ZCD
January 2003
STARTER
-
DISABLE
6
GND
1/17
L6565
DESCRIPTION (continued)
Converter's power capability variations with the mains voltage are compensated by line voltage feedforward.
At light load the device features a special function that automatically lowers the operating frequency still maintaining the operation as close to ZVS as possible. In addition to very low start-up and quiescent currents, this
feature helps keep low the consumption from the mains at light load and be Blue Angel and Energy Star compliant.
The IC includes also a disable function, an on-chip filter on current sense, an error amplifier with a precise reference voltage for primary regulation and an effective two-level overcurrent protection.
PIN CONNECTION (Top view, Minidip and SO8)
INV
1
8
Vcc
COMP
2
7
GD
VFF
3
6
GND
CS
4
5
ZCD
PIN DESCRIPTION
N°
Name
Function
1
INV
Inverting input of the error amplifier. The information on the output voltage is fed into the pin
through either a resistor divider (primary regulation) or an optocoupler (secondary feedback).
This pin can be grounded in some secondary feedback schemes (see pin 2).
2
COMP
Output of the error amplifier. Typically, a compensation network is placed between this pin and
the INV pin to achieve stability and good dynamic performance of the voltage control loop. With
secondary feedback, the pin can be also driven directly by an optocoupler to control PWM by
modulating the current sunk from the pin (with the INV pin grounded).
3
VFF
Line voltage feedforward. The information on the converter’s input voltage is fed into the pin
through a resistor divider and is used to change the setpoint of the pulse-by-pulse current
limitation (the higher the voltage, the lower the setpoint). If this function is not desired the pin will
be grounded and the current limitation setpoint will be maximum.
4
CS
Input to the PWM comparator. The primary current is sensed through a resistor, the resulting
voltage is applied to this pin and compared with an internal reference to determine MOSFET’s
turn-off. The internal reference is clamped at a value, which defines the pulse-by-pulse current
limitation setpoint, depending on the voltage at pin VFF. If the signal at the pin CS exceeds 2 V,
the gate driver will be disabled (Hiccup-mode OCP).
5
ZCD
Transformer’s demagnetization sensing input for Quasi-Resonant operation. Alternately,
synchronization input for an external signal. A negative-going edge triggers MOSFET’s turn-on.
The trigger circuit is blanked for a minimum of 3.5 µs after MOSFET turn-off, for safe operation
under short circuit conditions and frequency foldback. If the pin is grounded the IC will be
disabled.
6
GND
Ground. Current return for both the signal part of the IC and the gate driver.
7
GD
Gate driver output. The totem pole output stage is able to drive power MOSFET’s and IGBT’s
with a peak current of 400 mA (source and sink).
8
Vcc
Supply Voltage of both the signal part of the IC and the gate driver. An electrolytic capacitor is
connected between this pin and ground. A resistor connected from this pin to the converter’s
input bulk capacitor will be typically used to start up the device.
2/17
L6565
THERMAL DATA
Symbol
Rth j-amb
Parameter
Max. Thermal Resistance, Junction-to-ambient
SO8
Minidip
Unit
150
100
°C/W
ABSOLUTE MAXIMUM RATINGS
Symbol
Pin
IVcc
8
ICC + IZ
IGD
7
Output Totem Pole Peak Current (2 µs)
INV, COMP,
VFF, CS
Parameter
Value
Unit
30
mA
±700
mA
-0.3 to 7
V
50 (source)
-10 (sink)
mA
1
0.65
W
Junction Temperature Operating range
-40 to 150
°C
Storage Temperature
-55 to 150
°C
1, 2, 3 4 Analog Inputs & Outputs
5
IZCD
Zero Current Detector
Ptot
Power Dissipation @Tamb = 50°C
Tj
Tstg
(Minidip)
(SO8)
ELECTRICAL CHARACTERISTCS
(Tj = -25 to 125°C, VCC = 12V, Co = 1nF; unless otherwise specified)
Symbol
Parameter
Test Condition
Min.
Typ.
Max.
Unit
SUPPLY VOLTAGE
Vcc
Operating range
VCCOn
Turn-on threshold
12.5
13.5
14.5
V
VCCOff
Turn-off threshold
8.7
9.5
10.3
V
3.65
4
4.3
V
18
20
22
V
Hys
Hysteresis
VZ
Zener Voltage
After turn-on
Icc = 25 mA
10.3
18
SUPPLY CURRENT
Istart-up
Start-up Current
Before turn-on, VCC = 12V
45
70
µA
Quiescent Current
After turn-on
2.3
3.5
mA
Operating Supply Current
@ 70 kHz
3.5
5
mA
Iq
Quiescent Current
During Hiccup-mode OCP
3.5
mA
Iq
Quiescent Current
VZCD < VDIS, VCC>VCCOff
1.4
2.1
mA
-1
µA
Iq
ICC
1.6
LINE FEEDFORWARD
IVFF
Input Bias Current
VVFF
Operating Range
K
Gain
VVFF = 0 to 3 V
0 to 3
V
0.16
VVFF = 1.5V, VCOMP = 4V
ERROR AMPLIFIER
VINV
Voltage Feedback Input
Threshold
Line Regulation
IINV
Input Bias Current
Tamb = 25°C
2.465
12V < VCC < 18V
2.44
Vcc = 12 to 18V
2.5
2.535
V
2.56
2
5
mV
-0.1
-1
µA
3/17
L6565
ELECTRICAL CHARACTERISTCS (continued)
(Tj = -25 to 125°C, VCC = 12V, Co = 1nF; unless otherwise specified)
Symbol
Parameter
Min.
Typ.
Open loop
60
80
Source Current
VCOMP = 4V, VINV = 2.4 V
-2
-3.5
Sink Current
VCOMP = 4V, VINV = 2.6 V
2.5
4.5
GV
Voltage Gain
GB
Gain-Bandwidth Product
ICOMP
VCOMP
Test Condition
Max.
dB
1
Upper Clamp Voltage
ISOURCE = 0.5 mA
Lower Clamp Voltage
ISINK = 0.5 mA
5
2.25
VCS = 0
Unit
MHz
-5
mA
mA
5.5
V
2.55
V
-0.05
-1
µA
200
450
ns
V
CURRENT SENSE COMPARATOR
ICS
Input Bias Current
td(H-L)
Delay to Output
VCSx
Current Sense Reference Clamp
VCOMP = Upper clamp, VVFF = 0V
1.28
1.4
1.5
VCOMP = Upper clamp, VVFF = 1.5V
0.62
0.7
0.78
0
0.2
1.85
2.0
2.2
V
VCOMP = Upper clamp, VVFF = 3V
VCSdis
Hiccup-mode OCP level
ZERO CURRENT DETECTOR/ SYNCHRONIZATION
VZCDH
Upper Clamp Voltage
IZCD = 3mA
4.7
5.2
6.1
V
VZCDL
Lower Clamp Voltage
IZCD = - 3mA
0.3
0.65
1
V
VZCDA
Arming Voltage
(positive-going edge)
(1)
VZCDT
Triggering Voltage
(negative-going edge)
IZCDb
Input Bias Current
VZCD = 1 to 4.5 V
2.1
V
1.6
V
2
µA
IZCDsrc
Source Current Capability
-3
-10
mA
IZCDsnk
Sink Current Capability
3
10
mA
VDIS
Disable Threshold
IZCDr
Restart Current After Disable
VZCD < VDIS, Vcc > Vccoff
Blanking time after pin 7 high-tolow transition
VCOMP ≥ 3.2 V
3.5
VCOMP = 2.5 V
18
TBLANK
150
200
250
mV
-70
-150
-230
µA
µs
START TIMER
tSTART
Start Timer period
250
400
550
µs
IGDsource = 200mA
1.2
2
V
IGDsource = 20mA
0.7
GATE DRIVER
VOL
Dropout Voltage
VOH
2
IGDsink = 20mA
0.3
V
tf
Current Fall Time
40
100
ns
tr
Current Rise Time
40
100
ns
IGDoff
IGD sink current
Vcc = 4 V, VGD = 1 V
(1) Parameters guaranteed by design, not tested in production.
4/17
1
IGDsink = 200mA
5
10
mA
L6565
Figure 1. Supply current vs. Supply voltage
ICC
(mA)
Figure 4. Line feedforward characteristics
Vcsx [V]
1.5
Upper clamp
10
5.0 V
5
1
1
4.5 V
0.5
4.0 V
0.1
0.5
3.5 V
0.05
CL = 1nF
f = 70KHz
TA = 25°C
0.01
0.005
3.0 V
0
VCOMP = 2.5V
0
0
0
5
10
15
20
0.5
1
VCC(V)
Figure 2. Start-up & UVLO vs. Temperature
1.5
2
2.5
3
3.5
VVFF [V]
Figure 5. Pin 2 (COMP) V-I characteristics
VCOMP [V]
14
VCC-ON
(V)
13
6
Tj = 25 °C
Vpin1 = 0
5
4
12
3
11
Regulation
range
2
10
VCC-OFF
(V)
9
-25
1
0
0
25
50
75
100
125
0
1
2
3
4
ICOMP [mA]
T (°C)
Figure 3. Feedback reference vs. Temperature
VREF
(V)
D94IN048A
Figure 6. ZCD blanking time vs. COMP voltage
TBLANK [µs]
20
Tj = 25 °C
15
2.50
10
2.48
5
0
2
2.46
-50
0
50
100
T (°C)
3
4
5
6
VCOMP [V]
5/17
L6565
Figure 7. Gate-drive output saturation
Figure 10. Zener voltage at Vcc pin vs. Tj
Vpin7 [V]
Vz [V]
2.5
22
Tj = 25 °C
Vcc = 14.5 V
SINK
2
21
1.5
20
1
19
0.5
0
0
100
200
300
400
500
18
-50
0
50
100
150
Tj [°C]
IGD [mA]
Figure 11. Start-up timer period vs. Tj
Figure 8. Gate-drive output saturation
TSTART [µs]
Vpin7 [V]
0
Vcc - 0.5
450
Tj = 25 °C
Vcc = 14.5 V
SOURCE
Vcc --0.5
0.5
Vcc=12V
400
Vcc - 1.0
-1
350
Vcc --1.5
1.5
300
Vcc - 2.0
-2
Vcc - 0.5
-2.5
0
100
200
300
400
500
IGD [mA]
Icc [mA]
5
Vcc=12V
2
Quiescent
1
0.5
0.2
0.1
0.02
-50
Before Start-up
0
50
Tj [°C]
6/17
0
50
Tj [°C]
Figure 9. IC consumption vs. temperature
0.05
250
-50
100
150
100
150
L6565
APPLICATION INFORMATION
Quasi-resonant operation in offline flyback converters lies in synchronizing MOSFET's turn-on to the transformer's demagnetization. Detecting the resulting negative-going edge of the voltage across any winding of the
transformer can do this. The L6565 is provided with a dedicated pin that allows doing the job with a very simple
interface, just one resistor.
Variable frequency operation - as a result of different operating conditions in terms of input voltage and output
current - is inherent in such functionality. The system always works close to the boundary between DCM (Discontinuous Conduction Mode) and CCM (Continuous Conduction Mode) operation of the transformer. The operation is then identical to that of the so-called self-oscillating or Ringing Choke Converter (RCC).
Detailed Device Description
Internal Supply Block (see fig. 12)
A linear voltage regulator supplied by Vcc (pin 8) generates an internal 7V rail used for supplying the entire IC,
except for the gate driver that is supplied directly from Vcc. In addition, a bandgap circuit generates a precise
internal reference (2.5V±1% @ 25°C) used by the control loop to ensure a good regulation with primary feedback technique.
In figure 12 it is also shown the undervoltage lockout (UVLO) comparator with hysteresis used to enable the
chip as long as the Vcc voltage is high enough to ensure a reliable operation.
Figure 12. L6565 internal supply block
+Vin
Vcc
8
+
LIN.
REG.
UVLO
REF.
2.5V
7V bus
7/17
L6565
Zero Current Detection and Triggering Block (see fig. 13):
The Zero Current Detection (ZCD) block switches on the external MOSFET if a negative-going edge falling below 1.6 V is applied to the input (pin 5, ZCD). However, to ensure high noise immunity, the triggering block must
be armed first: prior to falling below 1.6V, the voltage on pin 5 must experience a positive-going edge exceeding
2.1 V.
This feature is typically used to detect transformer demagnetization for QR operation, where the signal for the
ZCD input is obtained from the transformer's auxiliary winding used also to supply the IC. Alternatively, this can
be used to synchronize MOSFET's turn-on to the negative-going edge of an external clock signal, in case the
device is not required to work in QR mode but as a standard PWM controller in a synchronized system (e.g.
monitor SMPS).
The triggering block is blanked for a certain time after the MOSFET has been turned off. This has two goals:
first, to prevent any negative-going edge that follows leakage inductance demagnetization from triggering the
ZCD circuit erroneously; second, to realize the Frequency Foldback function (see the relevant description).
Figure 13. Zero Current Detection and Triggering Block; Disable and Frequency Foldback Blocks
COMP
L6565
INV
E/A
+
2.5V
RZCD
5
ZCD
150µA
+Vin
5.2V
-
Q
BLANKING
TIME
1.6V
2.1V
+
PWM
blanking
START
7
R
Q
MONO
STABLE
S
STARTER
0.2V
0.3V
to line
FFWD
DRIVER
GD
starter STOP
DISABLE
+
A circuit is needed that turns on the external MOSFET at start-up since no signal is coming from the ZCD pin.
This is realized with an internal starter, which forces the driver to deliver a pulse to the gate of the MOSFET.
To minimize the external interface with the synchronization source (either the auxiliary winding or an external
clock), the voltage at the pin is both top and bottom limited by a double clamp, as illustrated in the internal diagram of the ZCD block of figure 13. The upper clamp is typically located at 5.2 V, while the lower clamp is at
one VBE above ground. The interface will then be made by just one resistor that has to limit the current sourced
by and sunk from the pin within the rated capability of the internal clamps.
Disable Block (see fig. 13):
The ZCD pin is used also to activate the Disable Block. If the voltage on the pin is taken below 150 mV the device will be shut down. To do so, it is necessary to override the source capability (10 mA max.) of the internal
lower clamp. While in disable, the current consumption of the IC will be reduced. To re-enable device operation,
the pull-down on the pin must be released.
Frequency Foldback Block (see fig. 13):
To prevent the switching frequency from reaching too high values, which is a typical drawback of QR operation,
8/17
L6565
the L6565 puts a limit on the minimum OFF-time of the switch. This is done by blanking the triggering block of
the ZCD circuit as mentioned before. The duration of the blanking time (3.5µs min.) is a function of the error
amplifier output VCOMP, as shown in the diagram of figure 6.
If the load current and the input voltage are such that the switch OFF-time falls below the minimum blanking
time of 3.5µs, the system will enter the "Frequency Foldback" mode, a sort of "ringing cycle skipping" illustrated
schematically in figure 14.
Figure 14. Frequency foldback: ringing cycle skipping as the load is progressively reduced
VDS
VDS
VDS
t
TFW
t
t
TV
TBLANKmin
TBLANK
Pin = Pin'
(limit condition)
TBLANK
Pin = Pin'' < Pin'
Pin = Pin''' < Pin''
In this mode, uneven switching cycles may be observed under some line/load conditions, due to the fact that
the OFF-time of the MOSFET is allowed to change with discrete steps (2·Tv), while the OFF-time needed for
cycle-by-cycle energy balance may fall in between. Thus one or more longer switching cycles will be compensated by one or more shorter ones and vice versa. However, this mechanism is absolutely normal and there is
no appreciable effect on the performance of the converter or on its output voltage.
Figure 15. Frequency Foldback: qualitative
frequency dependence on power
throughput
fsw
BURST MODE
00 00 00 00
0000
00 0 0 0
00 000 000 000
without frequency foldback
Vin fixed
with frequency foldback
Voltage Feedforward block (see fig. 17b):
The power that QR flyback converters with a fixed
overcurrent setpoint (like fixed-frequency systems)
are able to deliver changes with the input voltage
considerably. With wide-range mains, at maximum
line it can be more than twice the value at minimum
line, as shown by the upper curve in the diagram of
figure 16. The L6565 has the Line Feedforward function available to solve this issue.
Figure 16. Typical power capability change vs.
input voltage in ZVS QR flyback
converters
2.5
system not
compensated
Pin
Further load reductions involve lower values for
VCOMP, which increases the blanking time. Therefore, more and more ringing cycles will be skipped.
When the load is low enough, so many ringing cycles
need to be skipped that their amplitude becomes
very small and they can no longer trigger the ZCD circuit. In that case the internal starter of the IC will be
activated, resulting in burst-mode operation: a series
of few switching cycles spaced out by long periods
where the MOSFET is in OFF state.
Pinlim @ Vin
Pinlim @ Vinmin
2
1.5
1
system optimally
compensated
0.5
1
1.5
2
2.5
3
3.5
4
Vin
Vinmin
9/17
L6565
It acts on the clamp level of the control voltage Vcsx, that is on the overcurrent setpoint, so that it is a function
of the converter's input voltage sensed through a dedicated pin (#3, VFF): the higher the input voltage, the lower
the setpoint. This is illustrated in the diagram of figure 17a that shows the relationship between the voltage at
the pin VFF and Vcsx (with the error amplifier saturated high in the attempt of keeping output voltage regulation).
The schematic in figure 17b shows also how the function is included in the control loop. With a proper selection
of the external divider R1-R2 it is possible to achieve the optimum compensation described by the lower curve
in the diagram of figure 16.
In applications where this function is not wanted, e.g. because of a narrow input voltage range, the VFF pin can
be simply grounded, thus saving the resistor divider. The overcurrent setpoint will be then fixed at the maximum
value of about 1.4V (1.5V max.).
Line Feedforward is also beneficial to other characteristics of quasi-resonant converters: it improves their input
ripple rejection ability and limits the variation of the power stage's small-signal gain versus the line voltage.
Figure 17. a) Overcurrent setpoint vs. VFF voltage; b) Line Feedforward function block
Vcsx [V]
1.5
VCOMP = Upper clamp
1
a)
0.5
0
0
0.5
1
1.5
2
2.5
3
3.5
VVFF [V]
+Vin
R1
R2
Rs
COMP
2
VFF
CS
3
ZCD
4
5
ZCD
PWM
2V
Hiccup
-
2.5V
b)
10/17
S
7
Q
DRIVER
GD
R
(reset-dominant)
+
E/A
+
-
-
VOLTAGE
FEED
FORWARD
starter STOP
+
INV 1
STARTER
DISABLE
L6565
L6565
Error Amplifier Block (see fig. 17b):
The Error Amplifier (E/A) inverting input is used in primary feedback technique to compare a partition of the voltage generated by the auxiliary winding with the internal reference, to achieve converter's output voltage regulation (see "Application Ideas", fig. 24). With secondary feedback (typically using a TL431 at the secondary side
and an optocoupler to transfer output voltage information to the primary side through the isolation barrier) the
E/A can be used as an inverting level-shifter to achieve negative feedback and shape the loop gain (see "Application Ideas", fig. 23).
The E/A output is used typically for control loop compensation, realized with an RC network connected to the
inverting input. With other secondary feedback techniques, the output is driven directly by an emitter-grounded
optocoupler to modulate the duty cycle (the inverting input will be grounded in that case - see figure 23 in "Application Ideas").
Current Comparator, PWM Latch and Hiccup-mode OCP (see fig. 17b):
The current comparator senses the voltage across the current sense resistor (Rs) and, by comparing it with the
programming signal delivered by the feedforward block, determines the exact time when the external MOSFET
is to be switched off. The PWM latch avoids spurious switching of the MOSFET, which might result from the
noise generated ("double-pulse suppression").
A comparator senses the voltage on the current sense input and disables the gate driver if the voltage at the pin
exceeds 2 V. Such anomalous condition is typically generated by a short circuit on the secondary rectifier or on
the secondary winding. To re-enable the driver, first the IC must be turned off and then can be restarted, that is
the Vcc voltage must fall below the UVLO threshold.
When the gate driver is disabled the quiescent current of the IC is unchanged and, since no energy is coming
from the self-supply circuit, the Vcc capacitor will be discharged below the UVLO threshold after some time.
Then the device will initiate a new start-up cycle. In case of failure of the secondary diode the resulting behavior
will be a low-frequency intermittent operation (Hiccup-mode operation), with very low stress on the power circuit.
Gate Driver (see fig. 18):
A totem pole buffer, with 400mA source and sink capability, drives the external MOSFET. It is made up of a
high-side NPN Darlington and a low-side MOSFET. In this way there is no need of an external diode clamp to
prevent the voltage at the gate drive output (pin 7, GD) from being pulled too negative.
An internal pull-down circuit holds the output low when the device is in UVLO conditions, to ensure that the external MOSFET cannot be turned on accidentally (e.g. at power-on).
Figure 18. Gate driver with UVLO pull-down
Vcc
8
L6565
7
GD
Q
DRIVER
UVLO
6
GND
11/17
L6565
TYPICAL APPLICATIONS
Figure 19. 50W Wide Range Mains SMPS for 14" TV
C6 4700pF/ 4KV
F1
2A fuse
NTC1
16R
D1
1N4148
C25
C24 1nF
100nF
Vin
C23
88 to
264 Vac 100nF
R9
4.7M
3
B1 2KBP04M
L1
15mH
C1
150 µF
400 V
C26
1nF
R1
75k
R2
75k
R10
4.7M
C22
100 pF
T1
105 V
0.35 A
C2
C8
8.2 nF 180 pF
250 V 630V
R5
100 k
C7 4700pF/ 4KV
N1
C9
220 µF
D5
BYT01-400
8
160 V
1
D3
N2
STTA106
9
R8 22
D6
BYW98-100
4
N3
C4
47µF
25V
5
7
R3
3M
VFF
3
8
2
C3
1 nF
4
1
COMP
R4
16 K
10
5
Q1
STP7NB80FI
D2
R7 10 1N4148
CS
Vcc
6
INV
Naux
R6 100
GD
IC1
L6565
25 V
D4
1N4148
R20 22 k
ZCD
R11
0.47
C27
220nF
GND
1
IC3 PC817
R12
47 k
C12
100 µF 25V
R15
1.8 k
1
IC4
L7805
2
3
4
R14
1.5 k
DZ1
15 V
R13
3.3 k
C5
2.2 nF
14 V
1A
C10
470 µF
3
TRANSFORMER SPECS:
CORE: ETD29x16x10, N67 material or equivalent
IC2 TL431
≈ 1 mm air gap for a primary inductance of 285 µH
N1: 48 T (24T+24T series connected), 2xAWG28 (∅ 0.37 mm)
N2: 31 T, AWG28
N3: 5 T, AWG28
Naux: 5 T, AWG32 (∅ 0.24 mm)
P1
100 k
R16
220 k
2
C13
100 nF
1
3
R18
150 k
R17
4.7 k
2
Figure 20. 40W Wide Range Mains SMPS for inkjet printer
2200pF 4KV
2A fuse
16R
1N4148
BYW100-200
2KBP04M
1nF
Vin
88 to
100nF
264 Vac
100nF
28V / 0.7A
75 kΩ
56 kΩ
2W
75 kΩ
15mH
1nF
10 nF
250V
N2
2 x 470µF
35V
BYW98-100
12V / 1.5A
N1
STTA106
N3
2 x 1000µF
16V
GND
BYW100-50
10 Ω
47 kΩ
47µF
5V / 0.5A
1N4148
N5
5
3 MΩ
N4
470µF
16V
8
10 Ω
STP4NA80FP
7
L6565
220 Ω
3
10 nF
PC817A
4
16 kΩ
2
1
0.39 Ω
1/2 W
3.3 nF
PC817
100 nF
TL431
TRANSFORMER SPECS:
CORE: ETD29x16x10, 3C85 material or equivalent
≈1 mm air gap for a primary inductance of 700 µH
N1: 75 T, AWG25 (∅ 0.51 mm)
N2: 8 T, AWG25
N3: 7 T, AWG20 (∅ 0.89 mm)
N4: 3 T, AWG25
N5: 7 T, AWG32 (∅ 0.24 mm)
12/17
270 kΩ
3.9 kΩ
6
2.7 kΩ
5.1 kΩ
+5 V
50 mA
C11
47 µF
25V
L6565
APPLICATION IDEAS
Here follows a series of ideas/suggestions aimed at either improving performance or solving common application issues of L6565-based power supplies.
Figure 21. Enhanced turn-off for big MOSFET's drive
Vcc
8
7
GD
Q
DRIVER
BC327
L6565
Rs
6
GND
Figure 22. Latched shutdown on: a) feedback disconnection; b) overload or short circuit
Vcc
Vcc
8
8
L6565
L6565
2
2
COMP
COMP
BC327
1N4148
BC327
1N4148
BC337
BC337
a)
b)
Figure 23. Secondary Feedback loop configurations
Vout
Vout
Vout
L6565
2
1
Vcc
8
INV
COMP
L6565
2
COMP
1
RA
INV
8
RB
L6565
ICOMP
TL431
Vcc
TL431
1
TL431
4
CS
INV
a)
b)
Roff
Rs
c)
13/17
L6565
Figure 24. Primary Feedback loop configurations
Vcc
COMP
Vcc
COMP
8
2
RH
8
2
RH
1
1
-
INV
+
-
to VFF
block
E/A
INV
to VFF
block
E/A
+
RL
RL
2.5V
GND
2.5V
GND
L6565
6
a)
L6565
6
b)
Figure 25. Protection against secondary
feedback disconnection by primary
regulation take-over
Figure 27. Remote ON/OFF control
L6565
5
Vcc
15 V
ZCD
8
INV
1
L6565
OFF
2
BC337
ON
COMP
2.2 kΩ
Figure 28. Low-consumption start-up circuit
Figure 26. Leading edge blanking circuit for
enhanced primary regulation
BC327
+Vin
Vac
1N4148
Vcc
R
C START 1N4148
Vcc
470 pF
L6565
8
1N4148
CSUPPLY
L6565
2.7 kΩ
6
GND
GND
RELATED DOCUMENTATION
[1] "L6565, QUASI-RESONANT CONTROLLER” (AN1326)
[2] “25W QUASI-RESONANT FLYBACK CONVERTER FOR SET-TOP BOX APPLICATIONS USING THE
L6565” (AN1376)
[3] “EVAL6565N, 30W AC-DC ADAPTER WITH THE L6565 QUASI-RESONANT PWM CONTROLLER”
(AN1439).
14/17
L6565
mm
inch
DIM.
MIN.
A
TYP.
MAX.
MIN.
3.32
TYP.
MAX.
0.131
a1
0.51
B
1.15
1.65
0.045
0.065
b
0.356
0.55
0.014
0.022
b1
0.204
0.304
0.008
0.012
0.020
D
E
10.92
7.95
9.75
0.430
0.313
0.384
e
2.54
0.100
e3
7.62
0.300
e4
7.62
0.300
F
6.6
0.260
I
5.08
0.200
L
Z
3.18
OUTLINE AND
MECHANICAL DATA
3.81
1.52
0.125
0.150
Minidip
0.060
15/17
L6565
mm
DIM.
MIN.
TYP.
A
a1
inch
MAX.
MIN.
TYP.
1.75
0.1
0.25
a2
MAX.
0.069
0.004
0.010
1.65
0.065
a3
0.65
0.85
0.026
0.033
b
0.35
0.48
0.014
0.019
b1
0.19
0.25
0.007
0.010
C
0.25
0.5
0.010
0.020
c1
45° (typ.)
D (1)
4.8
5.0
0.189
0.197
E
5.8
6.2
0.228
0.244
e
1.27
e3
0.050
3.81
0.150
F (1)
3.8
4.0
0.15
0.157
L
0.4
1.27
0.016
0.050
M
S
0.6
0.024
8 ° (max.)
(1) D and F do not include mold flash or protrusions. Mold flash or
potrusions shall not exceed 0.15mm (.006inch).
16/17
OUTLINE AND
MECHANICAL DATA
SO8
L6565
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of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted
by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject
to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not
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17/17