L6739
Single-phase PWM controller with light-load efficiency optimization
Datasheet - production data
Applications
Memory and termination supply
Subsystem power supply (MCH, IOCH, PCI)
CPU and DSP power supply
Distributed power supply
General DC-DC converter
Description
VFQFPN16
Features
Flexible power supply from 5 V to 12 V bus
Power conversion input as low as 1.5 V
Light-load efficiency optimization
Embedded bootstrap switch
0.8 V internal reference
0.5% output voltage accuracy
Remote GND recovery
High current integrated drivers
Sensorless and programmable precise-OC
sense across inductor DCR
Overtemperature protection
Programmable oscillator up to 600 kHz
LS-less to manage pre-bias startup
Adjustable output voltage
The device flexibility allows to manage
conversions with power input VIN as low as 1.5 V
and the device supply voltage ranging from 5 V to
12 V bus.
The L6739 device features a proprietary algorithm
that allows light-load efficiency optimization,
boosting efficiency without compromising the
output voltage ripple.
VIN detector
OV protection
The L6739 is a single-phase step-down controller
with integrated high current drivers that provides
complete control logic and protection to realize
a DC-DC converter.
The integrated 0.8 V reference allows generation
of output voltages with ± 0.5% accuracy over line
and temperature variations.
The oscillator is programmable up to 600 kHz.
The L6739 provides a programmable overcurrent
protection and overvoltage protection. The current
information is monitored across the inductor DCR.
The L6739 device is available in a VFQFPN 16 3 x 3 mm package.
Disable function
Internal soft-start
VFQFPN 16 - 3 x 3 mm package
May 2014
This is information on a product in full production.
Table 1. Device summary
Order code
Package
Packing
L6739TR
VFQFPN16
Tape and reel
DocID026384 Rev 1
1/30
www.st.com
Contents
L6739
Contents
1
2
3
Typical application circuit and block diagram . . . . . . . . . . . . . . . . . . . . 4
1.1
Application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
1.2
Block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Connection diagram and pin description . . . . . . . . . . . . . . . . . . . . . . . . 6
2.1
Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
2.2
Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Electrical specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
3.1
Absolute maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
3.2
Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
4
Device description and operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
5
Soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
LS-less startup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
6
Output voltage setting and protections . . . . . . . . . . . . . . . . . . . . . . . . 13
Overcurrent . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Overcurrent threshold setting. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
7
Main oscillator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
8
High current embedded drivers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
9
2/30
8.1
Boot capacitor design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
8.2
Power dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Application details . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
9.1
Compensation network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
9.2
Layout guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
DocID026384 Rev 1
L6739
10
Contents
Application information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
10.1
Inductor design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
10.2
Output capacitor(s) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
10.3
Input capacitors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
11
Package information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
12
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
DocID026384 Rev 1
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Typical application circuit and block diagram
L6739
1
Typical application circuit and block diagram
1.1
Application circuit
Figure 1. Typical 3.3 V application circuit (LDO disabled)
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DocID026384 Rev 1
L6739
1.2
Typical application circuit and block diagram
Block diagram
Figure 3. Block diagram
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16
GND
All internal references, logic and driver return path are referenced to this pin. Connect to
the PCB GND ground plane and filter to VCC and VCCDR.
Thermal
PAD
The thermal pad connects the silicon substrate and makes good thermal contact with the
PCB. Use VIAs to connect to the PGND plane.
2.2
Function
Error amplifier output.
Connect with an RF - CF to FB. The device cannot be disabled by grounding this pin.
Error amplifier inverting input.
Connect with a resistor RFB to VSEN and with an RF - CF to COMP.
Output voltage monitor.
It manages OVP and UVP protections and PGOOD. Connect to the positive side of the
load for remote sensing. See Section 6 on page 13 for details.
Thermal data
Table 3. Thermal data
Symbol
Parameter
Value
Unit
RTHJA
Thermal resistance junction to ambient
(device soldered on 2s2p PC board)
45
°C/W
RTHJC
Thermal resistance junction to case
1
°C/W
TMAX
Maximum junction temperature
150
°C
TSTG
Storage temperature range
-40 to 150
°C
TJ
Junction temperature range
-40 to 125
°C
DocID026384 Rev 1
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Electrical specifications
L6739
3
Electrical specifications
3.1
Absolute maximum ratings
Table 4. Absolute maximum ratings
Symbol
Condition
Value
Unit
VCC
To GND
-0.3 to 15
V
VCCDR
To GND(1)
-0.3 to 6
V
VBOOT, VUGATE
To GND
To PHASE
-0.3 to 41
-0.3 to 6
V
VPHASE
To GND
-5 to 35
V
VLGATE
To GND(2)
-0.3 to 6
V
(1)
-0.3 to 7
V
-0.3 to 8
V
-0.3 to 3.6
V
EN, PGOOD
To GND
CSP, CSN
To
FB, COMP, VSEN, OSC, FBG
GND(3), (4)
To GND
1. Needs to be lower than VCC under any condition.
2. Needs to be lower than VCCDR under any condition.
3. Needs to be lower than VCC - 1.5 V under any condition.
4. Max. differential voltage to be limited within 100 mV.
Table 5. Recommended operative conditions
Symbol
Condition
VCC
VOUT (max.)
(1)
Value
Unit
4.75 to 13.2
V
7
V
1. VCC needs to be > 1.5 V higher than CSx pins to ensure proper operation of current sense.
Note:
8/30
Absolute maximum ratings are those values beyond which damage to the device may occur.
These are stress ratings only and functional operation of the device at these conditions is
not implied. Operating outside maximum recommended conditions for extended periods of
time may impact product reliability and results in device failures.
DocID026384 Rev 1
L6739
3.2
Electrical specifications
Electrical characteristics
Table 6. Electrical characteristics
(VCC = 12 V; VCCDR = open; TA = 25 C unless otherwise specified)
Symbol
Parameter
Test conditions
Min.
Typ.
Max.
Unit
Supply current and power-on
ICC
ICCDR
UVLOVCC
UVLOVCCDR
LDO
VCC supply current
VCCDR supply current
Turn-ON threshold
EN = HIGH
7.5
mA
EN = GND
6.0
mA
EN = HIGH, VCCDR = VCC = 5 V
5.5
mA
EN = GND, VCCDR = VCC = 5 V
4
mA
VCC = VCCDR = 5 V,
UGATE and LGATE = open,
EN = HIGH, UGATE ON,
LGATE OFF
0.4
mA
VCC = VCCDR = 5 V,
UGATE and LGATE = open,
EN = HIGH, UGATE OFF,
LGATE ON
0.4
mA
VCC = VCCDR = 5 V,
EN = GND
UGATE and LGATE = open
0.2
mA
VCC rising
4.1
Hysteresis
Turn-ON threshold
0.2
VCCDR rising
V
V
3.9
V
Hysteresis
0.15
V
Voltage
5.75
V
6.1
V
3.5
V
Turn-ON threshold
Turn-OFF threshold
Measured at the VCC pin
Oscillator enable and soft-start
FSW
Main oscillator accuracy
OSC = open
kOSC
Oscillator gain
Current sink/source from OSC
tSS
Soft-start time
OSC = open
4.5
5.12
5.7
msec
SS delay
OSC = open, before SS
4.5
5.12
5.7
msec
tSSdelay
VOSC
d
180
200
220
10
PWM ramp amplitude
kHz/A
2
Duty cycle
0
kHz
V
100
%
ENABLE
EN
Input logic high
EN rising
Input logic low
EN falling
DocID026384 Rev 1
1.1
V
0.5
V
9/30
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Electrical specifications
L6739
Table 6. Electrical characteristics (continued)
(VCC = 12 V; VCCDR = open; TA = 25 C unless otherwise specified)
Symbol
Parameter
Test conditions
Min.
Typ.
Max.
Unit
-0.5
-
0.5
%
Reference and error amplifier
Output voltage accuracy
A0
120
dB
15
MHz
Slew rate
8
V/s
RBOOT
Boot switch resistance
11
IUGATE
HS Source current(1)
BOOT - PHASE = 5.5 V;
CUGATE to PHASE = 3.3 nF
2
A
RUGATE
HS sink resistance
BOOT - PHASE = 5.5 V; 100 mA
ILGATE
LS source current(1)
VCCDR = 5.5 V;
CLGATE to GND = 5.6 nF
2
RLGATE
LS sink resistance
VCCDR = 5.5 V; 100 mA
0.5
1
17
20
23
mV
VSEN rising
0.890
0.920
0.950
V
Un-latch, VSEN falling
0.350
0.400
0.450
V
VSEN falling
0.570
0.600
0.630
V
GBWP
SR
DC gain
Vout to FBG
(1)
(1)
Gain bandwidth product
(1)
Gate driver
1.0
1.5
A
Current sense amplifier
VOCTH
OC current threshold
CSP - CSN; 7x masking
PGOOD and protection
PGOOD
OVP threshold
UVP threshold
Overtemperature protection
OTP
Thermal shutdown
threshold(1)
140
°C
Thermal shutdown
hysteresis(1)
40
°C
1. Guaranteed by design, not subject to test.
10/30
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L6739
4
Device description and operation
Device description and operation
The L6739 device is a single-phase PWM controller with embedded high current drivers that
provides complete control logic and protections to realize a general DC-DC step-down
converter. Designed to drive N-channel MOSFETs in a synchronous buck topology, with its
high level of integration, this 16-pin device allows a reduction of cost and size of the power
supply solution and also provides real-time PGOOD in a compact VFQFPN16 - 3 x 3 mm.
The L6739 is designed to operate from a 5 V or 12 V supply. The output voltage can be
precisely regulated to as low as 0.8 V with ± 0.5% accuracy over line and temperature
variations. The controller performs remote GND recovery to prevent losses and GND drops
to affect the regulation.
The switching frequency is internally set to 200 kHz and adjustable through the OSC pin.
The IC can be disabled by pulling the OSC pin low.
The L6739 device provides a simple control loop with a voltage-mode error amplifier. The
error amplifier features a 15 MHz gain bandwidth product and 8 V/µs slew rate, allowing
high regulator bandwidth for fast transient response.
To avoid load damages, the L6739 provides overcurrent protection, and overvoltage and
undervoltage protection. The overcurrent trip threshold is monitored through the inductor
DCR, assuring optimum precision, saving the use of an expensive and space consuming
sense resistor. The output voltage is monitored through the dedicated VSEN pin.
The L6739 implements soft-start by increasing the internal reference in closed loop
regulation. The low-side-less feature allows the device to perform the soft-start over prebiased output avoiding high current return through the output inductor and dangerous
negative spikes at the load side.
The L6739 device is available in a compact VFQFN16 - 3 x 3 mm package with an exposed
pad.
DocID026384 Rev 1
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Soft-start
5
L6739
Soft-start
The L6739 device implements a soft-start to smoothly charge the output filter avoiding high
inrush currents to be required to the input power supply. During this phase, the device
increases the internal reference from zero up to 0.8 V in closed loop regulation. The softstart is implemented only when VCC and VCCDR are above their own UVLO threshold and
the EN pin is set high.
When SS takes place, the IC initially waits for 1024 clock cycles and then starts ramping up
the reference in 1024 clock cycles in closed loop regulation. At the end of the digital softstart, the PGOOD signal is set free with 3x clock cycles delay.
Protections are active during this phase as follows:
Undervoltage is enabled when the reference voltage reaches 80% of the final value.
Overvoltage is always enabled.
FB disconnection is enabled.
Overtemperature protection is enabled.
Soft-start time depends on the programmed frequency, initial delay and reference ramp up
lasts for 1024 clock cycles. SS time and initial delay can be determined as follows:
Equation 1
1024
T SS ms = -----------------------------Fsw kHz
LS-less startup
In order to avoid any kind of negative undershoot on the load side during startup, the L6739
device performs a special sequence in enabling the drivers for both sections: during the
soft-start phase, the LS MOSFET is kept OFF until the first PWM pulse. This particular
sequence avoids the dangerous negative spike on the output voltage that can happen if
starting over a pre-biased output.
Low-side MOSFET turn-on is masked only from the control loop point of view: protections
are still allowed to turn-on the low-side MOSFET in the case of overvoltage, if needed.
Figure 5. LS-less startup (left) vs. non-LS-less startup (right)
12/30
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L6739
6
Output voltage setting and protections
Output voltage setting and protections
The L6739 device is capable of precisely regulating an output voltage as low as 0.8 V. In
fact, the device comes with a fixed 0.8 V internal reference that guarantees the output
regulated voltage to be within ± 0.5% tolerance over line and temperature variations
(excluding output resistor divider tolerance, when present).
Output voltage higher than 0.8 V can be easily achieved by adding a resistor ROS between
the FB pin and ground. Referring to Figure 1 on page 4, the steady state DC output voltage
is:
Equation 2
R FB
V OUT = V REF 1 + -----------
R OS
where VREF is 0.8 V.
The L6739 monitors the voltage at the VSEN pin and compares it to the internal reference
voltage in order to provide undervoltage and overvoltage protections, as well as PGOOD
signal. According to the level of VSEN, different actions are performed from the controller:
PGOOD
If the voltage monitored through VSEN exits from the PGOOD window limits, the device
de-asserts the PGOOD signal. PGOOD is asserted at the end of the soft-start phase
with 3x clock cycles delay.
Undervoltage protection (UV)
If the voltage at the VSEN pin drops below the UV threshold, the device turns off both
HS and LS MOSFETs, latching the condition. Cycle VCC or EN to recover.
UV is also active during SS acting as VIN detection protection. See the description
below.
Overvoltage protection (OV)
If the voltage at the VSEN pin rises over the OV threshold, overvoltage protection turns
off the HS MOSFET and turns on the LS MOSFET. The LS MOSFET is turned off as
soon as VSEN goes below Vref/2. The condition is latched, cycle VCC/EN to recover.
Note that, even if the device is latched, the device still controls the LS MOSFET and
can switch it on whenever VSEN rises above the OV threshold.
PreOVP protection
Monitors VSEN when IC is disabled. If VSEN surpasses the OV threshold, IC turns on
the low-side MOSFET to protect the load. On the EN rising edge, the protection is
disabled and the IC implements the SS procedure. PreOVP is disabled when EN is
high but the OV protection becomes operative.
VIN detection
UV protection active during SS allows the IC to detect whether input voltage VIN is
present. If UV is triggered during the soft-start, it resets the SS procedure: the controller
re-implements the initial delay and re-ramps up the reference with the same SS timings
described in Section 5. The UV protection is then preventing that IC starts-up when VIN
is not present.
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30
Output voltage setting and protections
L6739
Overtemperature protection (OT)
To avoid any damage to the device when reaching high temperature, when the junction
temperature reaches 140 °C, the device turns off both MOSFETs. If the junction
temperature drops below 100 °C, the device restarts with a new soft-start sequence.
Protections are active also during soft-start (See Section 5).
For proper operations, VCC needs to be at least 1.5 V higher than the programmed output
voltage.
Table 7. L6739 Protection at a glance
L6739
Comments
Overvoltage (OV)
VSEN = +15% above reference.
Action: IC latch; LS = ON until VSEN = 50% of Vref; PGOOD = GND.
Action (EN = 0): IC latch; LS = ON; reset by EN rising edge (PreOVP).
Undervoltage (UV)
VSEN = -25% below reference.
Action: IC latch; HiZ; PGOOD = GND.
Action (SS): SS reset (VIN detection).
PGOOD
Overcurrent (OC)
PGOOD is set to zero whenever VSEN falls outside the +15% / -25% of Vref.
Action: PGOOD transition coincides with OV/UV protection set.
Current monitor across inductor DCR.
Action: 1st threshold (20 mV): IC latch after 7 consecutive constant current events.
TJ > 140 °C.
Overtemperature (OT) Action: the device shuts down and restarts with a new soft-start sequence when the TJ
drops to 100 °C.
Overcurrent
The overcurrent function protects the converter from a shorted output or overload, by
sensing the output current information across the inductor DCR. This method reduces cost
and enhances converter efficiency by avoiding the use of expensive and space consuming
sense resistors.
The inductor DCR current sense is implemented by comparing and monitoring the
difference between the CSP and CSN pins. If the monitored voltage is bigger than the
internal thresholds, an overcurrent event is detected.
DCR current sensing requires time constant matching between the inductor and the reading
network:
Equation 3
L
------------- = R C
DCR
V CSP-CSN = DCR I OUT
The L6739 device monitors the voltage between CSP and CSN, when this voltage exceeds
the OC threshold, an overcurrent is detected. The IC works in constant current mode,
turning on the low-side MOSFET immediately while the OC persists and, in any case, until
the next clock cycle. After seven consecutive OC events, overcurrent protection is triggered
and the IC latches.
14/30
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L6739
Output voltage setting and protections
When overcurrent protection is triggered, the device turns off both LS and HS MOSFETs in
a latched condition.
To recover from an overcurrent protection triggered condition, VCC power supply or EN
must be cycled.
For proper current reading, the CSN pin must be filtered by 100 nF (typ.) MLCC to GND.
Note:
The device features an integrated OC thermal compensation in order to reduce the
influence of the temperature change of the inductor DCR. The internal temperature
coefficient compensation is 4000 ppm/°C (0.4%/°C).
Overcurrent threshold setting
The L6739 device detects OC when the difference between CSP and CSN is equal to
20 mV (typ.). By properly designing the current reading network, it is possible to program
the OC threshold as desired (See Figure 6).
Equation 4
20mV R1 + R2
I OCP = ---------------- ---------------------DCR
R2
Time constant matching is, in this case, designed considering:
Equation 5
L
------------- = R1//R2 C
DCR
This means that once inductor has been chosen, the two conditions above define the proper
values for R1 and R2.
Figure 6. Current reading network
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Main oscillator
7
L6739
Main oscillator
The controller embeds a programmable oscillator. The internal oscillator generates the
sawtooth waveform for the PWM charging with a constant current and resetting an internal
capacitor. The switching frequency, FSW, is internally fixed at 200 kHz.
The current delivered to the oscillator is typically 20 A (corresponding to the free running
frequency FSW = 200 kHz) and it may be varied using an external resistor (ROSC) typically
connected between the OSC pin and GND. As the OSC pin is fixed at 1.240 V, the
frequency is varied proportionally to the current sunk from the pin considering the internal
gain of 10 kHz/A (see Figure 7).
Connecting ROSC to GND, the frequency is increased (current is sunk from the pin),
according to the following relationships:
Equation 6
1.240V
kHz
F SW = 200kHz + ------------------- 10 ----------R OSC
A
Connecting ROSC to a positive voltage, the frequency is reduced (current is forced into the
pin), according to the following relationships:
Equation 7
V j – 1.240
kHz
F SW = 200kHz – -------------------------- 10 -----------
R OSC
A
where Vj is the positive voltage to which the ROSC resistor is connected.
Figure 7. ROSC vs. switching frequency
16/30
DocID026384 Rev 1
L6739
High current embedded drivers
8
High current embedded drivers
The L6739 device provides high current driving control. The driver for the high-side
MOSFET uses the BOOT pin for supply and the PHASE pin for return. The driver for the
low-side MOSFET uses the VCCDR voltage for supply and the GND pin for return. An
embedded linear regulator is provided to guarantee proper supply voltage to the high-side
and low-side drivers which are optimized for such regulated voltage to guarantee minimum
deadtimes and higher efficiency. The LDO can be bypassed, in case of low 5 V VCC
applications, by properly shorting VCCDR to VCC. In this case the LDO automatically
disables reducing consumptions.
The embedded driver embodies an anti shoot-through and adaptive deadtime control to
minimize the low-side body diode conduction time maintaining good efficiency and saving
the use of Schottky diodes: when the high-side MOSFET turns off, the voltage on its source
begins to fall; when the voltage reaches about 2 V, the low-side MOSFET gate drive voltage
is suddenly applied. When the low-side MOSFET turns off, the voltage at the LGATE pin is
sensed. When it drops below about 1 V, the high-side MOSFET gate drive voltage is
suddenly applied. If the current flowing in the inductor is negative, the source of the highside MOSFET never drops. To allow the low-side MOSFET to turn-on even in this case,
a watchdog controller is enabled: if the source of the high-side MOSFET doesn't drop, the
low-side MOSFET is switched on, so allowing the negative current of the inductor to
recirculate. This mechanism allows the system to regulate even if the current is negative.
8.1
Boot capacitor design
The bootstrap capacitor needs to be designed in order to show a negligible discharge due to
the high-side MOSFET turn-on. In fact, it must give a stable voltage supply to the high-side
driver during the MOSFET turn-on, also minimizing the power dissipated by the embedded
boot switch. Figure 8 gives some guidelines on how to select the capacitance value for the
bootstrap according to the desired discharge and depending on the selected MOSFET.
Figure 8. Bootstrap capacitor design
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High current embedded drivers
8.2
L6739
Power dissipation
It is important to consider the power that the device is going to dissipate in driving the
external MOSFETs in order to avoid surpassing the maximum junction operative
temperature.
Two main terms contribute in the device power dissipation: bias power and drivers' power.
Device power (PDC) depends on the static consumption of the device through the
supply pins and it is simply quantifiable as follows:
Equation 8
P DC = V CC I CC + V VCCDR I VCCDR
Driver's power is the power needed by the driver to continuously switch ON and OFF
the external MOSFETs; it is a function of the switching frequency and total gate charge
of the selected MOSFETs. It can be quantified considering that the total power PSW,
dissipated to switch the MOSFETs, is dissipated by three main factors: external gate
resistance (when present), intrinsic MOSFET resistance and intrinsic driver resistance.
This last term is the important one to be determined to calculate the device power
dissipation.
The total power dissipated to switch the MOSFETs for each phase featuring embedded
driver results:
Equation 9
P SW = F SW Q GHS VCCDR + Q GLS VCCDR
where QGHS is the total gate charge of the HS MOSFETs and QGLS is the total gate
charge of the LS MOSFETs.
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L6739
Application details
9
Application details
9.1
Compensation network
The control loop shown in Figure 9 is a voltage-mode control loop. The output voltage is
regulated to the internal reference (when present, an offset resistor between FB node and
GND can be neglected in control loop calculation).
Error amplifier output is compared to the oscillator sawtooth waveform to provide a PWM
signal to the driver section. The PWM signal is then transferred to the switching node with
VIN amplitude. This waveform is filtered by the output filter.
The converter transfer function is the small signal transfer function between the output of the
EA and VOUT. This function has a double pole at frequency FLC depending on the L-COUT
resonance and a zero at FESR depending on the output capacitor ESR. The DC gain of the
modulator is simply the input voltage VIN divided by the peak-to-peak oscillator voltage
VOSC.
Figure 9. PWM control loop
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The compensation network closes the loop joining VOUT and EA output with transfer
function ideally equal to -ZF/ZFB.
Compensation goal is to close to the control loop assuring high DC regulation accuracy,
good dynamic performance and stability. To achieve this, the overall loop needs high DC
gain, high bandwidth and good phase margin.
High DC gain is achieved giving an integrator shape to compensation network transfer
function. Loop bandwidth (F0dB) can be fixed choosing the right RF/RFB ratio, however, for
stability, it should not exceed FSW/2. To achieve a good phase margin, the control loop gain
has to cross the 0 dB axis with -20 dB/decade slope.
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Application details
L6739
For example, Figure 10 shows an asymptotic bode plot of a type III compensation.
Figure 10. Example of type III compensation
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Open loop converter singularities:
Equation 10
a)
1
F LC = ---------------------------------2 L C OUT
b)
1
F ESR = -------------------------------------------2 C OUT ESR
Compensation network singularities frequencies:
Equation 11
20/30
a)
1
F Z1 = -----------------------------2 R F C F
b)
1
F Z2 = ----------------------------------------------------2 R FB + R S C S
c)
1
F P1 = -------------------------------------------------CF CP
2 R F ---------------------
CF + CP
d)
1
F P2 = ------------------------------2 R S C S
DocID026384 Rev 1
L6739
Application details
To place the poles and zeros of the compensation network, the following suggestions may
be followed:
a)
Set the gain RF / RFB in order to obtain the desired closed loop regulator
bandwidth according to the approximated formula (suggested values for RFB are
in the range of some k):
Equation 12
RF
F 0dB V OSC
---------- = ------------ ------------------R FB
F LC
V IN
b)
Place FZ1 below FLC (typically 0.5 * FLC):
1
C F = ---------------------------- R F F LC
c)
Place FP1 at FESR:
CF
C P = ---------------------------------------------------------2 R F C F F ESR – 1
d)
Place FZ2 at FLC and FP2 at half of the switching frequency:
R FB
R S = --------------------------F SW
------------------ – 1
2 F LC
1
C S = ------------------------------ R S F SW
9.2
e)
Check that compensation network gain is lower than open loop EA gain before
F0dB.
f)
Check phase margin obtained (it should be greater than 45°) and repeat if
necessary.
Layout guidelines
The L6739 device provides control functions and high current integrated drivers to
implement high current step-down DC-DC converters. In this kind of application, a good
layout is very important.
The first priority when placing components for these applications has to be reserved to the
power section, minimizing the length of each connection and loop as much as possible. To
minimize noise and voltage spikes (EMI and losses) power connections (highlighted in
Figure 11) must be a part of a power plane and realized by wide and thick copper traces:
loop must be minimized. The critical components, i.e. the power MOSFETs, must be close
to one another. The use of a multi-layer printed circuit board is recommended.
The input capacitance (CIN), or at least a portion of the total capacitance needed, has to be
placed close to the power section in order to eliminate the stray inductance generated by
the copper traces. Low ESR and ESL capacitors are preferred, MLCCs are recommended
to be connected near the HS drain.
Use a proper number of VIAs when power traces have to move between different planes on
the PCB in order to reduce both parasitic resistance and inductance. Moreover, reproducing
DocID026384 Rev 1
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Application details
L6739
the same high current trace on more than one PCB layer reduces the parasitic resistance
associated to that connection.
Connect output bulk capacitors (COUT) as near as possible to the load, minimizing parasitic
inductance and resistance associated to the copper trace, also adding extra decoupling
capacitors along the way to the load when this results in being far from the bulk capacitors
bank.
Remote sense connection must be routed as parallel nets from the FBG/VSEN pins to the
load in order to avoid the pickup of any common mode noise. Connecting these pins in
points far from the load causes a non-optimum load regulation, increasing output tolerance.
Locate current reading components close to the device. The PCB traces connecting the
reading point must use dedicated nets, routed as parallel traces in order to avoid the pickup
of any common mode noise. It's also important, to avoid any offset in the measurement and,
to get a better precision, to connect the traces as close as possible to the sensing elements.
A small filtering capacitor can be added, near the controller, between VOUT and GND, on the
CSN line to allow higher layout flexibility.
Figure 11. Power connections (heavy lines)
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Gate traces and phase trace must be sized according to the driver RMS current delivered to
the power MOSFET. The device robustness allows the managing of applications with the
power section far from the controller without losing performance. However, when possible, it
is recommended to minimize the distance between the controller and power section.
Small signal components and connections to critical nodes of the application, as well as
bypass capacitors for the device supply, are also important. Locate the bypass capacitor
(VCC and bootstrap capacitor) and feedback compensation components as close to the
device as practical.
22/30
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L6739
Application details
Figure 12. Drivers turn-on and turn-off paths
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Application information
L6739
10
Application information
10.1
Inductor design
The inductance value is defined by a compromise between the dynamic response time, the
efficiency, the cost, and the size. The inductor must be calculated to maintain the ripple
current (IL) between 20% and 30% of the maximum output current (typ.). The inductance
value can be calculated with the following relationship:
Equation 13
V IN – V OUT V OUT
L = ------------------------------ -------------F SW I L
V IN
where FSW is the switching frequency, VIN is the input voltage and VOUT is the output
voltage. Figure 13 shows the ripple current vs. the output voltage for different values of the
inductor, with VIN = 5 V and VIN = 12 V.
Increasing the value of the inductance reduces the current ripple but, at the same time,
increases the converter response time to a dynamic load change. The response time is the
time required by the inductor to change its current from initial to final value. Until the inductor
has finished its charging time, the output current is supplied by the output capacitors.
Minimizing the response time can minimize the output capacitance required. If the
compensation network is well designed, during a load variation the device is able to set
a duty cycle value very different (0% or 80%) from the steady state one. When this condition
is reached, the response time is limited by the time required to change the inductor current.
Figure 13. Inductor current ripple vs. output voltage
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L6739
10.2
Application information
Output capacitor(s)
The output capacitors are basic components to define the ripple voltage across the output
and for the fast transient response of the power supply. They depend on the output voltage
ripple requirements, as well as any output voltage deviation requirement during a load
transient.
During steady state conditions, the output voltage ripple is influenced by both the ESR and
capacitive value of the output capacitors as follows:
Equation 14
V OUT_ESR = I L ESR
Equation 15
1
V OUT_C = I L --------------------------------------8 C OUT F SW
where IL is the inductor current ripple. In particular, the expression that defines VOUT_C
takes into consideration the output capacitor charge and discharge as a consequence of the
inductor current ripple.
During a load variation, the output capacitors supply the current to the load or absorb the
current stored in the inductor until the converter reacts. In fact, even if the controller
immediately recognizes the load transient and sets the duty cycle at 80% or 0%, the current
slope is limited by the inductor value. The output voltage has a drop that, also in this case,
depends on the ESR and capacitive charge/discharge as follows:
Equation 16
V OUT_ESR = I OUT ESR
Equation 17
L I OUT
V OUT_C = I OUT -------------------------------------2 C OUT V L
where VL is the voltage applied to the inductor during the transient response
( D MAX VIN – VOUT for the load appliance or VOUT for the load removal).
MLCC capacitors have typically low ESR to minimize the ripple but also have low
capacitance which does not minimize the voltage deviation during dynamic load variations.
On the contrary, electrolytic capacitors have large capacitance to minimize voltage deviation
during load transients while they do not show the same ESR values as the MLCC, resulting
therefore in higher ripple voltages. For these reasons, a mix between electrolytic and MLCC
capacitors is suggested to minimize ripple as well as reduce voltage deviation in dynamic
mode.
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Application information
10.3
L6739
Input capacitors
The input capacitor bank is designed considering mainly the input RMS current that
depends on the output deliverable current (IOUT) and the duty cycle (D) for the regulation as
follows:
Equation 18
I rms = I OUT D 1 – D
The equation reaches its maximum value, IOUT / 2, with D = 0.5. The losses depend on the
input capacitor ESR and, in the worst case, are:
Equation 19
P = ESR I OUT 2
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11
Package information
Package information
In order to meet environmental requirements, ST offers these devices in different grades of
ECOPACK® packages, depending on their level of environmental compliance. ECOPACK
specifications, grade definitions and product status are available at: www.st.com.
ECOPACK is an ST trademark.
Figure 14. VFQFPN 16 - 3 x 3 mm package outline
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Package information
L6739
Table 8. VFQFPN 16 - 3 x 3 mm package mechanical data
Dimensions (mm)
Symbol
A
Min.
Typ.
Max.
0.80
0.90
1.00
A1
0.02
A3
0.20
b
0.18
0.25
0.30
D
2.85
3.00
3.15
E
2.85
3.00
3.15
D2
1.70
1.80
1.90
E2
1.70
1.80
1.90
e
L
0.50
0.45
0.50
Figure 15. Recommended footprint
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L6739
12
Revision history
Revision history
Table 9. Document revision history
Date
Revision
21-May-2014
1
Changes
Initial release.
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L6739
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