L7987L
61 V, 2 A asynchronous step-down switching regulator with
adjustable current limitation
Datasheet - production data
Description
HTSSOP16 (RTH = 40 °C/W)
Features
2 A DC output current
4.5 V to 61 V operating input voltage
RDS,ON = 300 mΩ typ.
Adjustable fSW (250 kHz - 1.5 MHz)
Low IQ-SHD (11 µA typ. from VIN)
Low IQ (1 mA typ. - VIN 24 V - VOUT 3.3 V)
Output voltage adjustable from 0.8 V to VIN
Synchronization
Adjustable soft-start time
The L7987L device is a step-down monolithic
switching regulator able to deliver up to 2 A DC.
The output voltage adjustability ranges from 0.8 V
to almost VIN. The embedded switchover feature
on the VBIAS pin maximizes the efficiency at light
load. The adjustable current limitation, designed
to select the inductor RMS current accordingly
with the nominal output current, and the high
switching frequency capability make the size of
the application compact. Pulse-by-pulse current
sensing with digital frequency foldback
implements an effective constant current
protection over the different application
conditions. The peak current foldback decreases
the stress of the power components in heavy
short-circuit condition. The PGOOD open
collector output can also implement output
voltage sequencing during the power-up phase.
Multiple devices can be synchronized sharing the
SYNCH pin to prevent beating noise in low noise
applications like sensors with A/D conversion.
Adjustable current limitation
Advanced bootstrap capacitor management for
LDO operation
VBIAS improves efficiency at light load
PGOOD open collector output
Output voltage sequencing
Digital frequency foldback in short-circuit
Peak current foldback in short-circuit
Auto-recovery thermal shutdown
Applications
Designed for 24 V bus
Fail safe tolerant systems
Programmable logic controllers (PLCs)
May 2020
This is information on a product in full production.
DocID026362 Rev 4
1/42
www.st.com
Contents
L7987L
Contents
1
Application schematic and block diagram . . . . . . . . . . . . . . . . . . . . . . . 4
2
Pin settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2.1
Pin connection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2.2
Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
2.3
Maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
2.4
Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
2.5
ESD protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
3
Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
4
Functional description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
5
4.1
Oscillator and synchronization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .11
4.2
Soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
4.3
Error amplifier and light-load management . . . . . . . . . . . . . . . . . . . . . . . 16
4.4
Low VIN operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
4.5
Overcurrent protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
4.6
Overtemperature protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Application information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
5.1
Input capacitor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
5.2
Output capacitor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
5.3
Inductor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
5.4
Compensation network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
5.4.1
Type II compensation network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
5.4.2
Type III compensation network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
5.5
Thermal considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
5.6
Layout considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
6
Demonstration board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
7
Application ideas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
7.1
2/42
Positive buck-boost . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
DocID026362 Rev 4
L7987L
Contents
7.2
8
Negative buck-boost . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
Package information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
8.1
HTSSOP16 package information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
9
Ordering information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
10
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
DocID026362 Rev 4
3/42
42
Application schematic and block diagram
1
L7987L
Application schematic and block diagram
Figure 1. Application schematic
Figure 2. Block diagram
VCC
VBIAS
VIN
BOOT
EN
EN
ENABLE
VREG
BOOT
CHARGE
UVLO
ISNS
REF and BIAS
VRAMP
+
FB
CURRENT
SENSING
E/A
FB
-
HS
DRIVER
- PWM
COMP
+
EN
SOFT
START
LOGIC
and
CONTROL
TEMP
MONITOR
LX
LS
DRIVER
ISS
- PG
SS
COMP
+
FB
PGOOD
VRAMP
OSCILLATOR
& RAMP
COMP
4/42
FSW
MASTER
SLAVE
SYNCH
DocID026362 Rev 4
ILIM
FUNCTION
ILIM
GND
L7987L
Pin settings
2
Pin settings
2.1
Pin connection
Figure 3. Pin connection (top view)
DocID026362 Rev 4
5/42
42
Pin settings
2.2
L7987L
Pin description
Table 1. Pin description
Number
Pin
Description
1
VBIAS
Auxiliary input that can be used to supply part of the analog circuitry to increase the efficiency
at light load. Typically connected to the regulated output voltage or to an external voltage rail
higher than 3 V. Connect to the signal GND if not used or bypass with a 1 F ceramic capacitor
if supplied by the output voltage or by an auxiliary rail.
2
VIN
DC input voltage
3
VIN
DC input voltage
4
VCC
Filtered DC input voltage to the internal circuitry. Bypass to the signal GND by a 1 F ceramic
capacitor.
5
EN
Active high enable pin. Connect to the VCC pin if not used.
6
SS
An internal current generator (5 µA typ.) charges the external capacitor to implement the softstart.
7
SYNCH Master / slave synchronization
8
COMP
9
FB
10
FSW
A pull-down resistor to GND selects the switching frequency.
11
ILIM
A pull-down resistor to GND selects the peak current limitation.
12
PGOOD
13
LX
Switching node
14
LX
Switching node
15
BOOT
Connect an external capacitor (100 nF typ.) between BOOT and LX pins. The gate charge
required to drive the internal n-DMOS is recovered by an internal regulator during the off-time
16
GND
Signal GND
-
E.P.
Exposed pad must be connected to signal GND.
6/42
Output of the error amplifier. The designed compensation network is connected at this pin.
Inverting input of the error amplifier.
The PGOOD open collector output is driven low when the output voltage, sensed on the FB
pin, is out of regulation.
DocID026362 Rev 4
L7987L
2.3
Pin settings
Maximum ratings
Table 2. Absolute maximum ratings
Symbol
Description
Min.
Max.
Unit
VIN
-0.3
61
V
VCC
-0.3
61
V
VBOOT - GND
-0.3
65
V
VBOOT - VLX
-0.3
4
V
VBIAS
-0.3
VCC
V
EN
-0.3
VCC
V
PGOOD
-0.3
VCC
V
LX
-0.3
VIN + 0.3
V
SYNCH
-0.3
5.5
V
SS
-0.3
3.6
V
FSW
-0.3
3.6
V
COMP
-0.3
3.6
V
ILIM
-0.3
3.6
V
FB
-0.3
3.6
V
Operating temperature range
-40
150
°C
TSTG
Storage temperature range
-65
150
°C
TLEAD
Lead temperature (soldering 10 sec.)
260
°C
2
A
BOOT
TJ
High-side RMS current
IHS
2.4
Thermal data
Table 3. Thermal data
2.5
Symbol
Parameter
Value
Unit
RthJA
Thermal resistance junction-ambient (device soldered
on the STMicroelectronics® demonstration board)
40
°C/W
Value
Unit
HBM
2
KV
CDM
500
V
ESD protection
Table 4. ESD protection
Symbol
ESD
Test condition
DocID026362 Rev 4
7/42
42
Electrical characteristics
3
L7987L
Electrical characteristics
All the population tested at TJ = 25 °C, VIN = VCC = 24 V and VEN = 3 V unless otherwise
specified.
Table 5. Electrical characteristics
Symbol
VIN
Parameter
Operating input voltage
range
RDSON HS High-side RDSON
Switching frequency
fSW
IPK
ISKIP
VFOLD
Test condition
Min.
(1)
4.5
V
0.30
0.48
0.30
0.57
233
250
267
kHz
225
250
275
kHz
1350
1500
1650
kHz
(1)
FSW floating
FSW floating
(1)
Max. Unit
61
ISW = 0.5 A
ISW = 0.5 A
Typ.
Selected switching
frequency
RFSW = 10 k
Peak current limit
RILIM = 27 k; VFB = 0.6 V
(2)
2.65
3.05
3.45
A
RILIM = 100 k; VFB = 0.6 V
(2)
0.68
0.85
1.01
A
Selected peak current limit
Pulse skipping peak current
(2)
0.5
A
Feedback foldback level
(3)
400
mV
µs
TONMAX
Maximum On time
12
TONMIN
Minimum On time
120
TOFFMIN
Minimum Off time
(3)
150
360
ns
ns
VCC / VBIAS
VCCH
(1)
3.85
4.10
4.30
V
(1)
160
250
340
mV
Switch internal supply from VCC to VBIAS. (1)
VBIAS ramping up from 0 V.
2.84
2.90
2.96
V
VCC UVLO rising threshold
VCCHYST VCC UVLO hysteresis
VBIAS threshold
Hysteresis
SWO
VCC - VBIAS threshold
(3)
Switch internal supply from VCC to VBIAS.
VIN = VCC = 24 V, VBIAS falling from 24 V (1)
to GND.
Hysteresis
(3)
80
3.35
4.05
mV
4.90
750
V
mV
Power consumption
ISHTDWN
Shutdown current from VIN
VEN = GND
11
16
µA
IQUIESC
Quiescent current from VIN
and VCC
LX floating, VFB = 1 V, VBIAS = GND,
FSW floating
2.5
3.4
mA
IQOPVIN
Quiescent current from VIN
and VCC
1.0
1.4
mA
1.6
2.4
mA
IQOPVBIAS
8/42
Quiescent current from
VBIAS
LX floating, VFB = 1 V, VBIAS = 3.3 V,
FSW floating
DocID026362 Rev 4
L7987L
Electrical characteristics
Table 5. Electrical characteristics (continued)
Symbol
Parameter
Test condition
Min.
Typ.
Max. Unit
Enable
VEN
Device OFF level
0.06
0.30
V
Device ON level
0.35
0.90
V
Soft-start
TSSSETUP Soft start setup time
ISS CH
CSS charging current
Delay from UVLO rising to switching
activity
(3)
VSS = GND
640
4.3
5.0
µs
5.7
µA
Error amplifier
VFB
VFB
Voltage feedback
(1)
Voltage feedback
VCOMPH
VFB = GND; VSS = 3.2 V
VCOMPL
VFB = 1 V; VSS = 3.2 V
IFB
FB biasing current
IOSINK
AV0
Output stage sinking
capability
V
0.788 0.800 0.812
V
3.2
VFB = 3.6 V
VFB = GND; SS pin floating; VCOMP = 2 V
IOSOURCE
0.792 0.800 0.808
Unity gain buffer configuration (FB
connected to COMP).COMP voltage
variation due to IOSINK injection lower
than ± 0.1 · VFB.
Error amplifier gain
5
3.5
V
0.1
V
50
nA
(3)
3.1
mA
(3)
5
mA
(3)
100
dB
23
MHz
Unity gain buffer configuration (FB
connected to COMP). No load on COMP (3)
pin.
GBWP
3.35
Synchronization (fan out: 5 slave devices max.)
fSYN MIN
Synchronization frequency
VSYNOUT Master output amplitude
VSYNOW
Output pulse width
VSYNIH
SYNCH slave high level
input threshold
VSYNIL
SYNCH slave low level input
threshold
ISYN
VSYNIW
Slave SYNCH pull-down
current
FSW floating
280
ILOAD = 4 mA
2.45
kHz
ILOAD = 0 A; pin SYNCH floating
3.8
ILOAD = 0 A; pin SYNCH floating
155
225
275
2.0
VSYNCH = 5 V
400
Input pulse width
V
ns
V
650
1.0
V
900
µA
200
ns
PGOOD
VPGDTH
PGOOD rising threshold
VPGDHYST PGOOD hysteresis
VFB rising
VFB falling
DocID026362 Rev 4
0.67
(3)
0.70
30
0.73
V
mV
9/42
42
Electrical characteristics
L7987L
Table 5. Electrical characteristics (continued)
Symbol
Parameter
VPGDLOW PGOOD low level
IPGDLKG
PGOOD leakage current
Test condition
Min.
IPGD = 1 mA, VFB = GND
Typ.
Max. Unit
30
VPGOOD = 61 V; VFB = 0.8 V
mV
0.1
µA
Thermal shutdown
TSHDWN
Thermal shutdown
temperature
(3)
170
°C
THYS
Thermal shutdown
hysteresis
(3)
15
°C
1. Specifications referred to TJ from -40 to +125 °C. Specifications in the -40 to +125 °C temperature range are assured by
design, characterization and statistical correlation.
2. Parameter tested in static condition during testing phase. Parameter value may change over dynamic application condition.
3. Not tested in production.
10/42
DocID026362 Rev 4
L7987L
4
Functional description
Functional description
The L7987L device is based on a voltage mode, constant frequency control loop. The output
voltage VOUT, sensed by the feedback pin (FB), is compared to an internal reference
(0.8 V) providing an error signal on the COMP pin. The COMP voltage level is then
compared to a fixed frequency sawtooth ramp, which finally controls the on- and off-time of
the power switch.
The main internal blocks are shown in the block diagram in Figure 2 on page 4 and can be
summarized as follow.
4.1
The fully integrated oscillator that provides the sawtooth ramp to modulate the duty
cycle and the synchronization signal. Its switching frequency can be adjusted by an
external resistor. The input voltage feed-forward is implemented.
The soft-start circuitry to limit inrush current during the start-up phase.
The voltage mode error amplifier.
The pulse width modulator and the relative logic circuitry necessary to drive the internal
power switch.
The high-side driver for embedded N-channel power MOSFET switch and bootstrap
circuitry. A dedicated high resistance low-side MOSFET, for anti-boot discharge
management purposes, is also present.
The peak current limit sensing block, with programmable threshold, to handle overload
and short-circuit conditions including current foldback and a thermal shutdown block, to
prevent thermal runaway.
The voltage regulator and internal reference, to supply the internal circuitry and provide
a fixed internal reference. The switchover function from VCC to VBIAS can be
implemented for higher efficiency. This block also implements a voltage monitor
circuitry (UVLO) that checks the input and internal voltages.
The output voltage monitor circuitry which releases the PGOOD signal if the sensed
output voltage is above 87% of the target value.
Oscillator and synchronization
Figure 4 shows the block diagram of the oscillator circuit. The internal oscillator provides
a constant frequency clock, whose frequency depends on the resistor externally connected
between the FSW pin and ground.
Figure 4. Oscillator and synchronization
DocID026362 Rev 4
11/42
42
Functional description
L7987L
If the FSW pin is left floating, the programmed frequency is 250 kHz (typ.); if the FSW pin is
connected to an external resistor the programmed switching frequency can be increased up
to 1.5 MHz, as shown in Figure 5. The required RFSW value (expressed in k) is estimated
by Equation 1:
Equation 1
12500
F SW = 250kHz + ---------------R FSW
Figure 5. Switching frequency programmability
To improve the line transient performance, keeping the PWM gain constant versus the input
voltage, the input voltage feed-forward is implemented by changing the slope of the
sawtooth ramp, according to the input voltage change (Figure 6 a).
The slope of the sawtooth also changes if the oscillator frequency is programmed by the
external resistor. In this way a frequency feed-forward is implemented (Figure 6 b) in order
to keep the PWM modulator gain constant versus the switching frequency.
On the SYNCH pin the synchronization signal is generated. This signal has a phase shift of
180° with respect to the clock. This delay is useful when two devices are synchronized
connecting the SYNCH pins together. When SYNCH pins are connected, the device with
a higher oscillator frequency works as master, so the slave device switches at the frequency
of the master but with a delay of half a period. This helps reducing the RMS current flowing
through the input capacitor. Up to five L7987Ls can be connected to the same SYNCH pin;
however, the clock phase shift from master switching frequency to slaves input clock is
180°.
The L7987L device can be synchronized to work at a higher frequency, in the range 250 kHz
- 1500 kHz, providing an external clock signal on the SYNCH pin. The synchronization
changes the sawtooth amplitude, also affecting the PWM gain (Figure 6 c). This change
must be taken into account when the loop stability is studied. In order to minimize the
change of PWM gain, the free running frequency should be set (with a resistor on the FSW
pin) only slightly lower than the external clock frequency.
12/42
DocID026362 Rev 4
L7987L
Functional description
This pre-adjusting of the slave IC switching frequency keeps the truncation of the ramp
sawtooth negligible.
In case two or more (up to five) L7987L SYNCH pins are tied together, the L7987L IC with
higher programmed switching frequency is typically the master device; however, the
SYNCH circuit is also able to synchronize with a slightly lower external frequency, so the
frequency pre-adjustment with the same resistor on the FSW pin, as suggested above, is
required for a proper operation.
DocID026362 Rev 4
13/42
42
Functional description
L7987L
Figure 6. Feed-forward
14/42
DocID026362 Rev 4
L7987L
4.2
Functional description
Soft-start
The soft-start is essential to assure a correct and safe startup of the step-down converter. It
avoids inrush current surge and makes the output voltage increase monotonically.
The soft-start is performed by charging an external capacitor, connected between the SS pin
and ground, with a constant current (5 µA typ.). The SS voltage is used as reference of the
switching regulator and the output voltage of the converter tracks the ramp of the SS
voltage. When the SS pin voltage reaches 0.8 V level, the error amplifier switches to the
internal 0.8 V ± 1% reference to regulate the output voltage.
Figure 7. Soft-start
During the soft-start period the current limit is set to the nominal value.
The dVSS/dt slope is programmed in agreement with Equation 2:
Equation 2
I SS T SS
5A T SS
C SS = ------------------------ = --------------------------V REF
0.8V
Before starting the CSS capacitor charge, the soft-start circuitry turns-on the discharge
switch shown in Figure 7 for TSSDISCH minimum time, in order to completely discharge the
CSS capacitor.
As a consequence, the maximum value for the soft-start capacitor, which assures an almost
complete discharge in case of EN signal toggle, is provided by:
Equation 3
T SSDISCH
C SS –MAX ------------------------------------ 270nF
5 R SSDISCH
given TSSDISCH = 530 s and RSSDISCH = 380 typical values.
DocID026362 Rev 4
15/42
42
Functional description
L7987L
The enable feature allows to put the device into standby mode. With the EN pin lower than
the device OFF level the device is disabled and the power consumption is reduced to less
than 11 A (typ.). With the EN pin higher than the device ON level the device is enabled. If
the EN pin is left floating, an internal pull-down current ensures that the voltage at the pin
reaches the inhibit threshold and the device is disabled. The pin is also VCC compatible.
4.3
Error amplifier and light-load management
The error amplifier (E/A) provides the error signal to be compared with the sawtooth to
perform the pulse width modulation. Its non inverting input is internally connected to a 0.8 V
voltage reference and its inverting input (FB) and output (COMP) are externally available for
feedback and frequency compensation. In this device the error amplifier is a voltage mode
operational amplifier, therefore, with high DC gain and low output impedance.
The uncompensated error amplifier characteristics are summarized in Table 6.
Table 6. Error amplifier characteristics
Parameters
Value
Low frequency gain (A0)
100 dB
GBWP
23 MHz
Output voltage swing
0 to 3.5 V
Source/sink current capability
2 mA / 5 mA
In continuous conduction working mode (CCM), the transfer function of the power section
has two poles due to the LC filter and one zero due to the ESR of the output capacitor.
Different kinds of compensation networks can be used depending on the ESR value of the
output capacitor.
If the zero introduced by the output capacitor helps to compensate the double pole of the LC
filter, a type II compensation network can be used. Otherwise, a type III compensation
network must be used (see Section 5.4 on page 23 for details on the compensation network
design).
In case of light load (i.e.: if the output current is lower than the half of the inductor current
ripple) the L7987L device enters pulse-skipping working mode. The HS MOS is kept off if
the COMP level is below 200 mV (typ.); when this bottom level is reached the integrated
switch is turned on until the inductor current reaches ISKIP value. So, in discontinuous
conduction working mode (DCM), the HS MOS on-time is only related to the time necessary
to charge the inductor up to ISKIP level. Due to current sensing comparator delay, the actual
inductor charge current is slightly impacted by VIN and inductance level.
In order to let the bootstrap capacitor recharge, in case of extremely light load the L7987L is
able to pull-down the LX net through an integrated small LS MOS. In this way the bootstrap
recharge current can flow from VIN through CBOOT, LX and the LS MOS.
This mechanism is activated if the HS MOS has been kept turned-off for more than 3 ms
(typ.).
16/42
DocID026362 Rev 4
L7987L
4.4
Functional description
Low VIN operation
In normal operation (i.e.: VOUT programmed lower than input voltage) when the HS MOS is
turned off, a minimum off time (TOFFMIN) interval is performed.
In case the input voltage falls close or below the programmed output voltage (low dropout,
LDO) the L7987L control loop is able to keep the boot capacitor properly charged by limiting
the HS MOS on time to TONMAX. When this limit is reached the HS MOS is turned-off and a
pull-down resistor between LX and GND is turned on until one of the following conditions is
met:
A negative current limit (300 mA typ.) is reached
A timeout (1 s typ.) is reached.
So doing the L7987L device is able to work in low dropout operation, due to the advanced
boot capacitor management.
4.5
Overcurrent protection
The L7987L device implements an overcurrent protection by sensing the current flowing
through the power MOSFET. Due to the noise created by the switching activity of the power
MOSFET, the current sensing circuitry is disabled during the initial phase of the conduction
time. This avoids an erroneous detection of a fault condition. This interval is generally
known as “masking time” or “blanking time”. The masking time is about 120 ns.
If the overcurrent limit is reached, the power MOSFET is turned off implementing pulse-bypulse overcurrent protection. In the overcurrent condition, the device can skip turn-on pulses
in order to keep the inductor current constant and equal to the current limit, assuming only a
slight drift due to input and output voltage variation.
If, at the end of the “masking time”, the current is higher than the overcurrent threshold, the
power MOSFET is turned off and one pulse is skipped. If, at the following switching on,
when the “masking time” ends, the current is still higher than the overcurrent threshold, the
device skips two pulses. This mechanism is repeated and the device can skip up to seven
pulses (refer to Figure 8).
If at the end of the “masking time” the current is lower than the overcurrent threshold, the
number of skipped cycles is decreased by one unit.
As a consequence, the overcurrent/short-circuit protection acts by switching off the power
MOSFET and reducing the switching frequency down to one eighth of the default switching
frequency, in order to keep constant the output current close to the current limit.
DocID026362 Rev 4
17/42
42
Functional description
L7987L
Figure 8. OCP and frequency scaling
If the sensed output voltage, monitored through FB pin, falls below the VFOLD threshold
(400 mV typ.) the peak current limit threshold is reduced to 1/3 of the nominal value. This
additional feature helps to reduce the IC stress in case of output short-circuit.
As soon as the FB pin increases above the VFOLD threshold, the full peak current limit
threshold is restored. This foldback protection is disabled during the soft-start.
This kind of overcurrent protection is effective if the inductor can be completely discharged
during HS MOS turn-off time, in order to avoid the inductor current to run away. In case of
output short-circuit the maximum switching frequency can be computed by Equation 4.
Equation 4
8 V F + R DCR I LIM
1
F SW MAX ---------------------------------------------------------------------- ---------------------V IN – R ON + R DCR I LIM T ON MIN
Assuming VF = 0.6 V the freewheeling diode direct voltage, RDCR = 70 m the inductor
parasitic resistance, ILIM = IPK = 0.9 A the peak current limit during foldback protection,
RON = 0.30 the HS MOS resistance and TON,MIN = 120 ns the minimum HS MOS on
duration, the maximum FSW frequency which avoids the inductor current run away in case
of output short-circuit and VIN = 61 V is 728 kHz.
If the programmed switching frequency is higher than the above computed limit, an
estimation of the inductor current in case of output short-circuit fault is provided by
Equation 5:
Equation 5
F SW T ON V IN – 8 V F
I LIM = ----------------------------------------------------------------------------------------------------------------------8 R DCR + F SW T ON MIN R ON + R DCR
18/42
DocID026362 Rev 4
L7987L
Functional description
The peak current limit threshold (ILIM) can be programmed in the range 0.85 A - 3.0 A by
selecting the proper RILIM resistor, as suggested in Equation 6:
Equation 6
3A
R ILIM = 27k ---------I LIM
In any case, the maximum high-side MOS RMS current must not be allowed to exceed the
value shown in Table 2 on page 7.
Figure 9. Current limit and programming resistor
2&3WKUHVKROG >$@
3URJUDPPLQJ UHVLVWRU >N@
The minimum programmed current limit can't be lower than ISKIP = 0.5 A (typical), also in
case of foldback detection.
4.6
Overtemperature protection
It is recommended that the device never exceeds the maximum allowable junction
temperature. This temperature increase is mainly caused by the total power dissipated from
the integrated power MOSFET.
To avoid any damage to the device when reaching high temperature, the L7987L device
implements a thermal shutdown feature: when the junction temperature reaches 170 °C
(typ.) the device turns off the power MOSFET and shuts down.
When the junction temperature drops to 155 °C (typ.), the device restarts with a new softstart sequence.
DocID026362 Rev 4
19/42
42
Application information
L7987L
5
Application information
5.1
Input capacitor selection
The input capacitor must be rated for the maximum input operating voltage and the
maximum RMS input current.
Since the step-down converters input current is a sequence of pulses from 0 A to IOUT, the
input capacitor must absorb the equivalent RMS current which can be up to the load current
divided by two (worst case, with duty cycle of 50%). For this reason, the quality of these
capacitors must be very high to minimize the power dissipation generated by the internal
ESR, thereby improving system reliability and efficiency.
The RMS input current (flowing through the input capacitor) is roughly estimated by:
Equation 7
I CIN RMS I OUT D 1 – D
Actual DC/DC conversion duty cycle, D = VOUT/VIN, is influenced by a few parameters:
Equation 8
V OUT + V F
D MAX = ---------------------------------------------------V IN MIN – V SW MAX
V OUT + V F
D MIN = ---------------------------------------------------V IN MAX – V SW MIN
where VF is the freewheeling diode forward voltage and VSW the voltage drop across the
internal high-side MOSFET. Considering the range DMIN to DMAX it is possible to determine
the maximum ICIN,RMS flowing through the input capacitor.
The input capacitor value must be dimensioned to safely handle the input RMS current and
to limit the VIN and VCC ramp-up slew-rate to 0.5 V/s maximum, in order to avoid the
device active ESD protections turn-on.
Different capacitors can be considered:
20/42
Electrolytic capacitors
These are the most commonly used due to their low cost and wide range of operative
voltage. The only drawback is that, considering ripple current rating requirements, they
are physically larger than other capacitors.
Ceramic capacitors
If available for the required value and voltage rating, these capacitors usually have
a higher RMS current rating for a given physical dimension (due to the very low ESR).
The drawback is their high cost.
Tantalum capacitors
Small, good quality tantalum capacitors with very low ESR are becoming more
available. However, they can occasionally burn if subjected to very high current, for
example when they are connected to the power supply.
DocID026362 Rev 4
L7987L
Application information
The amount of the input voltage ripple can be roughly overestimated by Equation 9.
Equation 9
D 1 – D I OUT
V IN PP = ---------------------------------------------- + R ES IN I OUT
C IN F SW
In case of MLCC ceramic input capacitors, the equivalent series resistance (RES,IN) is
negligible.
In addition to the above considerations, a ceramic capacitor with an appropriate voltage
rating and with a value 1 F or higher should always be placed across VIN and power
ground and across VCC and the IC GND pins, as close as possible to the L7987L device.
This solution is necessary for spike filtering purposes.
5.2
Output capacitor selection
The output capacitor is very important in order to satisfy the output voltage ripple
requirement. Using a small inductor value is useful to reduce the size of the choke but
increases the current ripple. So, to reduce the output voltage ripple, a low ESR capacitor is
required. Nevertheless, the ESR of the output capacitor introduces a zero in the open loop
gain, which helps to increase the phase margin of the system. If the zero goes to very high
frequency, typical drawback in case of ceramic output capacitor application, a type III
compensation network must be designed.
The current in the output capacitor has a triangular waveform which generates a voltage
ripple across it. This ripple is due to the capacitive component (charge and discharge of the
output capacitor) and the resistive component (due to the voltage drop across its ESR). So
the output capacitor must be selected in order to have a voltage ripple compliant with the
application requirements.
The amount of the voltage ripple can be estimated starting from the current ripple obtained
by the inductor selection. Assuming IL the inductor current ripple, the output voltage ripple
is roughly overestimated by Equation 10.
Equation 10
I L
V OUT PP I L R ES OUT + ----------------------------------------8 F SW C OUT
Usually the resistive component of the ripple is much higher than the capacitive one, if the
output capacitor adopted is not a multi-layer ceramic capacitor (MLCC) with very low ESR
value.
The output capacitor is important also for loop stability: it fixes the double LC filter pole and
the zero due to its ESR.
The output capacitor is also the key component that provides the current to the load during
a load transient which exceeds the system bandwidth. So, if the high slew rate load
transient is required by the application, the output capacitor must be designed in order to
sustain the load transient or absorbs the energy stored in the inductor until the converter
reacts.
In fact, even if the controller detects immediately the load variation and sets the duty cycle at
100% or 0%, the output current slope is limited by the inductor value, the input and output
voltage.
DocID026362 Rev 4
21/42
42
Application information
L7987L
The output voltage has a drop or overshoot that depends on the ESR and capacitive
charge/discharge, as roughly estimated in Equation 11:
Equation 11
L I OUT
V OUT –LT I OUT R ES OUT + I OUT ---------------------------------------2 C OUT V L
where VL is the voltage applied to the inductor during the load appliance or load release.
Equation 12
D MAX V IN – V OUT
V L =
V OUT
MLCC capacitors have typically low ESR to minimize the ripple but also have low
capacitance that does not minimize the voltage deviation during dynamic load variations.
Electrolytic capacitors, on the other hand, have a large capacitance which minimizes voltage
deviation during load transients whereas they do not show the same ESR values as the
MLCCs, resulting then in higher ripple voltages.
A mix between an electrolytic and MLCC capacitor can be used to minimize ripple as well as
reducing voltage deviation in dynamic mode.
The high bandwidth error amplifier of the L7987L and external compensation feature let
design a wide range of output filter configurations (including all MLCC solutions) and
perform fast transient response.
5.3
Inductor selection
The inductance value fixes the current ripple flowing through the output capacitor. So the
minimum inductance value, in order to have the expected current ripple, must be selected.
The rule to fix the current ripple value is to have a ripple at 20% - 40% of the output current.
In the continuous conduction mode (CCM), the required inductance value can be calculated
by Equation 13:
Equation 13
V OUT
V OUT 1 – --------------
V IN
L = -------------------------------------------------L L F SW
In order to guarantee a maximum current ripple in every condition, Equation 13 must be
evaluated in case of maximum input voltage, assuming VOUT fixed.
Increasing the value of the inductance help to reduce the current ripple but, at the same
time, strongly impacts the converter response time to a dynamic load change. The response
time is the time required by the inductor to change its current from the initial to the final
value. Until the inductor has finished its charging (or discharging) time, the output current is
supplied (or recovered) by the output capacitors.
Further, if the compensation network is properly designed, during a load variation the device
is able to properly change the duty cycle so improving the control loop transient response.
When this condition is reached the response time is only limited by the time required to
change the inductor current, basically by VIN, VOUT and L.
Minimizing the response time, at the end, can help to decrease the output filter total cost
and to reduce the application area.
22/42
DocID026362 Rev 4
L7987L
5.4
Application information
Compensation network
The compensation network must assure stability and good dynamic performance. The loop
of the L7987L device is based on the voltage mode control. The error amplifier is an
operational amplifier with high bandwidth. So, by selecting the compensation network the
E/A is considered as ideal, that is, its bandwidth is much larger than the system one.
Figure 10. Switching regulator control loop simplified model
/
9,1
,QWHUQDO
VZLWFK
5 '&
3RZHU
VHFWLRQ
5 (6
52
&2
3:0
=6
96
58
95()
($
&203
=)
5'
&RPSHQVDWLRQ
QHWZRUN
$0
The transfer function of the PWM modulator, from the error amplifier output (COMP pin) to
the LX pin results in an almost constant gain, due to the voltage feed-forward which
generates a sawtooth with amplitude VS directly proportional to the input voltage:
Equation 14
V IN
1
G PWO = --------- = --------- = 30
VS
k FF
The synchronization of the device with an external clock provided through the SYNCH pin
can modify the PWM modulator gain (see Section 4.1 on page 11 to understand how this
gain changes and how to keep it constant in spite of the external synchronization).
DocID026362 Rev 4
23/42
42
Application information
L7987L
The transfer function of the power section (i.e.: the L-CO filters and the output load) is the
ratio of the parallel of CO and RO and the sum of L and the parallel of CO and RO, including
L and CO parasitics:
Equation 15
1
CO
s
S
RE
RO
RO
C
RD
S
RO
RE
CO
1
C
1
C
D
RD
L
R
s
S
SRE
RE
CO CO
s CO
s C
S
RD
RE
RO
RO O
C
RO
L
s
S
RE
2
RO
CO
L
s
s
C
GL
︵ ︶
given L, RDC, CO, RES and RO the parameters shown in Figure 10. The power section
transfer function can be rewritten as follows:
Equation 16
s
1 + --------------------------2 f zESR
G LC s = G LCO ----------------------------------------------------------------------------2- ;
s
s
1 + ------------------------------- + --------------------
2 Q f LC 2 f LC
RO
G LCO = -------------------------- 1
R O + R DC
Equation 17
1
f zESR = -------------------------------- ;
2 C O R ES
1
1
f LC = ------------------------------------------------------ ----------------------------------------------------R O + R ES
R O + R ES
2 LC O -------------------------- 2 LC O ------------------------R O + R DC
RO
Equation 18
LC O R O + R DC R O + R ES
LC O R O R O + R ES
Q = ----------------------------------------------------------------------------------------------------------- ------------------------------------------------------------------------L + C O R O R DC + R O R ES + R ES R DC
L + C O R O R ES
with the assumption that the inductor parasitic resistance, RDC, is negligible compared to
RO. The closed loop gain is then given by:
Equation 19
GLOOP(s) = GLC(s) • GPWO(s)
• GCOMP(s)
As noted in Section 5.2 on page 21, two different kinds of network can compensate the loop,
depending on the value of fzESR, lower or higher than the regulator required bandwidth.
In Section 5.4.1 and Section 5.4.2 the guidelines to select the type II and type III
compensation network are illustrated.
24/42
DocID026362 Rev 4
L7987L
5.4.1
Application information
Type II compensation network
If the equivalent series resistance (RES) of the output capacitor introduces a zero with
a frequency lower than the desired bandwidth (that is: 2• RES • CO > 1 / BW), this zero
helps stabilize the loop. Electrolytic capacitors show non-negligible ESR (> 30 m typically),
so with this kind of output capacitor the type II network combined with the zero of the ESR
allows to stabilize the loop.
Figure 11. Type II compensation network
9 287
95()
58
($
&203
)%
5)
5'
&)
&3
=)
$0
The type II compensation network transfer function, from VOUT to COMP, is computed in
Equation 20.
Equation 20
GCOMPII ( s)
Z F ( s)
1
1 sC F RF
RU
RU s (C F C P ) 1 sC F C P RF
s
2 f Z 1
s
1
2 f P1
1
s
2 f P 0
Equation 21
f Z1
1
;
2 C F RF
f P0
1
;
2 C F C P RU
f P1
1
2 C F C P RF
The following suggestions can be followed for a quite common compensation strategy,
assuming that CP
工商网监
湘ICP备2023018690号
