LED6000PHR

LED6000 3 A, 61 V monolithic current source with dimming capability Datasheet - production data Applications HTSSOP16 (RTH = 40 °C)  HB LED driving applications  Halogen bulb replacement Features Description  Up to 3 A DC output current The LED6000 is a step-down monolithic switching regulator designed to source up to 3 A DC current for high power LED driving. The 250 mV typical RSENSE voltage drop enhances the efficiency. Digital dimming is implemented by driving the dedicated DIM pin.  4.5 V to 61 V operating input voltage  RDS,ON = 250 m typ.  Adjustable fSW (250 kHz - 1.5 MHz)  Dimming function with dedicated pin  Low IQ shutdown (10 µA typ. from VIN)  Low IQ operating (2.4 mA typ.) The adjustable current limitation, designed to select the inductor RMS in accordance with the nominal output LED current, and the adjustability of the switching frequency allow the size of the application to be compact. The embedded switchover feature on the VBIAS pin maximizes efficiency. Multiple devices can be synchronized by sharing the SYNCH pin to prevent beating noise in low-noise applications, and to reduce the input current RMS value.  ± 3% output current accuracy  Synchronization  Enable with dedicated pin  Adjustable soft-start time  Adjustable current limitation  Low dropout operation (12 µs max.)  VBIAS improves efficiency at light-load The device is fully protected against overheating, overcurrent and output short-circuit.  Output voltage sequencing  Auto recovery thermal shutdown The LED6000 is available in an HTSSOP16 exposed pad package.  MLCC output capacitor Figure 1. Application schematic 4: /$) 7*/ %*. %*. 7*/@ 7*/@ 7#*"4 #005 7$$ - &/ $*/ $#005 -&% -9@ -9@ 7-&% 44 $065 '# '48 $7$$ $44 *-*. $' 3' (/% 36 $0.1 7-&% % $1 34/4 (/% (/% ". May 2015 This is information on a product in full production. DocID027777 Rev 2 1/49 www.st.com Contents LED6000 Contents 1 Block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 2 Pin settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 2.1 Pin connection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 2.2 Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 2.3 Maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 2.4 Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 2.5 ESD protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 3 Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 4 Functional description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 5 2/49 4.1 Oscillator and synchronization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .11 4.2 Soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 4.3 Digital dimming . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 4.4 Error amplifier and light-load management . . . . . . . . . . . . . . . . . . . . . . . 19 4.5 Low VIN operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 4.6 Overcurrent protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 4.7 Overtemperature protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 Application notes - step-down conversion . . . . . . . . . . . . . . . . . . . . . . 24 5.1 Input capacitor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 5.2 Output capacitor and inductor selection . . . . . . . . . . . . . . . . . . . . . . . . . . 25 5.3 Compensation strategy . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 5.4 Thermal considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 5.5 Layout considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31 5.6 Demonstration board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32 DocID027777 Rev 2 LED6000 6 7 Contents Application notes - alternative topologies . . . . . . . . . . . . . . . . . . . . . . 35 6.1 Inverting buck-boost . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35 6.2 Positive buck-boost . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38 6.3 Floating boost . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41 6.4 Compensation strategy for alternative topologies . . . . . . . . . . . . . . . . . . 44 Package information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46 HTSSOP16 package information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46 8 Ordering information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48 9 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48 DocID027777 Rev 2 3/49 49 Block diagram 1 LED6000 Block diagram Figure 2. Block diagram 4/49 DocID027777 Rev 2 LED6000 Pin settings 2 Pin settings 2.1 Pin connection Figure 3. Pin connection (top view) DocID027777 Rev 2 5/49 49 Pin settings 2.2 LED6000 Pin description Table 1. Pin description No. 1 6/49 Pin Description Auxiliary input that can be used to supply a part of the analog circuitry to increase the VBIAS efficiency at the light-load. Connect to GND if not used or bypass with a 1 µF ceramic capacitor if supplied by an auxiliary rail. 2 VIN DC input voltage 3 VIN DC input voltage 4 VCC Filtered DC input voltage to the internal circuitry. Bypass to the signal GND by a 1 µF ceramic capacitor. 5 EN Active high enable pin. Connect to the VCC pin if not used. 6 SS An internal current generator (5 µA typ.) charges the external capacitor to implement the soft-start. 7 SYNCH Master / slave synchronization 8 COMP 9 FB 10 FSW A pull-down resistor to GND selects the switching frequency. 11 ILIM A pull-down resistor to GND selects the peak current limitation. 12 DIM A PWM signal in this input pin implements the LED PWM current dimming. It's pulleddown by an internal 2 µA current. 13 LX Switching node 14 LX Switching node 15 BOOT 16 GND Signal GND -- E. P. The exposed pad must be connected to the signal GND. Output of the error amplifier. The designed compensation network is connected at this pin. Inverting input of the error amplifier Connect an external capacitor (100 nF typ.) between BOOT and LX pins. The gate charge required to drive the internal n-DMOS is recovered by an internal regulator during the off-time. DocID027777 Rev 2 LED6000 2.3 Pin settings Maximum ratings Table 2. Absolute maximum ratings Symbol Min. Max. Unit VIN -0.3 61 V VCC -0.3 61 V VBOOT - GND -0.3 65 V VBOOT - VLX -0.3 4 V VBIAS -0.3 VCC V EN -0.3 VCC V DIM -0.3 VCC V LX -0.3 VIN + 0.3 V SYNCH -0.3 5.5 V SS -0.3 3.6 V FSW -0.3 3.6 V COMP -0.3 3.6 V ILIM -0.3 3.6 V FB -0.3 3.6 V Operating temperature range -40 150 °C TSTG Storage temperature range -65 150 °C TLEAD Lead temperature (soldering 10 sec.) 260 °C 3 A BOOT TJ High-side RMS current IHS 2.4 Description Thermal data Table 3. Thermal data 2.5 Symbol Parameter Value Unit RthJA Thermal resistance junction ambient (device soldered on the STMicroelectronics® demonstration board) 40 °C/W ESD protection Table 4. ESD protection Symbol Test condition Value Unit ESD HBM 2 KV CDM 500 V DocID027777 Rev 2 7/49 49 Electrical characteristics 3 LED6000 Electrical characteristics All the population tested at TJ = 25 °C, VIN = VCC = 24 V, VBIAS = GND, VDIM = VEN = 3 V and RILIM = N. M. unless otherwise specified. Table 5. Electrical characteristics Symbol VIN Parameter Operating input voltage range RDSON HS High-side RDSON Switching frequency fSW IPK ISKIP Test condition Min. (1) 4.5 V 0.25 0.32  0.25 0.42  233 250 267 kHz 225 250 275 kHz 1350 1500 1650 kHz (1) FSW pin floating; FSW pin floating (1) Max. Unit 61 ILX = 0.5 A ILX = 0.5 A Typ. Selected switching frequency RFSW = 10 k Peak current limit ILIM pin floating; VFB = 0.2 V (2) 3.5 4.1 4.7 A Selected peak current limit RILIM = 100 k; VFB = 0.2 V (2) 0.68 0.85 1.01 A ILIM pin floating (2) 0.40 A RILIM = 100 k (2) 0.15 A Pulse skipping peak current TONMIN Minimum on-time TONMAX Maximum on-time TOFFMIN Minimum off-time 120 Refer to Section 4: Functional description for TONMAX details. (3) 150 ns 12 s 360 ns VCC / VBIAS VCCH (1) 3.85 4.10 4.30 V (1) 160 250 340 mV Switch internal supply from VCC to VBIAS. VBIAS ramping up from 0 V. (1) 2.82 2.90 2.98 V Hysteresis (3) Switch internal supply from VCC to VBIAS. VIN = VCC = 24 V, VBIAS falling from 24 V to GND. (1) Hysteresis (3) VCC UVLO rising threshold VCCHYST VCC UVLO hysteresis VBIAS threshold SWO VCC -VBIAS threshold 80 3.35 4.05 mV 4.90 750 V mV Power consumption ISHTDWN Shutdown current from VIN VEN = GND 10 15 A IQUIESC Quiescent current from VIN and VCC LX floating, VFB = 1 V, VBIAS = GND, FSW floating. 2.4 3.4 mA 8/49 DocID027777 Rev 2 LED6000 Electrical characteristics Table 5. Electrical characteristics (continued) Symbol Parameter IQOPVIN Quiescent current from VIN and VCC IQOPVBIAS Quiescent current from VBIAS Test condition Min. LX floating, VFB = 1 V, VBIAS = 3.3 V, FSW floating Typ. Max. Unit 0.9 1.4 mA 1.5 2.4 mA Enable VENL Device OFF level 0.06 0.30 V VENH Device ON level 0.35 0.90 V Soft-start TSSSETUP Soft-start setup time ISSCH CSS charging current Delay from UVLO rising to switching activity (3) VSS = GND s 640 4.3 5.0 5.7 A Error amplifier VFB Voltage feedback (1) VCOMPH VFB = GND; VSS = 3.2 V VCOMPL VFB = 3.2 V; VSS = 3.2 V IFB FB biasing current IOSINK AV0 Output stage sinking capability Unity gain buffer configuration (FB connected to COMP). No load on COMP pin. Error amplifier gain Unity gain buffer configuration (FB connected to COMP). No load on COMP pin. GBWP 0.250 0.258 V 0.240 0.250 0.260 V 3.20 3.35 3.50 V 0.1 V 50 nA VFB = 3.6 V VFB = GND; SS pin floating; VCOMP = 2 V IOSOURCE 0.242 5 (3) 3.1 mA (3) 5 mA (3) 100 dB (3) 23 MHz Synchronization (fan out: 5 slave devices max.) fSYN MIN Synchronization frequency VSYNOUT Master output amplitude VSYNOW Output pulse width VSYNIH SYNCH slave high level input threshold VSYNIL SYNCH slave low level input threshold ISYN VSYNIW Slave SYNCH pull-down current Pin FSW floating 280 ILOAD = 4 mA 2.45 kHz ILOAD = 0 A; pin SYNCH floating ILOAD = 0 A; pin SYNCH floating 3.8 150 215 280 V ns 2.0 V 1.0 VSYNCH = 5 V Input pulse width 450 150 DocID027777 Rev 2 700 950 A ns 9/49 49 Electrical characteristics LED6000 Table 5. Electrical characteristics (continued) Symbol Parameter Test condition Min. Typ. Max. Unit Dimming VDIMH DIM rising threshold VDIML DIM falling threshold VDIMPD TDIMTO DIM pull-down current 1.23 VDIM = 2 V Dimming timeout 0.75 1.00 0.5 1.5 1.7 V V 2.5 A (3) 42 ms Thermal shutdown TSHDWN Thermal shutdown temperature (3) 170 °C THYS Thermal shutdown hysteresis (3) 15 °C 1. Specifications referred to TJ from -40 to +125 °C. Specifications in the -40 to +125 °C temperature range are assured by design, characterization and statistical correlation. 2. Parameter tested in static condition during testing phase. Parameter value may change over dynamic application condition. 3. Not tested in production. 10/49 DocID027777 Rev 2 LED6000 4 Functional description Functional description The LED6000 device is based on a voltage mode, constant frequency control loop. The LEDs current, monitored through the voltage drop on the external current sensing resistor, RSNS, is compared to an internal reference (0.25 V) providing an error signal on the COMP pin. The COMP voltage level is then compared to a fixed frequency sawtooth ramp, which finally controls the on- and off-time of the power switch. The main internal blocks are shown in Figure 2: Block diagram on page 4 and can be summarized as follow. 4.1  The fully integrated oscillator that provides the sawtooth ramp to modulate the duty cycle and the synchronization signal. Its switching frequency can be adjusted by an external resistor. The input voltage feedforward is implemented.  The soft-start circuitry to limit the inrush current during the startup phase.  The voltage mode error amplifier.  The pulse width modulator and the relative logic circuitry necessary to drive the internal power switch.  The high-side driver for the embedded N-channel power MOSFET switch and bootstrap circuitry. A dedicated high resistance low-side MOSFET, for anti-boot discharge management purposes, is also present.  The peak current limit sensing block, with a programmable threshold, to handle the overload including a thermal shutdown block, to prevent the thermal runaway.  The bias circuitry, which includes a voltage regulator and an internal reference, to supply the internal circuitry and provide a fixed internal reference and manages the current dimming feature. The switchover function from VCC to VBIAS can be implemented for higher efficiency. This block also implements voltage monitor circuitry (UVLO) that checks the input and internal voltages. Oscillator and synchronization Figure 4 shows the block diagram of the oscillator circuit. The internal oscillator provides a constant frequency clock, whose frequency depends on the resistor externally connected between the FSW pin and ground. Figure 4. Oscillator and synchronization DocID027777 Rev 2 11/49 49 Functional description LED6000 If the FSW pin is left floating, the programmed frequency is 250 kHz (typ.); if the FSW pin is connected to an external resistor, the programmed switching frequency can be increased up to 1.5 MHz, as shown in Figure 5. The required RFSW value (expressed in k) is estimated by Equation 1: Equation 1 FSW  250kHz  12500 RFSW Figure 5. Switching frequency programmability To improve the line transient performance, keeping the PWM gain constant versus the input voltage, the input voltage feedforward is implemented by changing the slope of the sawtooth ramp, according to the input voltage change (Figure 6 a). The slope of the sawtooth also changes if the oscillator frequency is programmed by the external resistor. In this way a frequency feedforward is implemented (Figure 6 b) in order to keep the PWM modulator gain constant versus the switching frequency. On the SYNCH pin the synchronization signal is generated. This signal has a phase shift of 180° with respect to the clock. This delay is useful when two devices are synchronized connecting the SYNCH pins together. When SYNCH pins are connected, the device with a higher oscillator frequency works as a master, so the slave device switches at the frequency of the master but with a delay of half a period. This helps reducing the RMS current flowing through the input capacitor. Up to five LED6000s can be connected to the same SYNCH pin; however, the clock phase shift from master switching frequency to the slave input clock is 180°. The LED6000 device can be synchronized to work at a higher frequency, in the range of 250 kHz - 1500 kHz, providing an external clock signal on the SYNCH pin. The synchronization changes the sawtooth amplitude, also affecting the PWM gain (Figure 6 c). This change must be taken into account when the loop stability is studied. In order to 12/49 DocID027777 Rev 2 LED6000 Functional description minimize the change of the PWM gain, the free-running frequency should be set (with a resistor on the FSW pin) only slightly lower than the external clock frequency. This pre-adjusting of the slave IC switching frequency keeps the truncation of the ramp sawtooth negligible. In case two or more (up to five) LED6000 SYNCH pins are tied together, the LED6000 IC with higher programmed switching frequency is typically the master device; however, the SYNCH circuit is also able to synchronize with a slightly lower external frequency, so the frequency pre-adjustment with the same resistor on the FSW pin, as suggested above, is required for a proper operation. The SYNCH signal is provided as soon as EN is asserted high; however, if DIM is kept low for more than TDIMTO timeout, the SYNCH signal is no more available until DIM re-assertion high. DocID027777 Rev 2 13/49 49 Functional description LED6000 Figure 6. Feedforward 14/49 DocID027777 Rev 2 LED6000 4.2 Functional description Soft-start The soft-start is essential to assure a correct and safe startup of the step-down converter. It avoids an inrush current surge and makes the output voltage to increase monotonically. The soft-start is performed as soon as the EN and DIM pin are asserted high; when this occurs, an external capacitor, connected between the SS pin and ground, is charged by a constant current (5 µA typ.). The SS voltage is used as reference of the switching regulator and the output voltage of the converter tracks the ramp of the SS voltage. When the SS pin voltage reaches the 0.25 V level, the error amplifier switches to the internal 0.25 V reference to regulate the output voltage. Figure 7. Soft-start During the soft-start period the current limit is set to the nominal value. The dVSS/dt slope is programmed in agreement with Equation 2: Equation 2 C SS  I SS  TSS 5A  TSS  VREF 0.25V Before starting the CSS capacitor charge, the soft-start circuitry turns-on the discharge switch shown in Figure 7 for TSSDISCH minimum time, in order to completely discharge the CSS capacitor. As a consequence, the maximum value for the soft-start capacitor, which assures an almost complete discharge in case of the EN signal toggle, is provided by: Equation 3 C SS _ MAX  TSSDISCH  270nF 5  RSSDISCH given TSSDISCH = 530 µs and RSSDISCH = 380  typical values. The enable feature allows to put the device into the standby mode. With the EN pin lower than VENL the device is disabled and the power consumption is reduced to 10 A (typ.). If the EN pin is higher than VENH, the device is enabled. If the EN pin is left floating, an internal pull-down current ensures that the voltage at the pin reaches the inhibit threshold and the device is disabled. The pin is also VCC compatible. DocID027777 Rev 2 15/49 49 Functional description 4.3 LED6000 Digital dimming The switching activity is inhibited as long as the DIM pin is kept below the VDIML threshold. When the DIM is asserted low, the HS MOS is turned-off as soon as the minimum on-time is expired and the COMP pin is parked close to the maximum ramp peak value, in order to limit the input inrush current when the IC restarts the switching activity. The internal oscillator and, consequently, the IC quiescent current are reduced only if the DIM is kept low for more than TDIMTO timeout. The inductor current dynamic performance, when dimming input goes high, depends on the designed system response. The best dimming performance is obtained by maximizing the bandwidth and phase margin, when possible. As a general rule, the output capacitor minimization improves dimming performance in terms of the shorter LEDs current rising time and reduced inductor peak current. An oversized output capacitor value requires an extra current for the fast charge so generating an inductor current overshoot and oscillations. Refer also to Section 5.2 on page 25 for output capacitor design hints. The dimming performance depends on the current pulse shape specification of the final application. Figure 8. Dimming operation The ideal current pulse has rectangular shape; however, in any case it degenerates into a trapezoid or, at worst, into a triangle, depending on the ratio (tRISE + tFALL)/ TLED. 16/49 DocID027777 Rev 2 LED6000 Functional description Figure 9. Dimming operation (rising edge) - VIN = 44 V, 12 LED Figure 10. Dimming operation (falling edge) - VIN = 44 V, 12 LED DocID027777 Rev 2 17/49 49 Functional description LED6000 In Figure 11 a very short DIM pulse is shown, measured in the standard demonstration board, VIN = 44 V, 12 LEDs. The programmed LED current, 1 A, is reached at the end of the DIM pulse (35 µs). Figure 11. Dimming operation - short DIM pulse The above consideration is crucial when short DIM pulses are expected in the final application. Once the external power components and the compensation network are selected, a direct measurement to determine tRISE and tFALL is necessary to certify the achieved dimming performance. When the DIM is forced above the VDIMH threshold after TDIMTO has elapsed, a new softstart sequence is performed. 18/49 DocID027777 Rev 2 LED6000 4.4 Functional description Error amplifier and light-load management The error amplifier (E/A) provides the error signal to be compared with the sawtooth to perform the pulse width modulation. Its non-inverting input is internally connected to a 0.25 V voltage reference and its inverting input (FB) and output (COMP) are externally available for feedback and frequency compensation. In this device the error amplifier is a voltage mode operational amplifier, therefore, with the high DC gain and low output impedance. The uncompensated error amplifier characteristics are summarized inTable 6. Table 6. Error amplifier characteristics Parameters Value Low frequency gain (A0) 100 dB GBWP 23 MHz Output voltage swing 0 to 3.5 V Source/sink current capability 3.1 mA / 5 mA In the continuous conduction working mode (CCM), the transfer function of the power section has two poles due to the LC filter and one zero due to the ESR of the output capacitor. Different kinds of compensation networks can be used depending on the ESR value of the output capacitor. If the zero introduced by the output capacitor helps to compensate the double pole of the LC filter, a type II compensation network can be used. Otherwise, a type III compensation network must be used (see Section 5.3 on page 27 for details on the compensation network design). In case of the light-load (i.e.: if the output current is lower than the half of the inductor current ripple) the LED6000 enters the pulse-skipping working mode. The HS MOS is kept off if the COMP level is below 200 mV (typ.); when this bottom level is reached the integrated switch is turned on until the inductor current reaches ISKIP value. So, in the discontinuous conduction working mode (DCM), the HS MOS on-time is only related to the time necessary to charge the inductor up to the ISKIP level. The ISKIP threshold is reduced with increasing the RILIM resistor value, as shown also in Table 5 on page 8 and plotted in Figure 12, so allowing the LED6000 device work in the continuous conduction mode also in case lower current LEDs are selected. DocID027777 Rev 2 19/49 49 Functional description LED6000 Figure 12. ISKIP typical current and RILIM value However, due to the current sensing comparator delay, the actual inductor charge current is slightly impacted by the VIN and selected inductor value. In order to let the bootstrap capacitor recharge, in case of an extremely light-load the LED6000 is able to pull-down the LX net through an integrated small LS MOS. In this way the bootstrap recharge current can flow from the VIN through the CBOOT, LX and LS MOS. This mechanism is activated if the HS MOS has been kept turned-off for more than 3 ms (typ.). 4.5 Low VIN operation In normal operation (i.e.: VOUT programmed lower than input voltage) when the HS MOS is turned off, a minimum off time (TOFFMIN) interval is performed. In case the input voltage falls close or below the programmed output voltage (low dropout, LDO) the LED6000 control loop is able to increase the duty cycle up to 100%. However, in order to keep the boot capacitor properly recharged, a maximum HS MOS on-time is limited (TONMAX). When this limit is reached the HS MOS is turned-off and an integrated switch working as a pull-down resistor between the LX and GND is turned on, until one of the following conditions is met:  A negative current limit (300 mA typ.) is reached  A timeout (1 µs typ.) is reached. So doing the LED6000 device is able to work in the low dropout operation, due to the advanced boot capacitor management, and the effective maximum duty cycle is about 12 s / 13 s = 92%. 20/49 DocID027777 Rev 2 LED6000 4.6 Functional description Overcurrent protection The LED6000 implements overcurrent protection by sensing the current flowing through the power MOSFET. Due to the noise created by the switching activity of the power MOSFET, the current sensing circuitry is disabled during the initial phase of the conduction time. This avoids erroneous detection of fault condition. This interval is generally known as “masking time” or “blanking time”. The masking time is about 120 ns. If the overcurrent limit is reached, the power MOSFET is turned off implementing pulse by pulse overcurrent protection. In the overcurrent condition, the device can skip turn-on pulses in order to keep the output current constant and equal to the current limit. If, at the end of the “masking time”, the current is higher than the overcurrent threshold, the power MOSFET is turned off and one pulse is skipped. If, at the following switching on, when the “masking time” ends, the current is still higher than the overcurrent threshold, the device skips two pulses. This mechanism is repeated and the device can skip up to seven pulses (refer to Figure 13). If at the end of the “masking time” the current is lower than the overcurrent threshold, the number of skipped cycles is decreased by one unit. As a consequence, the overcurrent protection acts by turning off the power MOSFET and reducing the switching frequency down to one eighth of the default switching frequency, in order to keep constant the output current close to the current limit. Figure 13. OCP and frequency scaling This kind of overcurrent protection is effective if the inductor can be completely discharged during HS MOS turn-off time, in order to avoid the inductor current to run away. In case of the output short-circuit the maximum switching frequency can be computed by the following equation: Equation 4 FSW ,MAX  8  VF  RDCR  I LIM  1  VIN  RON  RDCR   I LIM TON ,MIN DocID027777 Rev 2 21/49 49 Functional description LED6000 Assuming VF = 0.6 V the free-wheeling diode direct voltage, RDCR = 30 m inductor parasitic resistance, ILIM = IPK = 4 A the peak current limit, RON = 0.25  HS MOS resistance and TON,MIN = 120 ns minimum HS MOS on duration, the maximum FSW frequency which avoids the inductor current runaway in case of the output short-circuit and VIN = 61 V is 801 kHz. If the programmed switching frequency is higher than the above computed limit, an estimation of the inductor current in case of the output short-circuit fault is provided by Equation 5: Equation 5 I LIM  FSW  TON VIN  8 VF 8  RDCR  FSW  TON ,MIN RON  RDCR  The peak current limit threshold (ILIM) can be programmed in the range 0.85 A - 4.0 A by selecting the proper RILIM resistor, as suggested in Equation 6. Equation 6 RILIM  20k  I PK I LIM IPK is the default LED6000 current limit in case of RILIM not mounted, as shown in Table 5 on page 8. Figure 14. Current limit and programming resistor 22/49 DocID027777 Rev 2 LED6000 4.7 Functional description Overtemperature protection It is recommended that the device never exceeds the maximum allowable junction temperature. This temperature increase is mainly caused by the total power dissipated by the integrated power MOSFET. To avoid any damage to the device when reaching high temperature, the LED6000 implements a thermal shutdown feature: when the junction temperature reaches 170 °C (typ.) the device turns off the power MOSFET and shuts-down. When the junction temperature drops to 155 °C (typ.), the device restarts with a new softstart sequence. DocID027777 Rev 2 23/49 49 Application notes - step-down conversion LED6000 5 Application notes - step-down conversion 5.1 Input capacitor selection The input capacitor must be rated for the maximum input operating voltage and the maximum RMS input current. Since the step-down converters input current is a sequence of pulses from 0A to IOUT, the input capacitor must absorb the equivalent RMS current which can be up to the load current divided by two (worst case, with duty cycle of 50%). For this reason, the quality of these capacitors must be very high to minimize the power dissipation generated by the internal ESR, thereby improving system reliability and efficiency. The RMS input current (flowing through the input capacitor) in step-down conversion is roughly estimated by: Equation 7 I CIN , RMS  I OUT  D  1  D  The actual DC/DC conversion duty cycle, D = VOUT/VIN, is influenced by a few parameters: Equation 8 DMAX  VOUT  VF VIN ,MIN  VSW ,MAX DMIN  VOUT  VF VIN ,MAX  VSW ,MIN where VF is the freewheeling diode forward voltage and VSW the voltage drop across the internal high-side MOSFET. Considering the range DMIN to DMAX it is possible to determine the maximum ICIN,RMS flowing through the input capacitor. The input capacitor value must be dimensioned to safely handle the input RMS current and to limit the VIN and VCC ramp-up slew rate to 0.5 V/µs maximum, in order to avoid the device active ESD protections turn-on. 24/49 DocID027777 Rev 2 LED6000 Application notes - step-down conversion Different capacitors can be considered:  Electrolytic capacitors These are the most commonly used due to their low cost and wide range of operative voltage. The only drawback is that, considering ripple current rating requirements, they are physically larger than other capacitors.  Ceramic capacitors If available for the required value and voltage rating, these capacitors usually have a higher RMS current rating for a given physical dimension (due to the very low ESR). The drawback is their high cost.  Tantalum capacitor Small, good quality tantalum capacitors with very low ESR are becoming more available. However, they can occasionally burn if subjected to a very high current, for example when they are connected to the power supply. The amount of the input voltage ripple can be roughly overestimated by Equation 9. Equation 9 VIN , PP  D  1  D   I OUT  RES , IN  I OUT C IN  FSW In case of MLCC ceramic input capacitors, the equivalent series resistance (RES,IN) is negligible. In addition to the above considerations, a ceramic capacitor with an appropriate voltage rating and with a value 1 µF or higher should always be placed across VIN and power ground and across VCC and the IC GND pins, as close as possible to the LED6000 device. This solution is necessary for spike filtering purposes. 5.2 Output capacitor and inductor selection The output capacitor is very important in order to satisfy the output voltage ripple requirement. Using a small inductor value is useful to reduce the size of the choke but increases the current ripple. So, to reduce the output voltage ripple, a low ESR capacitor is required. The current in the output capacitor has a triangular waveform which generates a voltage ripple across it. This ripple is due to the capacitive component (charge and discharge of the output capacitor) and the resistive component (due to the voltage drop across its ESR). So the output capacitor must be selected in order to have a voltage ripple compliant with the application requirements. The allowed LED current ripple (ILED,PP) is typically from 2% to 5% of the LED DC current. DocID027777 Rev 2 25/49 49 Application notes - step-down conversion LED6000 Figure 15. LEDs small signal model Based on the small signal model typically adopted for LEDs (as shown in Figure 15), the amount of the inductor current ripple which flows through the LEDs can be estimated by the following equation: Equation 10 I LED s   I L s   1  sCO RES 1  sCO  RES  RSNS  N  Rd  The typical LED dynamic resistance, for high current LEDs, is about 0.9  to 1 . The output capacitor, CO, is typically an MLCC ceramic capacitor in the range of 1 µF and with equivalent series resistance lower than 10 m, as a consequence the zero due to the time constant CO * RES is in the range of 10 MHz or above. Starting from Equation 10 it's possible to roughly estimate the required CO value: Equation 11 CO  8  2  I L ,PP  2  f SW  RSNS 1  N  Rd   I LED,PP In the above equation it has been assumed that the total inductor current ripple is well approximated by the first Fourier harmonic, 8/2 * IL,PP, due to the inductor current triangular shape. The inductance value fixes the current ripple flowing through the output capacitor and LEDs. So the minimum inductance value, in order to have the expected current ripple, must be selected. The rule to fix the current ripple value is to have a ripple at 30% - 60% of the programmed LEDs current. In the continuous conduction mode (CCM), the required inductance value can be calculated by Equation 12. 26/49 DocID027777 Rev 2 LED6000 Application notes - step-down conversion Equation 12  V VOUT  1  OUT VIN  L I L  FSW    In order to guarantee a maximum current ripple in every condition, Equation 12 must be evaluated in case of maximum input voltage, assuming VOUT fixed. Increasing the value of the inductance helps to reduce the current ripple but, at the same time, strongly impacts the converter response time in case of high frequency dimming requirements. On the other hand, with a lower inductance value the regulator response time is improved but the power conversion efficiency is impacted and the output capacitor must be increased to limit the current ripple flowing through the LEDs. As usually, the L-CO choice is a trade-off among multiple design parameters. 5.3 Compensation strategy The compensation network must assure stability and good dynamic performance. The loop of the LED6000 is based on the voltage mode control. The error amplifier is an operational amplifier with high bandwidth. So, by selecting the compensation network the E/A is considered as ideal, that is, its bandwidth is much larger than the system one. Figure 16. Switching regulator control loop simplified model DocID027777 Rev 2 27/49 49 Application notes - step-down conversion LED6000 The transfer function of the PWM modulator, from the error amplifier output (COMP pin) to the LX pin results in an almost constant gain, due to the voltage feedforward which generates a sawtooth with amplitude VS directly proportional to the input voltage: Equation 13 G PW 0  1 1  VS k FF  VIN For the LED6000 the feedforward gain is KFF = 1/30. The synchronization of the device with an external clock provided through the SYNCH pin can modify the PWM modulator gain (see Section 4.1 on page 11 to understand how this gain changes and how to keep it constant in spite of the external synchronization). The transfer function of the power section (i.e.: the voltage across RSNS resulting as a variation of the duty cycle) is: Equation 14 G LC ( s )  VSNS ( s )  d ( s) RSNS  V IN RSNS  s  L  R DC  N  Rd // 1  s  C O  R ES s  CO given L, RDC, CO, RES, RSNS and Rd the parameters shown in Figure 16. The power section transfer function can be rewritten as follows: Equation 15 1 G LC ( s )  G LC 0  1 s 2  f z  s s   2  Q  f LC  2  f LC    2 ; G LC 0  RSNS  V IN RSNS  N  Rd Equation 16 fz  1 ; 2  CO  RES  N  Rd  1 f LC  2 LCO N  Rd N  Rd  RSNS Equation 17 Q LCO  RSNS  N  Rd  N  Rd L  CO  RSNS  N  Rd with the assumption that the inductor parasitic resistance, RDC, and the output capacitor parasitic resistance, RES, are negligible compared to LED dynamic resistance, Rd. The closed loop gain is then given by: Equation 18 GLOOP (s)  GLC (s)  GPW 0 (s)  GCOMP (s) 28/49 DocID027777 Rev 2 LED6000 Application notes - step-down conversion As shown in Equation 16, fz depends on the output capacitor parasitic resistance and on the LEDs dynamic resistance. Following the considerations summarized in Section 5.2 on page 25, in the typical application the programmed control loop bandwidth (BW) might be higher than fz, so this zero helps stabilize the loop. If this assumption is verified, a type II compensation network is required for loop stabilization. In Figure 17 the type II compensation network is shown. Figure 17. Type II compensation network The type II compensation network transfer function, from VSNS to COMP, is computed in Equation 19. Equation 19 Equation 20 f Z1  1 ; 2  C F RF f P0  1 ; 2  C F  C P   RU f P1  1 2  C F // C P RF The following suggestions can be followed for a quite common compensation strategy, assuming that CP