End of Life. Last Available Purchase Date is 31-Dec-2014
Si9140
Vishay Siliconix
MP Controller For High Performance Process Power Supplies
FEATURES
Runs on 3.3- or 5-V Supplies
Adjustable, High Precision Output
Voltage
High Frequency Operation (>1 MHz)
High Efficiency Synchronous
Switching
Full Set of Protection Circuitry
2000-V ESD Rating (Si9140CQ/DQ)
DESCRIPTION
Siliconix’ Si9140 Buck converter IC is a high-performance,
surface-mount switchmode controller made to power the new
generation of low-voltage, high-performance microprocessors. The Si9140 has an input voltage range of 3 to
6.5 V to simplify power supply designs in desktop PCs. Its
high-frequency switching capability and wide bandwidth
feedback loop provide tight, absolute static and transient load
regulation. Circuits using the Si9140 can be implemented with
low-profile, inexpensive inductors, and will dramatically
minimize power supply output and processor decoupling
capacitance. The Si9140 is designed to meet the stringent
regulation requirements of new and future high-frequency
microprocessors, while improving the overall efficiency in new
“green” systems.
Today’s state-of-the-art microprocessors run at frequencies
over 100 MHz. Processor clock speeds are going up and so
are current requirements, but operating voltages are going
down. These simultaneous changes have made dedicated,
high-frequency, point-of-use buck converters an essential part
of any system design. These point-of-use converters must
operate at higher frequencies and provide wider feedback
bandwidths than existing converters, which typically operate
at less than 250 kHz and have feedback bandwidths of less
than 50 kHz. The Si9140’s 100-kHz feedback loop bandwidth
ensures a minimum improvement of one-half the required
output/decoupling capacitance, resulting in a tremendous
reduction in board size and cost of implementation.
With the microprocessing power of any PC representing an
investment of hundreds of dollars, designers need to ensure
that the reliable operation of the processor will not be affected
by the power supply. The Si9140 provides this assurance. A
demo board, the Si9140DB, is available.
Si9140CQ-T1 and Si9140DQ-T1 are available in lead free.
APPLICATION CIRCUIT
VIN
VCCP
R3
+
VOUT
C3
R2
R1
C1
L1
2 x Si4435DY
2 x Si4410DY
+
D1
Power-Good
R4
C2
U1
Si9140
C4
1
2
C5
3
R5
4
5
6
R6
C6
7
C7
8
VDD
VS
MON
DR
VGOOD
COMP
FB
NI
VREF
GND
16
15
14
DS
13
PGND
12
UVLOSET
11
COSC
10
ROSC
9
ENABLE
R13
C8
R7
R10
0.1%
C9
R9
R12
0.1%
R8
C10
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
R11
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1
Si9140
Vishay Siliconix
ABSOLUTE MAXIMUM RATINGS
Voltages Referenced to GND.
VDD, VS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 V
PGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . "0.3 V
VDD to VS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.3 V
Thermal Impedance (QJA)
16-Pin SOIC (Y Suffix) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 140_C/W
16-Pin TSSOP (Q Suffix) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 135_C/W
Linear Inputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.3 V to VDD +0.3 V
Logic Inputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.3 V to VDD +0.3 V
Peak Output Drive Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 350 mA
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65 to 150_C
Operating Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150_C
Power Dissipation (Package)a
16-Pin SOIC (Y Suffix)\b . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 900 mW
16-Pin TSSOP (Q Suffix)c . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 925 mW
Operating Temperature
C Suffix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0_ to 70_C
D Suffix . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −40_ to 85_C
Notes
a. Device mounted with all leads soldered or welded to PC board.
b. Derate 7.2 mW/_C above 25_C.
c. Derate 7.4 mW/_C above 25_C.
* . Exposure to Absolute Maximum rating conditions for extended periods may affect device reliability. Stresses above Absolute Maximum rating may cause permanent
damage. Functional operation at conditions other than the operating conditions specified is not implied. Only one Absolute Maximum rating should be applied at any
one time
RECOMMENDED OPERATING RANGE
Voltages Referenced to GND.
VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 V to 6.5 V
COSC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47 pF to 200 pF
VS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 V to 6.5 V
Linear Inputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0 to VDD
fOSC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 kHz to 2 MHz
Digital Inputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0 to VDD
ROSC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 kW to 250 kW
VREF Load Resistance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . >150 kW
SPECIFICATIONS
Limits
Test Conditions
Unless Otherwise Specifieda
Parameter
Symbol
C Suffix 0 to 70_C
D Suffix −40 to 85_C
3 V v VDD v 6.5 V, VDD = VS
GND = PGND
Minb
IREF = −10 mA
1.455
Typ
Maxb
Unit
Reference
Output Voltage
VREF
TA = 25_C
1.477
1.545
1.50
1.523
1.0
1.15
V
Oscillator
Maximum Frequencyc
Accuracy
ROSC Voltage
Voltage Stabilityc
Temperature Stabilityc
fMAX
VDD = 5 V, COSC = 47 pF, ROSC = 5.0 kW
2.0
fOSC
VDD = 5 V
COSC = 100 pF, ROSC = 7.50 kW, TA = 25_C
0.85
4 V v VDD v 6 V, Ref to 5 V, TA = 25_C
−8
VROSC
Df/f
1.0
Referenced to 25_C
MHz
V
8
"5
%
Error Amplifier (COSC = GND, OSC DISABLED)
Input Bias Current
IFB
Open Loop Voltage Gain
AVOL
Offset Voltage
VOS
Unity Gain Bandwidthc
BW
Output Current
Power Supply Rejectionc
IEA
VNI = VREF , VFB = 1.0 V
VNI = VREF
−1.0
47
55
−15
0
1.0
mA
15
mV
dB
10
Source (VFB = 1 V, NI = VREF)
Sink (VFB = 2 V, NI = VREF)
−2.0
0.4
MHz
−1.0
0.8
PSRR
3 V < VDD < 6.5 V
60
VUVLOHL
UVLOSET High to Low
VUVLOLH
UVLOSET Low to High
1.2
VHYS
VUVLOLH − VUVLOHL
175
mA
dB
UVLOSET Voltage Monitor
Under Voltage Lockout
Hysteresis
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2
0.85
1.0
1.15
V
mV
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
Si9140
Vishay Siliconix
SPECIFICATIONS
Limits
Test Conditions
Unless Otherwise Specifieda
C Suffix 0 to 70_C
D Suffix −40 to 85_C
Symbol
3 V v VDD v 6.5 V, VDD = VS
GND = PGND
Minb
IUVLO(SET)
VUVLO = 0 to VDD
−100
Output High Voltage
VOH
VS = VDD = 5 V, IOUT = −10 mA
4.7
Output Low Voltage
VOL
VS = VDD = 5 V, IOUT = 10 mA
Peak Output Current
ISOURCE
VS = VDD = 5 V, VOUT = 0 V
Peak Output Current
ISINK
VS = VDD = 5 V, VOUT = 5 V
Break-Before-Make
tBBM
VDD = 6.5 V
40
ENABLE Turn-On Delay
tdEN
ENABLE Delay to Output, ENLH, VDD = 5 V
1.5
ENABLE Logic Low
VENL
ENABLE Logic High
VENH
Parameter
Typ
Maxb
Unit
100
nA
UVLOSET Voltage Monitor
UVLO Input Current
Output Drive (DR and DS)
4.8
200
0.2
0.3
−380
−260
V
mA
300
nS
Logic
ENABLE Input Current
ms
0.2 VDD
V
0.8 VDD
IEN
ENABLE = 0 to VDD
−1.0
1.0
mA
VGOOD Comparator (Voltage-Good Comparator)
Input Offset Voltage
VOS
−45
VIN Common Mode Voltage = VREF, VDD = 5 V
0
Input Hysteresis
VINHYS
Input Bias Current
IBMON
VIN = VREF, VDD = 5 V
−1
0
Output Sink I
ISINK
VOUT = 5 V, VDD = 5 V
6
9
Output Low Voltage
VOL
IOUT = 2 mA, VDD = 5 V
45
mV
10
1
mA
mA
350
500
mV
Supply
Supply Current—Normal Mode
IDD
Supply Current—Standby Mode
fOSC = 1 MHz, ROSC = 7.50 kW
1.6
2.3
mA
ENABLE < 0.4 V
250
330
mA
Notes
a. 100 pF includes CSTRAY on COSC.
b. The algebraic convention whereby the most negative value is a minimum and the most positive a maximum, is used in this data sheet.
c. Guaranteed by design, not subject to production testing.
TYPICAL CHARACTERISTICS (25_C UNLESS OTHERWISE NOTED)
VREF vs. Supply Voltage
1.515
VREF vs. Temperature
1.510
1.510
1.505
1.505
1.500
V REF (V)
V REF (V)
VREF with 10 mA Load
1.500
VDD = 3 V
1.495
1.490
1.490
1.485
1.485
3.0
3.5
4.0
4.5
5.0
5.5
VDD − Supply Voltage (V)
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
6.0
6.5
VDD = 6 V
1.495
1.480
−50
−25
0
25
50
75
100
125
t − Temperature (_C)
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Si9140
Vishay Siliconix
TYPICAL CHARACTERISTICS (25_C UNLESS OTHERWISE NOTED)
VREF vs. Load Current
1.515
80
Error Amplifier Gain and Phase
0
60
1.510
Gain
Gain (dB)
3.0, 3.6 V
1.500
6.5 V
1.495
5.0 V
Phase
20
−60
0
−90
−120
−20
1.490
1.485
−150
−40
0
5
10
15
20
25
30
0.0001
0.001
VREF − Sourcing Current (mA)
260
CL = 10 pF
f = 1 MHz
100
70_C
TA = 85_C
250
1.6
70_C
Standby Current ( m A)
Normal Current (mA)
10
0.01
0.1
1
f − Frequency (MHz)
Standby Current
vs. Supply Voltage and Temperature
Supply Current
vs. Supply Voltage and Temperature
1.8
Phase (deg)
V REF (V)
−30
40
1.505
TA = 85_C
1.4
25_C
1.2
0_C
25_C
240
0_C
230
−40_C
220
−40_C
1.0
3.0
3.5
4.0
4.5
5.0
5.5
6.0
210
3.0
6.5
3.5
600
DR Sourcing Current vs. Supply Voltage
600
400
300
200
4
5.0
5.5
6.0
6.5
DR Sinking Current vs. Supply Voltage
400
300
200
3.5
4.0
4.5
5.0
5.5
VDD − Supply Voltage (V)
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4.5
500
DR Sinking Current (mA)
DR Sourcing Current (mA)
500
100
3.0
4.0
VDD − Supply Voltage (V)
VDD − Supply Voltage (V)
6.0
6.5
100
3.0
3.5
4.0
4.5
5.0
5.5
6.0
6.5
VDD − Supply Voltage (V)
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
Si9140
Vishay Siliconix
TYPICAL CHARACTERISTICS (25_C UNLESS OTHERWISE NOTED)
600
DS Sourcing vs. Supply Voltage
500
DS Sinking Current (mA)
DS Sourcing Current (mA)
500
400
300
200
100
3.0
DS Sinking Current vs. Supply Voltage
600
400
300
200
3.5
4.0
4.5
5.0
5.5
6.0
100
3.0
6.5
3.5
VDD − Supply Voltage (V)
1.2
4.0
4.5
5.0
5.5
6.0
6.5
VDD − Supply Voltage (V)
Switching Frequency vs. Supply Voltage
Frequency vs. ROSC/COSC
10.00
1.1
Switching Frequency (MHz)
Switching Frequency (MHz)
ROSC = 7.50 kW
COSC = 100 pF
1.0
0.9
1.00
4.99 kW
12.1 kW
24.9 kW
0.10
49.9 kW
100 kW
249 kW
0.8
3.0
0.01
3.5
4.0
4.5
5.0
5.5
6.0
6.5
40
200
VDD − Supply Voltage (V)
UVLO Hysteresis vs. Supply Voltage
215
60
195
UVLO Hysteresis (mV)
Output Delay (nS)
Enable Turn-OFF Delay to Output
70
50
40
30
20
3.0
300
COSC − Capacitance (pF)
175
155
135
3.5
4.0
4.5
5.0
5.5
VDD − Supply Voltage (V)
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
6.0
6.5
115
3.0
3.5
4.0
4.5
5.0
5.5
6.0
6.5
VDD − Supply Voltage (V)
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Si9140
Vishay Siliconix
TYPICAL CHARACTERISTICS (25_C UNLESS OTHERWISE NOTED)
Power Good Sinking Current (mA)
20
VGOOD Sinking Current vs. Supply Voltage
16
12
8
4
0
3.0
3.5
4.0
4.5
5.0
5.5
6.0
6.5
VDD − Supply Voltage (V)
PIN CONFIGURATIONS AND ORDERING INFORMATION
SOIC-16
TSSOP-16
VDD
1
16
VS
MON
2
15
DR
VGOOD
3
14
DS
COMP
4
13
PGND
VS
VDD
1
16
MON
2
15
DR
VGOOD
3
14
DS
COMP
4
13
PGND
FB
5
12
UVLOSET
NI
6
11
COSC
FB
5
12
UVLOSET
NI
6
11
COSC
VREF
7
10
VREF
7
10
ROSC
GND
8
9
GND
8
9
ENABLE
ROSC
ENABLE
Top View
Top View
ORDERING INFORMATION−SOIC-16
Part Number
Temperature Range
Si9140CY
Si9140CY-T1
ORDERING INFORMATIONTSSOP-16
Part Number
Si9140CQ
0_ to 70_C
Si9140CQ-T1
Si9140CY-T1—E3
Si9140CQ-T1—E3
Si9140DY
Si9140DQ
Si9140DY-T1
Si9140DY-T1—E3
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6
Temperature Range
−40_ to 85_C
Si9140DQ-T1
0_ to 70_C
−40_ to 85_C
Si9140DQ-T1—E3
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
Si9140
Vishay Siliconix
PIN DESCRIPTION
Pin 1: VDD
The positive power supply for all functional blocks except
output driver. A bypass capacitor of 0.1 mF (minimum) is
recommended.
Pin 2: MON
Non-inverting input of a comparator. Inverting input is tied
internally to reference voltage. This comparator is typically
used to monitor the output voltage and to flag the processor
when the output voltage falls out of regulation.
Pin 3: VGOOD
This is an open drain output. It will be held at ground when the
voltage at MON (Pin 2) is less than the internal reference. An
external pull-up resistor will pull this pin high if the MON pin (Pin
2) is higher than the VREF. (Refer to Pin 2 description.)
operation is disabled, supply current is reduced, the oscillator
stops and DS goes high while DR goes low.
Pin 10: ROSC
A resistor connected from this pin to ground sets the
oscillator’s capacitor COSC, charge and discharge current.
See the oscillator section of the description of operation.
Pin 11: COSC
An external capacitor is connected to this pin to set the
oscillator frequency.
fOSC ] R
0.75
COSC
OSC
(at VDD = 5.0 V)
Pin 4: COMP
Pin 12: UVLOSET
This pin is the output of the error amplifier. A compensation
network is connected from this pin to the FB pin to stabilize the
system. This pin drives one input of the internal pulse width
modulation comparator.
This pin will place the chip in the standby mode if the UVLOSET
voltage drops below 1.2 V. Once the UVLOSET voltage
exceeds 1.2 V, the chip operates normally. There is a built-in
hysteresis of 165 mV.
Pin 5: FB
The inverting input of the error amplifier. An external resistor
divider is connected to this pin to set the regulated output
voltage. The compensation network is also connected to this
pin.
Pin 6: NI
The non-inverting input of the error amplifier. In normal
operation it is externally connected to VREF or an external
reference.
Pin 13: PGND
The negative return for the VS supply.
Pin 14: DS
This CMOS push-pull output pin drives the external p-channel
MOSFET. This pin will be high in the standby mode. A
break-before-make function between DS and DR is built-in.
Pin 7: VREF
Pin 15: DR
This pin supplies a 1.5-V reference.
This CMOS push-pull output pin drives the external n-channel
MOSFET. This pin will be low in the standby mode. A
break-before-make function between the DS and DR is built-in.
Pin 8: GND (Ground)
Pin 9: ENABLE
Pin 16: VS
A logic high on this pin allows normal operation. A logic low
places the chip in the standby mode. In standby mode normal
The positive terminal of the power supply which powers the
CMOS output drivers. A bypass capacitor is required.
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
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Si9140
Vishay Siliconix
FUNCTIONAL BLOCK DIAGRAM
1.5-V Reference
Generator
VDD
UVLO
UVLOSET
VREF
VREF
VUVLO
GND
VUVLO
ENABLE
VS
COMP
VS
Error Amp
NI
+
FB
−
+
Driver
−
Logic
and
BBM
Control
COSC
PGND
PGND
VS
Driver
Oscillator
ROSC
DS
DR
PGND
VGOOD
MON
VREF
+
−
TIMING WAVEFORMS
5V
ENABLE
0V
1.5 V
VCOMP
VCOSC
1V
tBBM
DS
DR
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8
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
Si9140
Vishay Siliconix
DESCRIPTION OF OPERATION
Schematics of the Si9140 dc-to-dc conversion solutions for
high-performance PC microprocessors are shown in Figure 1
and 2 respectively. These solutions are geared to meet the
extremely demanding transient regulation and power
requirements of these new microprocessors at minimal cost
5V
(VIN)
VCCP
R3
100
+
R1
20 k
C1
2 x 220 mF
10 V
OS-CON
C4, 5.6 pF
3
R5
240 k
4
5
6
C6
0.1 mF
R6
7
4.99 k
8
C7
0.1 mF
VDD
VS
MON
DR
VGOOD
COMP
FB
NI
VREF
GND
Coiltronics
CTX07-12877
+
2.9 V
(VOUT)
C2
3 x 330 mF
6.3V OS-CON
U1
Si9140
2
D1
D1FS4
2 x Si4410DY
Power-Good
1
L1
1.5 mH
2 x Si4435DY
C3
0.1 mF
R2
10 k
R4
24.9 k
C5, 180 pF
and with a minimal parts count. The two solutions are nearly
identical, except for slight variations in output voltage, load
transient amplitude, and specified power. Figure 3 is a
schematic diagram for a 3.3-V logic converter.
16
15
14
DS
13
PGND
12
UVLOSET
11
COSC
10
ROSC
9
ENABLE
R7
100 k
R13
C8
10 k 1 mF
R12
13.3 k,
0.1%
C9
220 pF
R8
40.2 k
R9
11 k
C10, 180 pF
R10
14.2 k
0.1%
R11, 4.7 k
FIGURE 1. 2.9 V @ 10 A
5V
(VIN)
VCCP
R3
100
+
R1
20 k
C1
2 x 220 mF
10 V
OS-CON
R4
40.2 k
C4, 5.6 pF
C5, 180 pF
3
R6
C6
0.1 mF
4.99 k
C7
0.1 mF
VDD
VS
MON
DR
16
15
14
DS
4
13
PGND
COMP
5
12
UVLOSET
FB
6
11
C
NI
OSC
7
10
ROSC
VREF
9
8
ENABLE
GND
VGOOD
+
C2
3 x 330 mF 6.3V
OS-CON
U1
Si9140
2
Coiltronics
CTX07-12877
2.5 V
(VOUT)
D1
D1FS4
Si4410DY
Power-Good
1
R5
240 k
C3
0.1 mF
R2
10 k
L1
1.5 mH
2 x Si4435DY
R13
C8
10 k 1 mF
C9
220 pF
R9
11 k
R7
100 k
R12
13.3 k,
0.1%
R8
40.2 k
C10, 180 pF
R10
20 k
0.1%
R11, 4.7 k
FIGURE 2. 2.5 V @ 8.5 A
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
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Si9140
Vishay Siliconix
5V
(VIN)
R3
100
+
C3
0.1 mF
C1
2 x 220 mF
TPS
Tantalum
D1
D1FS4
Si4410DY
1
2
3
C5, 1000 pF R5
16.2 k
4
5
6
C6
0.1 mF
R6
7
4.99 k
8
C7
0.1 mF
VDD
VS
MON
DR
VGOOD
COMP
FB
NI
VREF
GND
16
15
14
DS
13
PGND
12
UVLOSET
11
COSC
10
ROSC
9
ENABLE
3.3 V
(VOUT)
+
C2
3 x 330 mF
TPS
Tantalum
U1
Si9140
C4, 330 pF
Coiltronics
CTX07-12891
L1
10 mH
Si4435DY
R13
C8
10 k 1 mF
C9
220 pF
R9
20 k
R7
100 k
R12, 13.3 k
R8
40.2 k
C10
1000 pF
R11
4.7 k
R10
11 k
FIGURE 3. 3.3 V@ 5 A
5V
(VIN)
R3
100
+
Si4435DY
C3
0.1 mF
C1
2 x 220 mF
TPS
Tantalum
Si4410DY
1
2
C5, 1000 pF
3
R5
16.2 k
4
5
6
R6
4.99 k
C6
0.1 mF
7
C7
0.1 mF
8
VDD
VS
MON
DR
VGOOD
COMP
FB
NI
VREF
GND
16
15
14
DS
13
PGND
12
UVLOSET
11
COSC
10
ROSC
9
ENABLE
Coiltronics
CTX07-12891
D1
D1FS4
+
1.5 V
(VOUT)
C2
3 x 330 mF
TPS
Tantalum
U1
Si9140
C4, 330 pF
L1
10 mH
R13
C8
10 k 1 mF
C9
220 pF
R9
20 k
R7
100 k
R12, 13.3 k
R8
40.2 k
C10
1000 pF
R11
4.7 k
FIGURE 4. 1.5-V Converter for GTL+ Bus @ 5 A
Figure 4 is a schematic diagram of a converter which produces
1.5 V for a GTL bus.
D Switch and Synchronous Rectification
MOSFETs—delivers the power to the load
Each of these dc-to-dc converters has four major sections:
D Inductor—filters and stores the energy
D PWM Controller—regulates the output voltage
D Input/Output Capacitor—filters the ripple
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10
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
Si9140
Vishay Siliconix
There are generally two types of controllers, voltage mode or
current mode. In voltage mode control, an error voltage is
generated by comparing the output voltage to the reference
voltage. The error voltage is then compared to an artificial
ramp, and the result is the duty cycle necessary to regulate the
output voltage. In current mode, an actual inductor current is
used, in place of the artificial ramp, to sense the voltage across
the current sense resistor.
The logic and timing sequence for voltage mode control is
shown in Figure 5. The Si9140 offers voltage mode control,
which is better suited for applications requiring both fast
transient response and high output current.
Current mode control requires a current sense resistor to
monitor the inductor current. A 10-mW sense resistor in a 10-A
design will dissipate 1 W, decreasing efficiency by 3.5%. Such
a design would require a 2-W resistor to satisfy derating criteria,
besides requiring additional board space. Voltage mode control
is a second-order LC system and has a faster natural transient
response compared to current mode control (first-order RC
system). Current mode has the advantage of providing an
inherently good line regulation. But the situations where line
voltage is fixed, as in the point-of-use conversion for
microprocessors, this feature is wasted. Current mode control
also provides automatic pulse-to-pulse current limiting. This
feature requires a current sense resistor as stated above. These
characteristics make voltage mode control ideal for high-end
microprocessor power supplies.
The Si9140 achieves the 5-mS transient response by
generating a 100-kHz closed-loop bandwidth. This is possible
only by switching above 400 kHz and utilizing an error amplifier
with at least a 10-MHz bandwidth. The Si9140 controller has
a 25-MHz unity gain bandwidth error amplifier. The switching
frequency must be at least four times greater than the desired
closed-loop bandwidth to prevent oscillation. To respond to
the stimuli, the error amplifier bandwidth needs to be at least
10 times larger than the desired bandwidth.
Phase
Phase (deg)
PWM Controller
The error amplifier of the PWM controller plays a major role in
determining the output voltage, stability, and the transient
response of the power supply. In the Si9140, the non-inverting
input of the error amplifier is available for use with an external
precision reference for tighter tolerance regulation. With a
two-pair lead-lag compensation network, it is easy to create a
stable 100-kHz closed loop converter with the Si9140 error
amplifier.
Gain (dB)
The functions of each circuit are explained in detail below.
Design equations are provided to optimize each application
circuit.
Gain
Frequency (Hz)
FIGURE 6. 100-kHz BW Synchronous Buck Converter
OSC
COMP
DS
DR
FIGURE 5. Voltage Mode Logic and Timing Diagram
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
The Si9140 solution requires only three 330-mF OS-CON
capacitors on the output of power supply to meet the 10-A
transient requirement. Other converter solutions on the market
with 20- to 50-kHz closed loop bandwidths typically require two
to five times the output capacitance specified above to match
the Si9140’s performance.
The theoretical issues and analytical steps involved in
compensating a feedback network are beyond the scope of
this application note. However, to ease the converter design
for today’s high-performance microprocessors, typical
component values for the feedback network are provided in
Table 1 for various combinations of output capacitance. Figure
6 shows the Bode plot (frequency domain) of the 2.9-V
converter shown schematically in Figure 1.
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11
Si9140
Vishay Siliconix
reference and 3.5% transient load regulation safely complies
with the "5% regulation requirement. If additional margin is
desired, an external precision reference can be used in place
of the internal 1.5-V reference.
TABLE 1.
FEEDBACK NETWORK COMPONENT VALUES
Total Output and
Decoupling Capacitance
C4
C5
R5
3 x 330
. . . . . . . . . Os-con
6 x 100 mFb . . . . . . . . . Tantalum
25 x 1 mFb . . . . . . . . . . Ceramic
5.6 pF
180 pF
240 k
2 x 330 mFa . . . . . . . . . Os-con
4 x 100 mFb . . . . . . . . . Tantalum
25 x 1 mFb . . . . . . . . . . Ceramic
10 pF
220 pF
200 k
3 x 330 mFa . . . . . . . . . Tantalum
4 x 100 mFb . . . . . . . . . Tantalum
25 x 1 mFb . . . . . . . . . . Ceramic
10 pF
100 pF
100 k
mFa
a.
b.
Switching and Synchronous Rectification MOSFETs
Power supply output capacitance.
mprocessor decoupling capacitance.
Figure 7 is the measured transient response (time domain) for
the 10-A step response. The measured transient response
shows the processor voltage regulating to 70 mV, well within
the 0.145-V regulation.
The Si9140’s switching frequency is determined by the
external ROSC and COSC values, allowing designers to set the
switching frequency of their choice. For applications where
space is the main constraint, the switching frequency can be
set as high as 2 MHz to minimize inductor and output capacitor
size. In applications where efficiency is the main concern, the
switching frequency can be set low to maximize battery life.
The switching frequency for high-performance processors
applications circuits are set for 400 kHz. The equation for
switching frequency is:
fOSC [
0.75
ROSC COSC
(at VDD = 5.0 V)
The precision reference is set at 1.5 V"1.5%. The reference
is capable of sourcing up to 1 mA. The combination of 1.5%
The synchronous gate drive outputs of Si9140 PWM controller
drive the high-side p-channel switch MOSFET and the
low-side n-channel synchronous rectifier MOSFET. The
physical difference between the non-synchronous to
synchronous rectification requires an additional MOSFET
across the free-wheeling diode (D1). The inductor current will
reach 0 A if the peak-to-peak inductor current equals twice the
output current. In synchronous rectification mode, current is
allowed to flow backwards from the inductor (L1) through the
synchronous MOSFET (Q3) and to the output capacitor (C2)
once the current reaches 0 A. Refer to schematic on Figure 1.
In non-synchronous rectification, the diode (D1) prevents the
current from flowing in the reverse direction. This minor
difference has a drastic affect on the performance of a power
supply. By allowing the current to flow in the reverse direction,
it preserves the continuous inductor current mode, maintaining
the wide converter bandwidth and improving efficiency. Also,
maintaining the continuous current mode during light load to
full load guarantees consistent transient response throughout
a wide range of load conditions.
The transition from stop clock and auto halt to active mode is
a perfect example. The microprocessor current can vary from
0.5 A to 10 A or greater during these transitions. If the
converter were to operate in discontinuous current mode
during the stop clock and auto halt modes, the transfer function
of the converter would be different compared to operation in
the active mode. In discontinuous current mode, the converter
bandwidth can be 10 to 15 times lower than the continuous
current mode (Figure 8). Therefore, the response time will also
be 10 to 15 times slower, violating the microprocessor’s
regulator requirements. This could result in unreliable
operation of the microprocessor.
mP
Voltage
2.9 V
mP
Current
10 A
5A
0A
a) Transient Response from 0- to 10-A Step Load
b) Transient Response from 10- to 0-A Step Load
FIGURE 7.
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12
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
Si9140
Vishay Siliconix
For these reasons, synchronous rectification is a must in
today’s microprocessors power supply design.
Pulseskipping modes are undesirable in high-performance
microprocessor power supplies, especially when the minimum
load current is as high as 500 mA. This pulse-skipping mode
disables the synchronous rectification during light load and
generates a random noise spectrum which may produce EMI
problems.
Worst case current of 10 A can be handled with two paralleled
Si4435DY and two paralleled Si4410DY MOSFETs, which
results in the efficiency levels shown in Figure 9.
100
VIN = 5 V
VOUT = 2.9 V
95
Efficiency (%)
Siliconix’ TrenchFETt technology has resulted in 20-mW
n-channel (Si4410DY) and 35-mW p-channel (Si4435DY)
MOSFETs in the SO-8 surface-mount package. These LITTLE
FOOTr products totally eliminate the need for an external
heatsink.
90
85
80
Phase
0
2
4
6
8
10
Gain
Frequency (Hz)
FIGURE 8. Non-Synchronous Converter BW
FIGURE 9. Efficiency
Phase (deg)
Gain (dB)
IOUT (A)
Good electrical designs must provide an adequate margin for
the specification, but they should not be grossly overdesigned
to lower costs. LITTLE FOOT power MOSFETs allow
designers to balance cost and performance considerations
without sacrificing either. If the design requires only an 8.5-A
continuous current, for example, one Si4410DY can be
eliminated. Table 2 shows the number of MOSFETs required
to handle the various output current levels of today’s highperformance microprocessors. For other output power levels,
the equations below should be used to calculate the power
handling capability of the MOSFET.
TABLE 2.
CONVERTER REQUIREMENTS (FIGURES 1, 2, AND 3)
IO (A)
Maximum
Quantity High-Side P-Channel
Si4435DY
Quantity Low-Side N-Channel
Si4410DY
Quantity Input (C1-C3)
Capacitor Os-con 220 mF
5A
1
1
1
8.5 A
2
1
2
10 A
2
2
2
14.5 A
3
2
3
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
www.vishay.com
13
Si9140
Vishay Siliconix
PDissipation in switch + IRMS SW 2
IRMS SW +
RSW )
Ǹǒ
IPEAK 2 ) IPP 2 ) I PEAK
Q SW
IPPǓ
V IN
2
3
RMS RECT
+
Ǹǒ
I PEAK 2 ) I PP 2 ) I PEAK
I PP
Ǔ
)
I PP
RRECT )
Q RECT
V O2
f OSC
L
IPEAK +
PIN +
VO
h
tC
V IN
2
IRMSSW
RSW
IRMSRECT
RRECT
QSW
QRECT
VIN
VO
IO
fOSC
h
tC
V IN
PIN – (0.5 VO
VO
2
fOSC
f OSC
(V IN – V )
O
3 V
IN
IPP = IPEAK + DI
DI +
VO
VO
V IN
PDissipation in synchronous rectification + IRMS RECT2
I
f OSC
DI)
IO
=
=
=
=
=
=
=
=
=
=
=
=
Switch rms current
Switch on resistance
Synchronous rectifier rms current
Synchronous rectifier on resistance
Total gate charge of switch
Total gate charge of synchronous rectifier
Input voltage
Output voltage
Output current
Switching frequency
efficiency
Crossover time
Current
IO
IPEAK
IPP
0A
time
Inductor
The size and value of the inductor are critical in meeting overall
circuit dimensional requirements and in assuring proper
transient voltage regulation. The size of the core is determined
by the output power, the material of the core, and the operating
frequency. To handle higher output power, the core must be
larger. Luckily, a higher switching frequency will lower the
inductance value, decreasing the core size. However, a higher
switching frequency can also mean greater core loss.
In applications where the dc flux density is high and the ac flux
density swing is only 100 to 200 gauss, the core loss will be
www.vishay.com
14
negligible compared to the wire loss. Kool Mu is the best
material to use at 500 kHz to deliver 30 W in the minimum
volume. Ferrite has a lower core cost and loss at this
frequency, but the core size is fairly large. If the power supply
is designed on the motherboard and space is not a critical
issue, ferrite is a better choice.
The higher switching frequency reduces the core size by
decreasing the amount of energy that must be stored between
switching periods. It also accelerates the transient response to
the load by decreasing the inductance value. The inductance
is calculated with following equation:
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
Si9140
Vishay Siliconix
L+
V IN
V O2
DI f OSC
DI = desired output current ripple. Typically DI = 25% of maximum
output current.
Finally, the time required to ramp up the current in the inductor
can be reduced with smaller inductance. A quick response
from the power supply relaxes the decoupling capacitance
required at the microprocessor, reducing the overall solution
cost and size.
Input Capacitor
The input capacitor’s function is to filter the raw power and
serve as the local power source to eliminate power-up and
transient surge failures. The type and characteristics of input
capacitors are determined by the input power and inductance
of the step-down converter. The ripple current handling
requirement usually dominates the selection criteria. The
capacitance required to maintain regulation will automatically
be achieved once it meets the ripple current requirement. The
following equation calculates the ripple current of the input
capacitor:
IRIPPLE + ǸIRMSSW 2 – I IN2
An aluminum-electrolytic capacitor from Sanyo (OS-CON),
AVX (TPS Tantalum), or Nichicon (PL series) should be used
in high-power (30-W) applications to handle the ripple current.
The Sanyo capacitor is smaller and handles higher ripple
current than Nichicon, but at higher cost than the Nichicon
product. The AVX Tantalum capacitor has the best
capacitance and current handling capability per volume ratio,
but it takes extra surface area compared to OS-CON or PL
series. The TPS capacitors, lead time and cost have
increased drastically in the recent past due to high demand,
causing designers to shy away from the TPS Tantalum
capacitors. Nichicon capacitors can be used to provide an
economical solution if space is available or a large bulk
capacitance is already present on the input line. The number
Document Number: 70026
S-40699—Rev. H, 19-Apr-04
of Sanyo (OS-CON) input capacitors required to handle
various output currents are specified in Table 2.
Output Capacitor
To regulate the microprocessor’s input voltage within 145 mV
during 10-A load transients, a large output capacitance with
low ESR is required. The output capacitor of the power supply
and decoupling capacitors at the microprocessor must hold up
the processor voltage until the power supply responds to the
change. Even with fastest known switching solution, it still
takes three 330-mF OS-CON capacitors to handle the load
transient. If it weren’t for the 10-A load transient, the output
capacitor would not need a low ESR value. The fundamental
output ripple current in a continuous step-down converter is
much lower than the input ripple current. Maintaining voltage
regulation during transients requires an ESR in the range of
30 mW.
For microprocessors with lower transient
requirements, the number of output and decoupling capacitors
can be reduced. The lower transient requirements also allows
greater consideration for Tantalum or Nichicon PL series
capacitors.
Conclusion
The Si9140 synchronous Buck controller’s ability to switch up
to 1 MHz combined with a 25-MHz error amplifier provides the
best solution in powering high- performance microprocessors.
The high switching frequency reduces inductor size without
compromising output ripple voltage. The wide converter
bandwidth generated with the help of a 25-MHz error amplifier
reduces the amount of decoupling capacitors required to
handle the extreme transient requirement. The Si9140’s
synchronous fixed-frequency operation eliminates the pulse
skipping mode that generates random unpredictable
EMI/EMC problems in desktop and notebook computers. The
synchronous rectification also allows the converter to operate
in continuous current mode, independent of output load
current. This preserves the wide closed-loop converter
bandwidth required to meet the transient demand of the
microprocessor as it transitions from stop clock and auto halt
to active mode. The synchronous rectification improves the
efficiency of the converter by substituting the much smaller I2R
MOSFET loss for the VI diode loss. The need for heatsinking
is eliminated by using low rDS(on) TrenchFETs (Si4410DY and
Si4435DY).
www.vishay.com
15
Package Information
Vishay Siliconix
SOIC (NARROW):
16-LEAD (POWER IC ONLY)
JEDEC Part Number: MS-012
MILLIMETERS
16
15
14
13
12
11
10
Dim
A
A1
B
C
D
E
e
H
L
Ĭ
9
E
1
2
3
4
5
6
7
8
INCHES
Min
Max
Min
Max
1.35
1.75
0.053
0.069
0.10
0.20
0.004
0.008
0.38
0.51
0.015
0.020
0.18
0.23
0.007
0.009
9.80
10.00
0.385
0.393
3.80
4.00
0.149
0.157
1.27 BSC
0.050 BSC
5.80
6.20
0.228
0.244
0.50
0.93
0.020
0.037
0_
8_
0_
8_
ECN: S-40080—Rev. A, 02-Feb-04
DWG: 5912
H
D
C
All Leads
e
Document Number: 72807
28-Jan-04
B
A1
L
Ĭ
0.101 mm
0.004 IN
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1
Package Information
Vishay Siliconix
TSSOP: 16-LEAD
DIMENSIONS IN MILLIMETERS
Symbols
Min
Nom
Max
A
-
1.10
1.20
A1
0.05
0.10
0.15
A2
-
1.00
1.05
0.38
B
0.22
0.28
C
-
0.127
-
D
4.90
5.00
5.10
E
6.10
6.40
6.70
E1
4.30
4.40
4.50
e
-
0.65
-
L
0.50
0.60
0.70
L1
0.90
1.00
1.10
y
-
-
0.10
θ1
0°
3°
6°
ECN: S-61920-Rev. D, 23-Oct-06
DWG: 5624
Document Number: 74417
23-Oct-06
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1
PAD Pattern
www.vishay.com
Vishay Siliconix
RECOMMENDED MINIMUM PAD FOR TSSOP-16
0.193
(4.90)
0.171
0.014
0.026
0.012
(0.35)
(0.65)
(0.30)
(4.35)
(7.15)
0.281
0.055
(1.40)
Recommended Minimum Pads
Dimensions in inches (mm)
Revision: 02-Sep-11
1
Document Number: 63550
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