SIC414CD-T1-GE3

SiC414 Vishay Siliconix microBUCKTM SiC414 6 A, 28 V Integrated Buck Regulator with 5 V LDO DESCRIPTION The Vishay Siliconix SiC414 is an advanced stand-alone synchronous buck regulator featuring integrated power MOSFETs, bootstrap switch, and an internal 5 V LDO in a space-saving MLPQ 4 x 4 - 28 pin package. The SiC414 is capable of operating with all ceramic solutions and switching frequencies up to 1 MHz. The programmable frequency, synchronous operation and selectable power-save allow operation at high efficiency across the full range of load current. The internal LDO may be used to supply 5 V for the gate drive circuits or it may be bypassed with an external 5 V for optimum efficiency and used to drive external n-channel MOSFETs or other loads. Additional features include cycle-by-cycle current limit, voltage softstart, under-voltage protection, programmable over-current protection, soft shutdown and selectable power-save. The Vishay Siliconix SiC414 also provides an enable input and a power good output. FEATURES • • • • • • • • • • • • • • High efficiency > 95 % 6 A continuous output current capability Integrated bootstrap switch Integrated 5 V/200 mA LDO with bypass logic Temperature compensated current limit Pseudo fixed-frequency adaptive on-time control All ceramic solution enabled Programmable input UVLO threshold Independent enable pin for switcher and LDO Selectable ultra-sonic power-save mode Internal soft-start and soft-shutdown 1 % internal reference voltage Power good output and over voltage protection Halogen-free according to IEC 61249-2-21 definition • Compliant to RoHS directive 2002/95/EC PRODUCT SUMMARY Input Voltage Range Output Voltage Range Operating Frequency Continuous Output Current Peak Efficiency Package 3 V to 28 V 0.75 V to 5.5 V 200 kHz to 1 MHz 6A 95 % at 300 kHz MLPQ 4 mm x 4 mm APPLICATIONS • • • • • • Notebook, desktop and server computers Digital HDTV and digital consumer applications Networking and telecommunication equipment Printers, DSL and STB applications Embedded applications Point of load power supplies TYPICAL APPLICATION CIRCUIT LDO_EN VOUT PGOOD EN/PSV (Tri-State) 3.3 V 28 27 26 25 24 TON ENL AGND EN/PSV LX 1 FB 23 22 PGOOD ILIM VOUT LX 21 LX 20 PGND 19 2 V5V 3 AGND VIN 4 VOUT 5 VIN 6 VLDO PAD1 AGND PAD3 LX PAD2 VIN P G ND PGND 18 PGND 17 PGND 16 P G ND VIN VIN VIN VIN 8 9 10 11 12 LX 7 BST LX 15 13 14 SiC414 (MLP 4 x 4-28L) Document Number: 65726 S10-1091-Rev. B, 03-May-10 www.vishay.com 1 SiC414 Vishay Siliconix PIN CONFIGURATION (TOP VIEW) EN/PSV TON ENL PGOOD 21 20 PAD3 LX PAD2 VIN 10 11 12 13 14 19 18 17 16 15 8 9 AGND 28 27 26 25 FB V5V AGND VOUT VIN VLDO BST 1 2 3 4 5 6 7 PAD1 AGND 24 23 22 LX LX PGND PGND PGND PGND LX VIN VIN VIN VIN LX PGND ILIM PIN DESCRIPTION Pin Number 1 2 3, 26, P1 4 5, 8 to 11, P2 6 7 12, 15, 20, 21, 24, P3 13, 14, 16 to 19 22 23 25 27 28 Symbol FB V5V AGND VOUT VIN VLDO BST LX PGND PGOOD ILIM EN/PSV tON ENL Description Feedback input for switching regulator. Connect to an external resistor divider from output to program the output voltage. 5 V power input for internal analog circuits and gate drives. Connect to external 5 V supply or configure the LDO for 5 V and connect to VLDO . Analog ground. Output voltage input to the SiC414. Additionally, may be used to bypass LDO to supply VLDO directly. Input supply voltage. LDO output. Bootstrap pin. A capacitor is connected between BST to LX to develop the floating voltage for the high-side gate drive. Switching (Phase) node. Power ground. Open-drain power good indicator. High impedance indicates power is good. An external pull-up resistor is required. Current limit sense point - to program the current limit connect a resistor from ILIM to LX. Tri-state pin. Enable input for switching regulator. Connect EN to AGND to disable the switching regulator. Float pin for forced continuous and pull high for power-save mode. On-time set input. Set the on-time by a series resistor to the input supply voltage. Enable input for the LDO. Connect ENL to AGND to disable the LDO. ORDERING INFORMATION Part Number SiC414CD-T1-GE3 SiC414DB Package MLPQ44-28 Evaluation board www.vishay.com 2 PGND LX Document Number: 65726 S10-1091-Rev. B, 03-May-10 SiC414 Vishay Siliconix FUNCTIONAL BLOCK DIAGRAM 2 V5V V5V 22 PGOOD 25 EN/PSV 5, 8 to 11, PAD2 VIN VIN AGND 3, 26, PAD1 Reference Control and Status V5V DL BST 7 Soft Start LX 12, 15, 20, 21, 24 PAD3 V5V PGND Zero Cross Detector Bypass Comparator VLDO 6 Y A B MUX VIN LDO ENL Valley1-Limit 13, 14, 16 to 19 ILIM 23 FB 1 TON 27 VOUT 4 + FB Comparator On-Time Generator Gate Drive Control 28 ABSOLUTE MAXIMUM RATINGS TA = 25 °C, unless otherwise noted Parameter LX to PGND Voltage LX to PGND Voltage (transient - 100 ns) VIN to PGND Voltage EN/PSV, PGOOD, ILIM, to AGND BST Bootstrap to LX; V5V to PGND AGND to PGND EN/PSV, PGOOD, ILIM, VOUT, VLDO, FB, FBL to GND tON to PGND BST to PGND VAG-PG Symbol VLX VLX VIN Min. - 0.3 -2 - 0.3 - 0.3 - 0.3 - 0.3 - 0.3 - 0.3 - 0.3 Max. + 30 + 30 + 30 V5V + 0.3 + 6.0 + 0.3 + (V5V + 0.3) + (V5V - 1.5) + 35 V Unit Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating/conditions for extended periods may affect device reliability. RECOMMENDED OPERATING CONDITIONS Parameter Input Voltage V5V to PGND VOUT to PGND Symbol VIN V5V VOUT Min. 3.0 3.0 0.75 Typ. Max. 28 5.5 5.5 V Unit Note: For proper operation, the device should be used within the recommended conditions. THERMAL RESISTANCE RATINGS Parameter Storage Temperature Maximum Junction Temperature Operation Junction Temperature Symbol TSTG TJ TJ Min. - 40 - 25 Typ. Max. + 150 150 + 125 °C Unit Document Number: 65726 S10-1091-Rev. B, 03-May-10 www.vishay.com 3 SiC414 Vishay Siliconix THERMAL RESISTANCE RATINGS Thermal Resistance, Junction-to-Ambientb High-Side MOSFET Low-Side MOSFET PWM Controller and LDO Thermal Resistance Peak IR Reflow Temperature 25 20 50 °C/W TReflow 260 °C Notes: a. This device is ESD sensitive. Use of standard ESD handling precautions is required. b. Calculated from package in still air, mounted to 3 x 4.5 (in), 4 layer FR4 PCB with thermal vias under the exposed pad per JESD51 standards. Exceeding the above specifications may result in permanent damage to the device or device malfunction. Operation outside of the parameters specififed in the Electrical Characteristics section is not recommended. ELECTRICAL SPECIFICATIONS Test Conditions Unless Specified VIN = 12 V, V5V = 5 V, TA = + 25 °C for typ., - 25 °C to + 85 °C for min. and max., TJ = < 125 °C Sensed at ENL pin, rising edge Sensed at ENL pin, falling edge EN/PSV = High Measured at V5V pin, rising edge Measured at V5V pin, falling edge EN/PSV, ENL = 0 V, VIN = 28 V VIN Supply Current IIN Standby mode: ENL = V5V, EN/PSV = 0 V EN/PSV, ENL = 0 V V5V Supply Current IV5V EN/PSV = V5V, no load (fSW = 25 kHz), VFB > 750 mVb fSW = 250 kHz, EN/PSV = floating, no load Controller FB On-Time Threshold Frequency Rangeb Bootstrap Switch Resistance Timing On-Time Minimum On-Timeb Minimum Off-Timeb Soft Start Soft Start Timeb Analog Inputs/Outputs VOUT Input Resistance Current Sense Zero-Crossing Detector Threshold Voltage Power Good Power Good Threshold Voltage Power Good Threshold Voltage Start-Up Delay Time Fault (noise-immunity) Delay Timeb Power Good Leakage Current Power Good On-Resistance PG_VTH_UPPER PG_VTH_LOWER PG_Td PG_ICC PG_ILK PG_RDS-ON VFB > internal reference 750 mV VFB < internal reference 750 mV VEN = 0 V VEN = 0 V VEN = 0 V VEN = 0 V 10 + 20 - 10 2 5 1 % ms µs µA  VSense-th LX-PGND - 3.5 0.5 + 4.7 mV RO-IN 500 k tSS IOUT = ILIM/2 1.7 ms tON tON tOFF Continuous mode operation VIN = 15 V, VOUT = 5 V, fSW = 300 kHz, Rton = 133 k 1350 1500 80 320 1650 ns VFB-TH FPWM Static VIN and load, - 40 °C to + 85 °C continuous mode 0.7425 200 10 0.750 0.7575 1000 V kHz  b Parameter Input Supplies VIN UVLO Threshold Voltagea VIN UVLO Hysteresis V5V UVLO Threshold Voltage V5V UVLO Hysteresis Symbol VIN_UV+ VIN_UVVIN_UV_HY V5V_UV+ V5V_UVV5V_UV_HY Min. 2.4 2.235 2.5 2.4 Typ. 2.6 2.4 0.2 2.9 2.7 0.2 8.5 130 3 2 10 Max. 2.95 2.565 3.0 2.9 20 Unit V µA 7 mA www.vishay.com 4 Document Number: 65726 S10-1091-Rev. B, 03-May-10 SiC414 Vishay Siliconix ELECTRICAL SPECIFICATIONS Test Conditions Unless Specified VIN = 12 V, V5V = 5 V, TA = + 25 °C for typ., - 25 °C to + 85 °C for min. and max., TJ = < 125 °C Parameter Fault Protection ILIM Source Current Valley Current Limit ILIM Comparator Offset Voltage Output Under-Voltage Fault Smart Power-Save Protection Threshold Voltageb Over-Voltage Protection Threshold Over-Voltage Fault Delayb Over Temperature Shutdownb Logic Inputs/Outputs Logic Input High Voltage Logic Input Low Voltage EN/PSV Input Bias Current ENL Input Bias Current FBL, FB Input Bias Current Linear Dropout Regulator VLDO Accuracy LDO Current Limit VLDO to VOUT Switch-Over Thresholdc Symbol ILIM Min. Typ. 8 Max. Unit µA RILIM = 5 k VILM-LK VOUV_Fault PSAVE_VTH With respect to AGND VFB with respect to Internal 500 mV reference, 8 consecutive clocks VFB with respect to internal 500 mV reference VFB with respect to internal 500 mV reference tOV-Delay TShut VIN+ VINIENFBL_ILK VLDOACC LDO_ILIM VLDO-BPS VLDO-NBPS RLDO VOUT = 5 V From VIN to VVLDO, VVLDO = + 5 V, IVLDO = 100 mA 10 °C hysteresis 3 -8 4 0 - 25 + 10 5.3 +8 A mV % % + 20 5 150 1 0.4 - 10 11 -1 4.9 134 - 140 - 450 2 1.2 5.0 85 200 + 140 + 450 + 10 18 +1 5.1 V mA mV  V µA µs °C EN, ENL, PSV EN/PSV = V5V or AGND VIN = 28 V FBL, FB = V5V or AGND VLDO load = 10 mA Start-up and foldback, VIN = 12 V Operating current limit, VIN = 12 V V VLDO to VOUT Non-Switch-Over Thresholdc VLDO to VOUT Switch-Over Resistance LDO Drop Out Voltaged Notes: a. VIN UVLO is programmable using a resistor divider from VIN to ENL to AGND. The ENL voltage is compared to an internal reference. b. Guaranteed by design. c. The switch-over threshold is the maximum voltage diff erential between the VLDO and VOUT pins which ensures that VLDO will internally switch-over to VOUT. The non-switch-over threshold is the minimum voltage diff erential between the VLDO and VOUT pins which ensures that VLDO will not switch-over to VOUT. d. The LDO drop out voltage is the voltage at which the LDO output drops 2 % below the nominal regulation point. Document Number: 65726 S10-1091-Rev. B, 03-May-10 www.vishay.com 5 SiC414 Vishay Siliconix ELECTRICAL CHARACTERISTICS 90 80 VIN = 12 V, VOUT = 1 V, FSW = 500 kHz 70 Efficiency (%) Efficiency (%) 60 50 40 30 20 40 10 0 0 1 2 3 4 5 6 7 IOUT (A) 30 0 1 2 3 4 5 6 7 IOUT (A) 70 80 VIN = 12 V, VOUT = 1 V, FSW = 500 kHz (at 6 A) 90 60 50 Efficiency vs. IOUT (in Continuous Conduction Mode) 1.05 1.04 1.03 1.02 VOUT (V) 1.01 1 0.99 0.98 0.97 0.96 0.95 0 1 2 3 4 5 6 7 IOUT (A) VIN = 12 V, VOUT = 1 V, FSW = 500 kHz VOUT (V) 1.05 1.04 1.03 Efficiency vs. IOUT (in Power-Save-Mode) VIN = 12 V, VOUT = 1 V, FSW = 500 kHz (at 6 A) 1.02 1.01 1 0.99 0.98 0.97 0.96 0.95 0 1 2 3 4 5 6 7 IOUT (A) VOUT vs. IOUT (in Continuous Conduction Mode) 1.05 1.04 1.03 1.02 VOUT (V) VOUT (V) 1.01 1 0.99 0.98 0.97 0.96 0.95 3 6 9 12 15 18 21 24 VIN (V) VOUT = 1 V, IOUT = 0 A 1.05 1.04 1.03 1.02 VOUT vs. IOUT (in Power-Save-Mode) VOUT = 1 V, IOUT = 6 A 1.01 1 0.99 0.98 0.97 0.96 0.95 3 6 9 12 15 18 21 24 VIN (V) VOUT vs. VIN at IOUT = 0 A (in Continuous Conduction Mode, FSW = 500 kHz) www.vishay.com 6 VOUT vs. VIN at IOUT = 6 A (in Continuous Conduction Mode, FSW = 500 kHz) Document Number: 65726 S10-1091-Rev. B, 03-May-10 SiC414 Vishay Siliconix ELECTRICAL CHARACTERISTICS 1.05 1.04 1.03 VOUT = 1 V, IOUT = 0 A 1.01 1 0.99 0.98 0.97 0.96 0.95 3 6 9 12 15 18 21 24 VIN (V) VOUT Ripple (mV) 1.02 VOUT (V) 35 30 25 20 15 10 5 0 0 5 10 VIN (V) 15 20 25 VOUT = 1 V, IOUT = 6 A, FSW = 500 kHz 50 45 40 VOUT vs. VIN (IOUT = 0 A in Power-Save-Mode) 50 45 40 VOUT = 1 V, IOUT = 0 A, FSW = 500 kHz VOUT Ripple (mV) 30 25 20 15 10 5 0 0 5 10 VIN (V) 15 20 25 VOUT Ripple (mV) 35 35 30 25 20 15 10 5 0 0 50 45 40 VOUT Ripple vs. VIN (IOUT = 6 A in Continuous Conduction Mode) VOUT = 1 V, IOUT = 0 A, PSV Mode 5 10 VIN (V) 15 20 25 VOUT Ripple vs. VIN (IOUT = 0 A in Continuous Conduction Mode) 550 525 500 370 FSW (kHz) FSW (kHz) 475 450 425 400 120 375 350 0 1 2 3 4 IOUT (A) 5 6 7 70 20 0 VIN = 12 V, VOUT = 1 V, FSW = 500 kHz (at 6 A) 320 270 220 170 520 470 420 VOUT Ripple vs. VIN (IOUT = 0 A in Power-Save-Mode) VIN = 12 V, VOUT = 1 V, FSW = 500 kHz (at 6 A) 1 2 3 4 5 6 7 IOUT (A) FSW vs. IOUT (in Continuous Conduction Mode) Document Number: 65726 S10-1091-Rev. B, 03-May-10 FSW vs. IOUT (in Power-Save-Mode) www.vishay.com 7 SiC414 Vishay Siliconix ELECTRICAL CHARACTERISTICS VOUT Ripple in Continuous Conduction Mode (No Load) (VIN = 12 V, VOUT = 1 V, FSW = 500 kHz) VOUT Ripple in Power Save Mode (No Load) (VIN = 12 V, VOUT = 1 V) Output Current 2 A/div. 5 µs/div. Output Voltage 50 mV/div. 5 µs/div. AC Coupling Output Current 2 A/div. 5 µs/div. Output Voltage 50 mV/div. 5 µs/div. AC Coupling Transient Response in Continuous Conduction Mode (0.2 A - 6 A) (VIN = 12 V, VOUT = 1 V, FSW = 500 kHz) Transient Response in Continuous Conduction Mode (6 A - 0.2 A) (VIN = 12 V, VOUT = 1 V, FSW = 500 kHz) Output Current 2 A/div. 5 µs/div. Output Voltage 50 mV/div. 5 µs/div. AC Coupling Output Current 2 A/div. 5 µs/div. Output Voltage 50 mV/div. 5 µs/div. AC Coupling Transient Response in Power Save Mode (0.2 A - 6 A) (VIN = 12 V, VOUT = 1 V, FSW = 500 kHz at 6A) Transient Response in Power Save Mode (6 A - 0.2 A) (VIN = 12 V, VOUT = 1 V, FSW = 500 kHz at 6 A) www.vishay.com 8 Document Number: 65726 S10-1091-Rev. B, 03-May-10 SiC414 Vishay Siliconix ELECTRICAL CHARACTERISTICS Start-up with VIN Ramping up (VIN = 12 V, VOUT = 1 V, FSW = 500 kHz) Over-Current Protection (VIN = 12 V, VOUT = 1 V, FSW = 500 kHz) APPLICATIONS INFORMATION SiC414 Synchronous Buck Converter The SiC414 is a step down synchronous buck dc-to-dc converter with integrated power FETs and programmable LDO. The SiC414 is capable of 6 A operation at very high efficiency in a tiny 4 mm x 4 mm - 28 pin package. The programmable operating frequency range of 200 kHz to 1 MHz, enables the user to optimize the solution for minimum board space and optimum efficiency. The buck controller employs pseudo-fixed frequency adaptive on-time control. This control scheme allows fast transient response thereby lowering the size of the power components used in the system. Input Voltage Range The SiC414 requires two input supplies for normal operation: VIN and V5V. VIN operates over the wide range from 3 V to 28 V. V5V requires a 5 V supply input that can be an external source or the internal LDO configured to supply 5 V. Pseudo-Fixed Frequency Adaptive On-Time Control The PWM control method used for the SiC414 is pseudo-fixed frequency, adaptive on-time, as shown in figure 1. The ripple voltage generated at the output capacitor ESR is used as a PWM ramp signal. This ripple is used to trigger the on-time of the controller. The adaptive on-time is determined by an internal oneshot timer. When the one-shot is triggered by the output ripple, the device sends a single on-time pulse to the highside MOSFET. The pulse period is determined by VOUT and VIN; the period is proportional to output voltage and inversely proportional to input voltage. With this adaptive on-time arrangement, the device automatically anticipates the on-time needed to regulate VOUT for the present VIN condition and at the selected frequency. VIN tON VLX CIN Q1 VLX L Q2 ESR FB + COUT VFB FB threshold VOUT Figure 1 - Output Ripple and PWM Control Method The adaptive on-time control has significant advantages over traditional control methods used in the controllers today. • Reduced component count by eliminating DCR sense or current sense resistor as no need of a sensing inductor current. • Reduced saves external components used for compensation by eliminating the no error amplifier and other components. • Ultra fast transient response because of fast loop, absence of error amplifier speeds up the transient response. • Predictable frequency spread because of constant on-time architecture. • Fast transient response enables operation with minimum output capacitance Overall, superior performance compared to fixed frequency architectures. Document Number: 65726 S10-1091-Rev. B, 03-May-10 www.vishay.com 9 SiC414 Vishay Siliconix On-Time One-Shot Generator (tON) and Operating Frequency The SiC414 have an internal on-time one-shot generator which is a comparator that has two inputs. The FB Comparator output goes high when VFB is less than the internal 750 mV reference. This feeds into the gate drive and turns on the high-side MOSFET, and also starts the one-shot timer. The one-shot timer uses an internal comparator and a capacitor. One comparator input is connected to VOUT, the other input is connected to the capacitor. When the on-time begins, the internal capacitor charges from zero volts through a current which is proportional to VIN. When the capacitor voltage reaches VOUT, the on-time is completed and the high-side MOSFET turns off. The figure 2 shows the on-chip implementation of on-time generation. FB comparator FB 750 mV + Gate drives Q1 VLX VOUT R1 R2 To FB pin Figure 3 - Output Voltage Selection As the control method regulates the valley of the output ripple voltage, the DC output voltage VOUT is off set by the output ripple according to the following equation. VOUT = 0.75 x (1 + R1/R2) + VRIPPLE/2 Enable and Power-Save Inputs The EN/PSV and ENL inputs are used to enable or disable the switching regulator and the LDO. When EN/PSV is low (grounded), the switching regulator is off and in its lowest power state. When off, the output of the switching regulator soft-discharges the output into a 15  internal resistor via the VOUT pin. When EN/PSV is allowed to float, the pin voltage will float to 1.5 V. The switching regulator turns on with power-save disabled and all switching is in forced continuous mode. When EN/PSV is high (above 2.0 V), the switching regulator turns on with ultra-sonic power-save enabled. The SiC414 ultra-sonic power-save operation maintains a minimum switching frequency of 25 kHz, for applications with stringent audio requirements. The ENL input is used to control the internal LDO. This input serves a second function by acting as a VIN UVLO sensor for the switching regulator. The LDO is off when ENL is low (grounded). When ENL is a logic high but below the VIN UVLO threshold (2.6 V typical), then the LDO is on and the switcher is off. When ENL is above the VIN UVLO threshold, the LDO is enabled and the switcher is also enabled if the EN/PSV pin is not grounded. Forced Continuous Mode Operation The SiC414 operates the switcher in Forced Continuous Mode (FCM) by floating the EN/PSV pin (see figure 4). In this mode one of the power MOSFETs is always on, with no intentional dead time other than to avoid cross-conduction. This feature results in uniform frequency across the full load range with the trade-off being poor efficiency at light loads due to the high-frequency switching of the MOSFETs. DH VOUT VIN Rton L ESR VOUT FB + One-shot timer DL Q2 COUT On-time = K x Rton x (VOUT/VIN) Figure 2 - On-Time Generation This method automatically produces an on-time that is proportional to VOUT and inversely proportional to VIN. Under steady-state conditions, the switching frequency can be determined from the on-time by the following equation. fSW = VOUT tON x VIN The SiC414 uses an external resistor to set the ontime which indirectly sets the frequency. The on-time can be programmed to provide operating frequency from 200 kHz to 1 MHz using a resistor between the tON pin and ground. The resistor value is selected by the following equation. (t - 10 ns) x VIN Rton = ON 25 pF x VOUT The maximum RTON value allowed is shown by the following equation. Rton_MAX = VIN_MIN 15 µA VOUT Voltage Selection The switcher output voltage is regulated by comparing VOUT as seen through a resistor divider at the FB pin to the internal 750 mV reference voltage, see figure 3. www.vishay.com 10 Document Number: 65726 S10-1091-Rev. B, 03-May-10 SiC414 Vishay Siliconix FB ripple voltage (VFB) FB threshold (750 mV) Because the on-times are forced to occur at intervals no greater than 40 µs, the frequency will not fall below ~ 25 kHz. Figure 5 shows ultra-sonic power-save operation. Benefits of Ultrasonic Power-Save Having a fixed minimum frequency in power-save has some significant advantages as below: • The minimum frequency of 25 kHz is outside the audible range of human ear. This makes the operation of the SiC414 very quiet. • The output voltage ripple seen in power-save mode is significant lower than conventional power-save, which improves efficiency at light loads. • Lower ripple in power-save also makes the power component selection easier. Inductor current DC load current On-time (tON) DH on-time is triggered when VFB reaches the FB threshold DH DL DL drives high when on-time is completed. DL remains high until VFB falls to the FB threshold. Figure 4 - Forced Continuous Mode Operation Ultrasonic Power-Save Operation The SiC414 provides ultra-sonic power-save operation at light loads, with the minimum operating frequency fixed at 25 kHz. This is accomplished using an internal timer that monitors the time between consecutive high-side gate pulses. If the time exceeds 40 µs, DL drives high to turn the low-side MOSFET on. This draws current from VOUT through the inductor, forcing both VOUT and VFB to fall. When VFB drops to the 750 mV threshold, the next DH on-time is triggered. After the on-time is completed the high-side MOSFET is turned off and the low-side MOSFET turns on, the low-side MOSFET remains on until the inductor current ramps down to zero, at which point the low-side MOSFET is turned off. minimum fSW ~ 25 kHz FB ripple voltage (VFB) FB threshold (750 mV) Inductor current (0 A) Figure 6 - Ultrasonic Power-Save Operation Mode Figure 6 shows the behavior under power-save and continuous conduction mode at light loads. Smart Power-Save Protection Active loads may leak current from a higher voltage into the switcher output. Under light load conditions with power-savepower-save enabled, this can force VOUT to slowly rise and reach the over-voltage threshold, resulting in a hard shutdown. Smart power-save prevents this condition. When the FB voltage exceeds 10 % above nominal (exceeds 825 mV), the device immediately disables power-save, and DL drives high to turn on the low-side MOSFET. This draws current from VOUT through the inductor and causes VOUT to fall. When VFB drops back to the 750 mV trip point, a normal tON switching cycle begins. This method prevents a hard OVP shutdown and also cycles energy from VOUT back to VIN. It also minimizes operating power by avoiding forced conduction mode operation. Figure 7 shows typical waveforms for the smart power-save feature. On-time (tON) DH DH on-time is triggered when VFB reaches the FB threshold DL After the 40 µs time-out, DL drives high if VFB has not reached the FB threshold. Figure 5 - Ultrasonic power-save Operation Document Number: 65726 S10-1091-Rev. B, 03-May-10 www.vishay.com 11 SiC414 Vishay Siliconix VOUT drifts up to due to leakage current flowing into COUT Smart power save threshold (825 mV) FB threshold DH and DL off High-side drive (DH) Single DH on-time pulse after DL turn-off Low-side drive (DL) DL turns on when smart PSAVE threshold is reached DL turns off FB threshold is reached Normal DL pulse after DH on-time pulse VOUT discharges via inductor and low-side MOSFET Normal VOUT ripple Note that because the low-side MOSFET with low RDS(ON) is used for current sensing, the PCB layout, solder connections, and PCB connection to the LX node must be done carefully to obtain good results. Refer to the layout guidelines for information. Soft-Start of PWM Regulator Soft-start is achieved in the PWM regulator by using an internal voltage ramp as the reference for the FB Comparator. The voltage ramp is generated using an internal charge pump which drives the reference from zero to 750 mV in ~ 1.2 mV increments, using an internal ~ 500 kHz oscillator. When the ramp voltage reaches 750 mV, the ramp is ignored and the FB comparator switches over to a fixed 750 mV threshold. During soft-start the output voltage tracks the internal ramp, which limits the start-up inrush current and provides a controlled softstart profile for a wide range of applications. Typical softstart ramp time is 850 µs. During soft-start the regulator turns off the low-side MOSFET on any cycle if the inductor current falls to zero. This prevents negative inductor current, allowing the device to start into a pre-biased output. Power Good Output The power good (PGOOD) output is an open-drain output which requires a pull-up resistor. When the output voltage is 10 % below the nominal voltage, PGOOD is pulled low. It is held low until the output voltage returns above - 8 % of nominal. PGOOD is held low during start-up and will not be allowed to transition high until soft-start is completed (when VFB reaches 750 mV) and typically 2 ms has passed. PGOOD will transition low if the VFB pin exceeds + 20 % of nominal, which is also the over-voltage shutdown threshold (900 mV). PGOOD also pulls low if the EN/PSV pin is low when V5V is present. Output Over-Voltage Protection Over-voltage protection becomes active as soon as the device is enabled. The threshold is set at 750 mV + 20 % (900 mV). When VFB exceeds the OVP threshold, DL latches high and the low-side MOSFET is turned on. DL remains high and the controller remains off , until the EN/PSV input is toggled or V5V is cycled. There is a 5 µs delay built into the OVP detector to prevent false transitions. PGOOD is also low after an OVP event. Output Under-Voltage Protection When VFB falls 25 % below its nominal voltage (falls to 562.5 mV) for eight consecutive clock cycles, the switcher is shut off and the DH and DL drives are pulled low to tristate the MOSFETs. The controller stays off until EN/PSV is toggled or V5V is cycled. V5V UVLO, and POR Under-voltage lock-out (UVLO) circuitry inhibits switching and tri-states the DH/DL drivers until V5V rises above 3.9 V. An internal Power-On Reset (POR) occurs when V5V exceeds 3.9 V, which resets the fault latch and soft-start Document Number: 65726 S10-1091-Rev. B, 03-May-10 Figure 7 - Smart Power-Save Current Limit Protection The SiC414 features programmable current limit capability, which is accomplished by using the RDS(ON) of the lower MOSFET for current sensing. The current limit is set by RILIM resistor. The RILIM resistor connects from the ILIM pin to the LX pin which is also the drain of the low-side MOSFET. When the low-side MOSFET is on, an internal ~ 10 µA current flows from the ILIM pin and the RILIM resistor, creating a voltage drop across the resistor. While the low-side MOSFET is on, the inductor current flows through it and creates a voltage across the RDS(ON). The voltage across the MOSFET is negative with respect to ground. If this MOSFET voltage drop exceeds the voltage across RILIM, the voltage at the ILIM pin will be negative and current limit will activate. The current limit then keeps the low-side MOSFET on and will not allow another high-side on-time, until the current in the low-side MOSFET reduces enough to bring the ILIM voltage back up to zero. This method regulates the inductor valley current at the level shown by ILIM in figure 8. IPEAK Inductor Current ILOAD ILIM Time Figure 8 - Valley Current Limit Setting the valley current limit to 6 A results in a 6 A peak inductor current plus peak ripple current. In this situation, the average (load) current through the inductor is 6 A plus onehalf the peak-to-peak ripple current. The internal 10 µA current source is temperature compensated at 4100 ppm in order to provide tracking with the RDS(ON). The RILIM value is calculated by the following equation. RILIM = 1250 x ILIM x [0.088 x (5 V - V5V) + 1] www.vishay.com 12 SiC414 Vishay Siliconix counter to prepare for soft-start. The SiC414 then begins a soft-start cycle. The PWM will shut off if V5V falls below 3.6 V. LDO Regulator The device features an integrated LDO regulator with a fixed output voltage of 5 V. There is also an enable pin (ENL) for the LDO that provides independent control. The LDO voltage can also be used to provide the bias voltage for the switching regulator. A minimum capacitance of 1 µF referenced to AGND is normally required at the output of the LDO for stability. If the LDO is providing bias power to the device, then a minimum 0.1 µF capacitor referenced to AGND is required, along with a minimum 1 µF capacitor referenced to PGND to filter the gate drive pulses. Refer to the layout guidelines section. LDO Start-up Before start-up, the LDO checks the status of the following signals to ensure proper operation can be maintained. 1. ENL pin 2. VLDO output 3. VIN input voltage When the ENL pin is high and VIN is above the UVLO point, the LDO will begin start-up. During the initial phase, when the LDO output voltage is near zero, the LDO initiates a current-limited start-up (typically 85 mA) to charge the output capacitor. When VLDO has reached 90 % of the final value (as sensed at the FBL pin), the LDO current limit is increased to ~ 200 mA and the LDO output is quickly driven to the nominal value by the internal LDO regulator. VVLDO final 90 % of VVLDO final Voltage regulating with ~ 200 mA current limit In the first method, the LDO is already in regulation and the dc-to-dc converter is later enabled. As soon as the PGOOD output goes high, the 32 cycles are started. The voltages at the VLDO and VOUT pins are then compared; if the two voltages are within ± 300 mV of each other, the VLDO pin connects to the VOUT pin using an internal switch, and the LDO is turned off. In the second method, the dc-to-dc converter is already running and the LDO is enabled. In this case the 32 cycles are started as soon as the LDO reaches 90 % of its final value. At this time, the VLDO and VOUT pins are compared, and if within ± 300 mV the switch-over occurs and the LDO is turned off. Benefits of having a switchover circuit The switchover function is designed to get maximum efficiency out of the dc-to-dc converter. The efficiency for an LDO is very low especially for high input voltages. Using the switchover function we tie any rails connected to VLDO through a switch directly to VOUT. Once switchover is complete LDO is turned off which saves power. This gives us the maximum efficiency out of the SiC414. If the LDO output is used to bias the SiC414, then after switchover the VOUT self biases the SiC414 and operates in self-powered mode. Steps to follow when using the on chip LDO to bias the SiC414: • Always tie the V5V to VLDO before enabling the LDO • Enable the LDO before enabling the switcher • LDO has a current limit of 40 mA at start-up, so do not connect any load between VLDO and ground • The current limit for the LDO goes up to 200 mA once the VLDO reaches 90 % of its final values and can easily supply the required bias current to the IC. Switch-over Limitations on VOUT and VLDO Because the internal switch-over circuit always compares the VOUT and VLDO pins at start-up, there are limitations on permissible combinations of VOUT and VLDO. Consider the case where VOUT is programmed to 1.5 V and VLDO is programmed to 1.8 V. After start-up, the device would connect VOUT to VLDO and disable the LDO, since the two voltages are within the ± 300 mV switch-over window. To avoid unwanted switch-over, the minimum difference between the voltages for VOUT and VLDO should be ± 500 mV. It is not recommended to use the switch-over feature for an output voltage less than 3 V since this does not provide sufficient voltage for the gate-source drive to the internal p-channel switch-over MOSFET. Switch-Over MOSFET Parasitic Diodes The switch-over MOSFET contains parasitic diodes that are inherent to its construction, as shown in figure 10. Constant current startup Figure 9 - LDO Start-Up LDO Switchover Function The SiC414 includes a switch-over function for the LDO. The switch-over function is designed to increase efficiency by using the more efficient dc-to-dc converter to power the LDO output, avoiding the less efficient LDO regulator when possible. The switch-over function connects the VLDO pin directly to the VOUT pin using an internal switch. When the switch-over is complete the LDO is turned off, which results in a power savings and maximizes efficiency. If the LDO output is used to bias the SiC414, then after switch-over the device is self-powered from the switching regulator with the LDO turned off. The switch-over logic waits for 32 switching cycles before it starts the switch-over. There are two methods that determine the switch-over of VLDO to VOUT. Document Number: 65726 S10-1091-Rev. B, 03-May-10 www.vishay.com 13 SiC414 Vishay Siliconix Switchover control VLDO Switchover MOSFET VOUT will not allow any PWM switching until the LDO output has reached 90 % of it's final value. On-Chip LDO Bias the SiC414 The following steps must be followed when using the onchip LDO to bias the device. • Connect V5V to VLDO before enabling the LDO. • The LDO has an initial current limit of 40 mA at start-up, therefore, do not connect any external load to VLDO during start-up. • When VLDO reaches 90 % of its final value, the LDO current limit increases to 200 mA. At this time the LDO may be used to supply the required bias current to the device. Attempting to operate in self-powered mode in any other configuration can cause unpredictable results and may damage the device. Design Procedure When designing a switch mode power supply, the input voltage range, load current, switching frequency, and inductor ripple current must be specified. The maximum input voltage (VINMAX) is the highest specified input voltage. The minimum input voltage (VINMIN) is determined by the lowest input voltage after evaluating the voltage drops due to connectors, fuses, switches, and PCB traces. The following parameters define the design: • Nominal output voltage (VOUT) • Static or DC output tolerance • Transient response • Maximum load current (IOUT) There are two values of load current to evaluate - continuous load current and peak load current. Continuous load current relates to thermal stresses which drive the selection of the inductor and input capacitors. Peak load current determines instantaneous component stresses and filtering requirements such as inductor saturation, output capacitors, and design of the current limit circuit. The following values are used in this design: • VIN = 12 V ± 10 % • VOUT = 1.05 V ± 4 % • fSW = 250 kHz • Load = 6 A maximum Frequency Selection Selection of the switching frequency requires making a trade-off between the size and cost of the external filter components (inductor and output capacitor) and the power conversion efficiency. The desired switching frequency is 250 kHz which results from using component selected for optimum size and cost. A resistor (RTON) is used to program the on-time (indirectly setting the frequency) using the following equation. (t - 10 ns) x VIN Rton = ON 25 pF x VOUT Parastic diode V5V Parastic diode Figure 10 - Switch-over MOSFET Parasitic Diodes There are some important design rules that must be followed to prevent forward bias of these diodes. The following two conditions need to be satisfied in order for the parasitic diodes to stay off. • V5V  VLDO • V5V  VOUT If either VLDO or VOUT is higher than V5V, then the respective diode will turn on and the SiC414 operating current will flow through this diode. This has the potential of damaging the device. ENL Pin and VIN UVLO The ENL pin also acts as the switcher under-voltage lockout for the VIN supply. The VIN UVLO voltage is programmable via a resistor divider at the VIN, ENL and AGND pins. ENL is the enable/disable signal for the LDO. In order to implement the VIN UVLO there is also a timing requirement that needs to be satisfied. If the ENL pin transitions low within 2 switching cycles and is < 0.4 V, then the LDO will turn off but the switcher remains on. If ENL goes below the VIN UVLO threshold and stays above 1 V, then the switcher will turn off but the LDO remains on. The VIN UVLO function has a typical threshold of 2.6 V on the VIN rising edge. The falling edge threshold is 2.4 V. Note that it is possible to operate the switcher with the LDO disabled, but the ENL pin must be below the logic low threshold (0.4 V maximum). ENL Logic Control of PWM Operation When the ENL input is driven above 2.6 V, it is impossible to determine if the LDO output is going to be used to power the device or not. In self-powered operation where the LDO will power the device, it is necessary during the LDO start-up to hold the PWM switching off until the LDO has reached 90 % of the final value. This is to prevent overloading the current-limited LDO output during the LDO start-up. However, if the switcher was previously operating (with EN/ PSV high but ENL at ground, and V5V supplied externally), then it is undesirable to shut down the switcher. To prevent this, when the ENL input is taken above 2.6 V (above the VIN UVLO threshold), the internal logic checks the PGOOD signal. If PGOOD is high, then the switcher is already running and the LDO will run through the start-up cycle without affecting the switcher. If PGOOD is low, then the LDO www.vishay.com 14 Document Number: 65726 S10-1091-Rev. B, 03-May-10 SiC414 Vishay Siliconix To select RTON, use the maximum value for VIN, and for tON use the value associated with maximum VIN. tON = VOUT VINMAX. x fSW tON = 318 ns at 13.2 VIN, 1.05 VOUT, 250 kHz Substituting for RTON results in the following solution RTON = 154.9 k, use RTON = 154 k. Inductor Selection In order to determine the inductance, the ripple current must first be defined. Low inductor values result in smaller size but create higher ripple current which can reduce efficiency. Higher inductor values will reduce the ripple current and voltage and for a given DC resistance are more efficient. However, larger inductance translates directly into larger packages and higher cost. Cost, size, output ripple, and efficiency are all used in the selection process. The ripple current will also set the boundary for power-save operation. The switching will typically enter power-save mode when the load current decreases to 1/2 of the ripple current. For example, if ripple current is 4 A then power-save operation will typically start for loads less than 2 A. If ripple current is set at 40 % of maximum load current, then powersave will start for loads less than 20 % of maximum current. The inductor value is typically selected to provide a ripple current that is between 25 % to 50 % of the maximum load current. This provides an optimal trade-off between cost, efficiency, and transient performance. During the DH on-time, voltage across the inductor is (VIN - VOUT). The equation for determining inductance is shown next. L= (VIN - VOUT) x tON IRIPPLE Capacitor Selection The output capacitors are chosen based on required ESR and capacitance. The maximum ESR requirement is controlled by the output ripple requirement and the DC tolerance. The output voltage has a DC value that is equal to the valley of the output ripple plus 1/2 of the peak-to-peak ripple. Change in the output ripple voltage will lead to a change in DC voltage at the output. The design goal is that the output voltage regulation be ± 4 % under static conditions. The internal 500 mV reference tolerance is 1 %. Allowing 1 % tolerance from the FB resistor divider, this allows 2 % tolerance due to VOUT ripple. Since this 2 % error comes from 1/2 of the ripple voltage, the allowable ripple is 4 %, or 42 mV for a 1.05 V output. The maximum ripple current of 4.4 A creates a ripple voltage across the ESR. The maximum ESR value allowed is shown by the following equations. ESRMAX = VRIPPLE IRIPPLEMAX = 42 mV 2.9 A ESRMAX = 9.5 mΩ The output capacitance is usually chosen to meet transient requirements. A worst-case load release, from maximum load to no load at the exact moment when inductor current is at the peak, determines the required capacitance. If the load release is instantaneous (load changes from maximum to zero in < 1 µs), the output capacitor must absorb all the inductor's stored energy. This will cause a peak voltage on the capacitor according to the following equation. L (IOUT + COUT_MIN = 1 xI )2 2 RIPPLEMAX (VPEAK)2 - (VOUT)2 Example In this example, the inductor ripple current is set equal to 50 % of the maximum load current. Thus ripple current will be 50 % x 6 A or 3 A. To find the minimum inductance needed, use the VIN and TON values that correspond to VINMAX. L= (13.2 - 1.05) x 318 ns = 1.28 µH 3A Assuming a peak voltage VPEAK of 1.150 (100 mV rise upon load release), and a 10 A load release, the required capacitance is shown by the next equation. 1.3 µH (6 + COUT_MIN = 1 x 2.9)2 2 (1.15)2 - (1.05)2 COUT_MIN = 328 µF A slightly larger value of 1.3 µH is selected. This will decrease the maximum IRIPPLE to 2.9 A. Note that the inductor must be rated for the maximum DC load current plus 1/2 of the ripple current. The ripple current under minimum VIN conditions is also checked using the following equations. TON_VINMIN = 25 pF x RTON x VOUT VINMIN IRIPPLE = (VIN - VOUT) x TON L (10.8 - 1.05) x 384 ns = 2.88 A 1.3 µH If the load release is relatively slow, the output capacitance can be reduced. At heavy loads during normal switching, when the FB pin is above the 750 mV reference, the DL output is high and the low-side MOSFET is on. During this time, the voltage across the inductor is approximately - VOUT. This causes a down-slope or falling di/dt in the inductor. If the load dI/dt is not much faster than the - dI/dt in the inductor, then the inductor current will tend to track the falling load current. This will reduce the excess inductive energy that must be absorbed by the output capacitor, therefore a smaller capacitance can be used. IRIPPLE_VIN = Document Number: 65726 S10-1091-Rev. B, 03-May-10 www.vishay.com 15 SiC414 Vishay Siliconix The following can be used to calculate the needed capacitance for a given dILOAD/dt: Peak inductor current is shown by the next equation. ILPK = IMAX + 1/2 x IRIPPLEMAX ILPK = 6 + 1/2 x 2.9 = 7.45 A Rate of change of load current = dILOAD/dt IMAX = maximum load release = 6 A I I L x LPK - MAX x dt VOUT dlLOAD COUT = ILPK x 2 (VPK - VOUT) CTOP VOUT R1 R2 To FB pin Example Figure 11 - Capacitor Coupling to FB Pin Load dlLOAD 2.5 A = µs dt This would cause the output current to move from 10 A to zero in 4 µs as shown by the following equation. 1.3 µH x 7.45 6 x 1 µs 1.05 2.5 2 (1.15 - 1.05) COUT = 7.45 x COUT = 254 µF Note that COUT is much smaller in this example, 254 µF compared to 328 µF based on a worst-case load release. To meet the two design criteria of minimum 254 µF and maximum 9 m ESR, select two capacitors rated at 150 µF and 18 m ESR. It is recommended that an additional small capacitor be placed in parallel with COUT in order to filter high frequency switching noise. Stability Considerations Unstable operation is possible with adaptive on-time controllers, and usually takes the form of double-pulsing or ESR loop instability. Double-pulsing occurs due to switching noise seen at the FB input or because the FB ripple voltage is too low. This causes the FB comparator to trigger prematurely after the 250 ns minimum off-time has expired. In extreme cases the noise can cause three or more successive on-times. Double-pulsing will result in higher ripple voltage at the output, but in most applications it will not affect operation. This form of instability can usually be avoided by providing the FB pin with a smooth, clean ripple signal that is at least 10 mVp-p, which may dictate the need to increase the ESR of the output capacitors. It is also imperative to provide a proper PCB layout as discussed in the Layout Guidelines section. Another way to eliminate doubling-pulsing is to add a small (~ 10 pF) capacitor across the upper feedback resistor, as shown in figure 11. This capacitor should be left unpopulated until it can be confirmed that double-pulsing exists. Adding the CTOP capacitor will couple more ripple into FB to help eliminate the problem. An optional connection on the PCB should be available for this capacitor. ESR loop instability is caused by insufficient ESR. The details of this stability issue are discussed in the ESR Requirements section. The best method for checking stability is to apply a zero-to-full load transient and observe the output voltage ripple envelope for overshoot and ringing. Ringing for more than one cycle after the initial step is an indication that the ESR should be increased. One simple way to solve this problem is to add trace resistance in the high current output path. A side effect of adding trace resistance is output decreased load regulation. ESR Requirements A minimum ESR is required for two reasons. One reason is to generate enough output ripple voltage to provide10 mVp-p at the FB pin (after the resistor divider) to avoid doublepulsing. The second reason is to prevent instability due to insufficient ESR. The on-time control regulates the valley of the output ripple voltage. This ripple voltage is the sum of the two voltages. One is the ripple generated by the ESR, the other is the ripple due to capacitive charging and discharging during the switching cycle. For most applications the minimum ESR ripple voltage is dominated by the output capacitors, typically SP or POSCAP devices. For stability the ESR zero of the output capacitor should be lower than approximately one-third the switching frequency. The formula for minimum ESR is shown by the following equation. ESRMIN = 3 2 x π x COUT x fSW For applications using ceramic output capacitors, the ESR is normally too small to meet the above ESR criteria. In these applications it is necessary to add a small virtual ESR network composed of two capacitors and one resistor, as shown in figure 12. This network creates a ramp voltage www.vishay.com 16 Document Number: 65726 S10-1091-Rev. B, 03-May-10 SiC414 Vishay Siliconix across CL, analogous to the ramp voltage generated across the ESR of a standard capacitor. This ramp is then capacitive-coupled into the FB pin via capacitor CC. L Highside RL CL R1 Lowside CC COUT This trace resistance should be optimized so that at full load the output droops to near the lower regulation limit. Passive droop minimizes the required output capacitance because the voltage excursions due to load steps are reduced as seen at the load. The use of 1 % feedback resistors contributes up to 1 % error. If tighter DC accuracy is required, 0.1 % resistors should be used. The output inductor value may change with current. This will change the output ripple and therefore will have a minor effect on the DC output voltage. The output ESR also affects the output ripple and thus has a minor effect on the DC output voltage. Switching Frequency Variations The switching frequency will vary depending on line and load conditions. The line variations are a result of fixed propagation delays in the on-time one-shot, as well as unavoidable delays in the external MOSFET switching. As VIN increases, these factors make the actual DH on-time slightly longer than the ideal on-time. The net effect is that frequency tends to falls slightly with increasing input voltage. The switching frequency also varies with load current as a result of the power losses in the MOSFETs and the inductor. For a conventional PWM constant-frequency converter, as load increases the duty cycle also increases slightly to compensate for IR and switching losses in the MOSFETs and inductor. A constant on-time converter must also compensate for the same losses by increasing the effective duty cycle (more time is spent drawing energy from VIN as losses increase). The on-time is essentially constant for a given VOUT/VIN combination, to off set the losses the off-time will tend to reduce slightly as load increases. The net effect is that switching frequency increases slightly with increasing load. FB pin R2 Figure 12 - Virtual ESR Ramp Current Dropout Performance The output voltage adjusts range for continuous-conduction operation is limited by the fixed 250 ns (typical) minimum off-time of the one-shot. When working with low input voltages, the duty-factor limit must be calculated using worst-case values for on and off times. The duty-factor limitation is shown by the next equation. DUTY = TON(MIN) TON(MIN) x TOFF(MAX) The inductor resistance and MOSFET on-state voltage drops must be included when performing worst-case dropout duty-factor calculations. System DC Accuracy (VOUT Controller) Three factors affect VOUT accuracy: the trip point of the FB error comparator, the ripple voltage variation with line and load, and the external resistor tolerance. The error comparator off set is trimmed so that under static conditions it trips when the feedback pin is 750 mV, 1 %. The on-time pulse from the SiC414 in the design example is calculated to give a pseudo-fixed frequency of 250 kHz. Some frequency variation with line and load is expected. This variation changes the output ripple voltage. Because constant on-time converters regulate to the valley of the output ripple, ½ of the output ripple appears as a DC regulation error. For example, if the output ripple is 50 mV with VIN = 6 V, then the measured DC output will be 25 mV above the comparator trip point. If the ripple increases to 80 mV with VIN = 25 V, then the measured DC output will be 40 mV above the comparator trip. The best way to minimize this effect is to minimize the output ripple. To compensate for valley regulation, it may be desirable to use passive droop. Take the feedback directly from the output side of the inductor and place a small amount of trace resistance between the inductor and output capacitor. Document Number: 65726 S10-1091-Rev. B, 03-May-10 www.vishay.com 17 SiC414 P5 3 4 2 1 3 4 2 2 4 3 1 1 28 25 4 7 VIN J4 Probe Test Pin 1 1 1 1 ENL EN B1 VIN C19 Open C15 10 µF C20 10 µF C10 10 µF R8 10K Open R12 1R R9 C24 C14 Open 0.1 µF C11 0.1 µF C13 0.01 µF 1 VOUT BST PGND PGND PGND PGND PGND PGND AGND AGND AGND 13 14 16 17 18 19 1 P3 V5V 3 26 29 1 C29 22 µF Optional 3 4 2 R15 10 K R39 0R R30 78.7 K PGOOD V5V 5 2 4 3 1 www.vishay.com 18 M4 1 1 Vishay Siliconix M1 M2 M3 1 1 1 1 1 1 V5V R1 Open P8 Step_I_Sense Si4812BDY Q1 1 1 1 5 J3 Probe Test Pin P9 SiC414 EVALUATION BOARD SCHEMATIC EN_PSV R2 Open 1 P6 P1 VIN ENL C5 0.1 µF 1K 5 R3 R4 1R01 R5 100 K C1 22 µF C2 22 µF C3 22 µF C4 22 µF VCTRL P2 VIN_GND R6 100 K R7 0R C6 0.1 µF L1 1 µH C21 10 µF C22 10 µF P12 LDTRG 1 J5 Probe Test Pin VO C7 0.1 µF 5 P10 VO P11 VO_GND B3 C18 C16 C23 C17 220 µF 220 µF 220 µF 220 µF 1 VO + C12 150 µF C8 10 µF C9 10 µF 1 2 V5V VLDO PGOOD RTON 27 22 FB 1 ILIM 23 6 5 8 9 10 11 30 VIN VIN VIN VIN VIN VIN U1 SiC414 LX LX LX LX LX LX 31 24 21 20 15 12 + + + + R10 10 K C25 100 pF 1 B2 VIN_GND Figure 13. Evaluation Board Schematic 0 C32 1 nF C30 180 pF P7 1 V5V R13 1 K C26 4.7 µF C27 Open C28 R11 1 µF B4 VO_GND C31 Open R23 16.5 K 1 J7 Probe Test Pin 5 J6 Probe Test Pin P4 VLDO Document Number: 65726 S10-1091-Rev. B, 03-May-10 SiC414 Vishay Siliconix BILL OF MATERIALS Qty. 1 4 4 1 1 5 3 1 1 1 2 1 1 2 3 1 1 1 1 1 4 1 4 1 12 4 Ref. Designator U1 C16, C18, C17, C23 C15, C20, C21, C22 L1 Q1 C1, C2, C3, C4, C29 C8, C9, C10 C26 C12 R4 R7, R11 R39 R3 R5, R6 R8, R10, R15 C6 R23 R13 C30 R30 C7, C11, C14, C28 C5 B1, B2, B3, B4 C13 P1, P2, P3, P4, P5, P6, P7, P8, P9, P10, P11, P12 M1, M2, M3, M4 Description SiC414 COT Buck Converter 220 µF, 10 V D 10 µF.16V.X7R.B, 1206 1.0 µH Si4812BDY-E3 CAP, 22 µF, 16 V, 1210 CAP10 µF 25 V 1210 4.7 µF 10 V 0805 CAP, Radial 150 µF 35 V 1 , 2512 Res 0  0R 50 V 0402 Res, 1k, 50 V, 0402 Res, 100k, 0603 Res.10k, 50 V, 0603 CAP CER 1.0 µF 35 V X7R 0805 Res 16.5k 1/10 W 1% 0603 SMD Res, 1K, 50 V, 0402 CAP, 180 pF, 0402 Res 78.7k 1/10 W 1 % 0603 SMD CAP, 0.1 µF 50 V 0603 CAP, 0.1 µF, 10 V, 0402 Solder Banana CAP, 0.01 µF, 50 V, 0402 Probe Hook Nylon on Stand off 0.01 µF 50 V SM0402 Terminal 22 µF 10 µF 4.7 µF 150 µF 1.0  0 0 1.0k 100k 10k 1.0 µF 16.5k 1.0k 180 pF 78.7k 0.1 µF 0.1 µF 16 V 25 V 10 V 35 V 200 V 50 V 50 V 50 V 50 V 50 V 35 V 50 V 50 V 50 V 50 V 50 V 10 V 220 µF 10 µF 1.0 µH 10 V 16 V Value Voltage Footprint MLPQ-28 4 x 4 mm SM593D SM1206 IHLP2525 SO-8 SM1210 SM1210 SM0805 Radial SM2512 SM0603 SM0402 SM0402 SM0603 SM0603 SM0805 SM0603 SM0402 SM0402 SM0603 SM0603 SM0402 Part Number SiC414 593D227X0010E2TE3 GRM31CR71C106KAC7L IHLP2525EZER1R0M01 Si4812BDY GRM32ER71C226ME18L TMK325B7106MM-T LMK212B7475KG-T EU-FM1V151 CRCW25121R00FKEG CRCW0603 0000ZOEA CRCW04020000ZOED CRCW04021K00FKED CRCW0603 100K FKEA CRCW060310KFKED GMK212B7105KG-T CRCW060316K5FKEA CRCW04021K00FKED VJ0402A181JXACW1BC CRCW060378K7FKEA VJ0603Y104KXACW1BC VJ0402Y104MXQCW1BC 575-6 VJ0402Y103KXACW1BC 0 8834 Manufacturer Vishay Vishay Murata Vishay Vishay Murata Taiyo Yuden Taiyo Yuden Panasonic Vishay Vishay Vishay Vishay Vishay Vishay Murata Vishay Vishay Vishay Vishay Vishay Vishay Keystone Vishay Keystone Keystone Document Number: 65726 S10-1091-Rev. B, 03-May-10 www.vishay.com 19 SiC414 Vishay Siliconix PCB LAYOUT OF THE EVALUATION BOARD Figure 14. Top Layer Figure 15. Mid Layer1 Figure 16. Mid Layer2 Figure 17. Bottom Layer www.vishay.com 20 Document Number: 65726 S10-1091-Rev. B, 03-May-10 SiC414 Vishay Siliconix PACKAGE DIMENSIONS AND MARKING INFO 56 PIN 1 Dot by Marking 2x A 0.10 C B 1 2 E 28L T/SLP (4.0 mm x 4.0 mm) E2-1 b 0.4000 e 3 E2-2 (Ne-1)X e Ref. K2 E2-3 B CA B D 2x 0.10 C A D2-3 K1 D2-1 PIN 1 Identification L D2-2 (Nd-1)X e Ref. Bottom View Top View Notes: 1. Use millimeters as the primary measurement. 2. Dimensioning and tolerances conform to ASME Y14.5M. - 1994. 3. N is the number of terminals. Nd is the number of terminals in X-direction and Ne is the number of terminals in Y-direction. A 4. Dimensions b applies to plated terminal and is measured between 0.15 mm and 0.30 mm from terminal tip. 0.08 C 5. The pin #1 identifier must be existed on the top surface of the package by using identification mark or other feature of package body. 0.000-0.0500 6. Exact shape and size of this feature is optional. 7. Package warpage max. 0.08 mm. 8. Applied only for terminals. 4 0.10 C 0.2030 Ref. Side View Dimensions A(8) A1 A2 b(4) D e E L N(3) Nd(3) Ne(3) D2-1 D2-2 D2-3 E2-1 E2-2 E2-3 K1 K2 Millimeters Min. 0.70 0.00 0.175 Nom. 0.75 0.20 Ref. 0.225 4.00 BSC 0.45 BSC 4.00 BSC 0.30 0.40 28 7 7 0.912 0.908 0.908 2.43 1.30 0.58 1.062 1.058 1.058 2.58 1.45 0.73 0.46 BSC 0.40 BSC 1.162 1.158 1.158 2.68 1.55 0.83 0.036 0.036 0.036 0.096 0.051 0.023 0.50 0.012 0.275 0.007 Max. 0.80 0.05 Min. 0.027 0.000 Inches Nom. 0.029 0.008 Ref. 0.009 0.157 BSC 0.018 BSC 0.157 BSC 0.016 28 7 7 0.042 0.042 0.042 0.102 0.057 0.029 0.018 BSC 0.016 BSC 0.046 0.046 0.046 0.105 0.061 0.033 0.020 0.011 Max. 0.031 0.002 Document Number: 65726 S10-1091-Rev. B, 03-May-10 www.vishay.com 21 SiC414 Vishay Siliconix RECOMMENDED LAND PATTERN 1.29 K X Dimensions C G H H1 H2 K H G Z Millimeters (3.95) 3.20 2.58 0.73 1.45 1.06 0.45 0.30 0.75 4.70 H2 (C) 1.29 P X Y Z H1 Y P K 2.58 Notes: a. Controlling dimensions are in millimeters (angles in degrees). b. This land pattern is for reference purposes only. Consult your manufacturing group to ensure your company’s manufacturing guidelines are met. c. Square package-dimensions apply in both X and Y directions. Vishay Siliconix maintains worldwide manufacturing capability. Products may be manufactured at one of several qualified locations. Reliability data for Silicon Technology and Package Reliability represent a composite of all qualified locations. For related documents such as package/tape drawings, part marking, and reliability data, see www.vishay.com/ppg?65726. www.vishay.com 22 Document Number: 65726 S10-1091-Rev. B, 03-May-10 PAD Pattern Vishay Siliconix PowerPAK® MLP44-28L Land Pattern Recommended Land Pattern 1.29 0.30 1.06 1 2 1.45 3 2.58 3.20 0.75 1.06 0.45 2.58 4.70 3.95 0.73 Recommended Land Pattern vs. Case Outline 0.30 0.400 0.06 0.06 0.06 0.75 1 2 3 Document Number: 70567 Revision: 17-May-10 1.29 www.vishay.com 1 Legal Disclaimer Notice Vishay Disclaimer ALL PRODUCT, PRODUCT SPECIFICATIONS AND DATA ARE SUBJECT TO CHANGE WITHOUT NOTICE TO IMPROVE RELIABILITY, FUNCTION OR DESIGN OR OTHERWISE. Vishay Intertechnology, Inc., its affiliates, agents, and employees, and all persons acting on its or their behalf (collectively, “Vishay”), disclaim any and all liability for any errors, inaccuracies or incompleteness contained in any datasheet or in any other disclosure relating to any product. Vishay makes no warranty, representation or guarantee regarding the suitability of the products for any particular purpose or the continuing production of any product. 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Customers using or selling Vishay products not expressly indicated for use in such applications do so at their own risk and agree to fully indemnify and hold Vishay and its distributors harmless from and against any and all claims, liabilities, expenses and damages arising or resulting in connection with such use or sale, including attorneys fees, even if such claim alleges that Vishay or its distributor was negligent regarding the design or manufacture of the part. Please contact authorized Vishay personnel to obtain written terms and conditions regarding products designed for such applications. No license, express or implied, by estoppel or otherwise, to any intellectual property rights is granted by this document or by any conduct of Vishay. Product names and markings noted herein may be trademarks of their respective owners. Document Number: 91000 Revision: 11-Mar-11 www.vishay.com 1