XD33035 DIP-24
XL33035 SOP-24
Brushless DC
Motor Controller
DIP/SOP
The 33035 is a high performance second generation monolithic
brushless DC motor controller containing all of the active functions
required to implement a full featured open loop, three or four phase
motor control system. This device consists of a rotor position decoder
for proper commutation sequencing, temperature compensated
reference capable of supplying sensor power, frequency
programmable sawtooth oscillator, three open collector top drivers,
and three high current totem pole bottom drivers ideally suited for
driving power MOSFETs.
Also included are protective features consisting of undervoltage
lockout, cycle−by−cycle current limiting with a selectable time
delayed latched shutdown mode, internal thermal shutdown, and a
unique fault output that can be interfaced into microprocessor
controlled systems.
Typical motor control functions include open loop speed, forward or
reverse direction, run enable, and dynamic braking. The 33035 is
designed to operate with electrical sensor phasings of 60°/300° or
120°/240°, and can also efficiently control brush DC motors.
Features
•
•
•
•
•
•
•
•
•
•
•
10 to 30 V Operation
Undervoltage Lockout
6.25 V Reference Capable of Supplying Sensor Power
Fully Accessible Error Amplifier for Closed Loop Servo
Applications
High Current Drivers Can Control External 3−Phase MOSFET
Bridge
Cycle−By−Cycle Current Limiting
Pinned−Out Current Sense Reference
Internal Thermal Shutdown
Selectable 60°/300° or 120°/240° Sensor Phasings
Can Efficiently Control Brush DC Motors with External MOSFET
H−Bridge
NCV Prefix for Automotive and Other Applications Requiring
Unique Site and Control Change Requirements; AEC−Q100
Qualified and PPAP Capable
1
Top Drive
Output
BT 1
24 CT
AT 2
23 Brake
Fwd/Rev
3
22 60°/120° Select
SA
4
21 AB
SB
5
20 BB
SC
6
19 CB
Output Enable
7
18 VC
Reference Output
8
17 VCC
Current Sense
Noninverting Input
9
16 Gnd
Sensor
Inputs
Oscillator 10
Error Amp
11
Noninverting Input
Error Amp
Inverting Input 12
15
Bottom
Drive
Outputs
Current Sense
Inverting Input
14 Fault Output
13
Error Amp Out/
PWM Input
XD33035 DIP-24 XL33035 SOP-24
Representative Schematic Diagram
VM
Fault
N
14
4
S
S
N
5
2
Rotor
Position
Decoder
6
Fwd/Rev
60°/120°
Enable
Vin
1
3
22
24
7
Undervoltage
17
Lockout
Motor
Output
Buffers
18
Reference
Regulator
8
Speed
Set
Faster
21
11
Error Amp
Thermal
Shutdown
12
20
PWM
RT
R
13
Q
S
CT
10
Oscillator
S
19
Q
9
R
15
16
23
Brake
This device contains 285 active transistors.
2
Current Sense
Reference
XD33035 DIP-24 XL33035 SOP-24
MAXIMUM RATINGS
Rating
Symbol
Value
Unit
VCC
40
V
−
Vref
V
IOSC
30
mA
Error Amp Input Voltage Range (Pins 11, 12, Note 1)
VIR
−0.3 to Vref
V
Error Amp Output Current (Source or Sink, Note 2)
IOut
10
mA
VSense
−0.3 to 5.0
V
Fault Output Voltage
VCE(Fault)
20
V
Fault Output Sink Current
ISink(Fault)
20
mA
Top Drive Voltage (Pins 1, 2, 24)
VCE(top)
40
V
Top Drive Sink Current (Pins 1, 2, 24)
ISink(top)
50
mA
Bottom Drive Supply Voltage (Pin 18)
VC
30
V
IDRV
100
mA
−
−
−
2000
200
2000
V
V
V
PD
RθJA
867
75
mW
°C/W
PD
RθJA
650
100
mW
°C/W
TJ
150
°C
TA
−40 to + 85
−40 to +125
°C
Tstg
−65 to +150
°C
Power Supply Voltage
Digital Inputs (Pins 3, 4, 5, 6, 22, 23)
Oscillator Input Current (Source or Sink)
Current Sense Input Voltage Range (Pins 9, 15)
Bottom Drive Output Current (Source or Sink, Pins 19, 20, 21)
Electrostatic Discharge Sensitivity (ESD)
Human Body Model (HBM) Class 2, JESD22 A114−C
Machine Model (MM) Class A, JESD22 A115−A
Charged Device Model (CDM), JESD22 C101−C
Power Dissipation and Thermal Characteristics
P Suffix, Dual In Line, Case 724
Maximum Power Dissipation @ TA = 85°C
Thermal Resistance, Junction−to−Air
DW Suffix, Surface Mount, Case 751E
Maximum Power Dissipation @ TA = 85°C
Thermal Resistance, Junction−to−Air
Operating Junction Temperature
Operating Ambient Temperature Range (Note 3)
33035
33035
Storage Temperature Range
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the
Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect
device reliability.
3
XD33035 DIP-24 XL33035 SOP-24
ELECTRICAL CHARACTERISTICS (VCC = VC = 20 V, RT = 4.7 k, CT = 10 nF, TA = 25°C, unless otherwise noted.)
Characteristic
Symbol
Min
Typ
Max
Unit
5.9
5.82
6.24
−
6.5
6.57
−
1.5
30
mV
REFERENCE SECTION
Reference Output Voltage (Iref = 1.0 mA)
TA = 25°C
(Note 4)
Vref
V
Line Regulation (VCC = 10 to 30 V, Iref = 1.0 mA)
Regline
Load Regulation (Iref = 1.0 to 20 mA)
Regload
−
16
30
mV
Output Short Circuit Current (Note 5)
ISC
40
75
−
mA
Reference Under Voltage Lockout Threshold
Vth
4.0
4.5
5.0
V
Input Offset Voltage (Note 4)
VIO
−
0.4
10
mV
Input Offset Current (Note 4)
IIO
−
8.0
500
nA
Input Bias Current (Note 4)
IIB
−
−46
−1000
nA
80
−
dB
ERROR AMPLIFIER
Input Common Mode Voltage Range
VICR
Open Loop Voltage Gain (VO = 3.0 V, RL = 15 k)
AVOL
70
Input Common Mode Rejection Ratio
CMRR
55
86
−
dB
Power Supply Rejection Ratio (VCC = VC = 10 to 30 V)
PSRR
65
105
−
dB
VOH
VOL
4.6
−
5.3
0.5
−
1.0
fOSC
22
25
28
kHz
Output Voltage Swing
High State (RL = 15 k to Gnd)
Low State (RL = 15 k to Vref)
(0 V to Vref)
V
V
OSCILLATOR SECTION
Oscillator Frequency
ΔfOSC/ΔV
−
0.01
5.0
%
Sawtooth Peak Voltage
VOSC(P)
−
4.1
4.5
V
Sawtooth Valley Voltage
VOSC(V)
1.2
1.5
−
V
Input Threshold Voltage (Pins 3, 4, 5, 6, 7, 22, 23)
High State
Low State
VIH
VIL
3.0
−
2.2
1.7
−
0.8
Sensor Inputs (Pins 4, 5, 6)
High State Input Current (VIH = 5.0 V)
Low State Input Current (VIL = 0 V)
IIH
IIL
−150
−600
−70
−337
−20
−150
Forward/Reverse, 60°/120° Select (Pins 3, 22, 23)
High State Input Current (VIH = 5.0 V)
Low State Input Current (VIL = 0 V)
IIH
IIL
−75
−300
−36
−175
−10
−75
Output Enable
High State Input Current (VIH = 5.0 V)
Low State Input Current (VIL = 0 V)
IIH
IIL
−60
−60
−29
−29
−10
−10
Vth
85
101
115
VICR
−
3.0
−
V
IIB
−
−0.9
−5.0
μA
Top Drive Output Sink Saturation (Isink = 25 mA)
VCE(sat)
−
0.5
1.5
V
Top Drive Output Off−State Leakage (VCE = 30 V)
IDRV(leak)
−
0.06
100
μA
tr
tf
−
−
107
26
300
300
VOH
VOL
(VCC −2.0)
−
(VCC −1.1)
1.5
−
2.0
Frequency Change with Voltage (VCC = 10 to 30 V)
LOGIC INPUTS
V
μA
μA
μA
CURRENT−LIMIT COMPARATOR
Threshold Voltage
Input Common Mode Voltage Range
Input Bias Current
mV
OUTPUTS AND POWER SECTIONS
Top Drive Output Switching Time (CL = 47 pF, RL = 1.0 k)
Rise Time
Fall Time
ns
Bottom Drive Output Voltage
High State (VCC = 20 V, VC = 30 V, Isource = 50 mA)
Low State (VCC = 20 V, VC = 30 V, Isink = 50 mA)
V
4
XD33035 DIP-24 XL33035 SOP-24
ELECTRICAL CHARACTERISTICS (VCC = VC = 20 V, RT = 4.7 k, CT = 10 nF, TA = 25°C, unless otherwise noted.)
Characteristic
Symbol
Min
Typ
Max
Unit
tr
tf
−
−
38
30
200
200
Fault Output Sink Saturation (Isink = 16 mA)
VCE(sat)
−
225
500
mV
Fault Output Off−State Leakage (VCE = 20 V)
IFLT(leak)
−
1.0
100
μA
Vth(on)
VH
8.2
0.1
8.9
0.2
10
0.3
ICC
−
−
−
−
12
14
3.5
5.0
16
20
6.0
10
OUTPUTS AND POWER SECTIONS
Bottom Drive Output Switching Time (CL = 1000 pF)
Rise Time
Fall Time
Under Voltage Lockout
Drive Output Enabled (VCC or VC Increasing)
Hysteresis
ns
V
Power Supply Current
Pin 17 (VCC = VC = 20 V)
Pin 17 (VCC = 20 V, VC = 30 V)
Pin 18 (VCC = VC = 20 V)
Pin 18 (VCC = 20 V, VC = 30 V)
1.
2.
3.
4.
mA
IC
The input common mode voltage or input signal voltage should not be allowed to go negative by more than 0.3 V.
The compliance voltage must not exceed the range of − 0.3 to Vref.
33035: T A = −40°C to +85°C;
Maximum package power dissipation limits must be observed.
5
XD33035 DIP-24 XL33035 SOP-24
4.0
,
OSC OSCILLATOR FREQUENCY CHANGE (%)
VCC = 20 V
TA = 25°C
CT = 1.0 nF
10
VCC = 20 V
VC = 20 V
RT = 4.7 k
CT = 10 nF
2.0
0
- 2.0
1.0
1.0
CT = 100 nF
CT = 10 nF
10
100
Δf
f OSC , OSCILLATOR FREQUENCY (kHz)
100
- 4.0
- 55
- 25
0
RT, TIMING RESISTOR (kΩ)
40
48
60
80
Phase
0
100
125
VCC = 20 V
VC = 20 V
TA = 25°C
Source Saturation
(Load to Ground)
-1.6
140
VCC = 20 V
VC = 20 V
VO = 3.0 V
RL = 15 k
CL = 100 pF
TA = 25°C
10 k
75
Vref
- 0.8
120
Gain
16
- 8.0
-16
- 24
1.0 k
0
100
24
8.0
φ, EXCESS PHASE (DEGREES)
Vsat , OUTPUT SATURATION VOLTAGE (V)
A VOL, OPEN LOOP VOLTAGE GAIN (dB)
56
32
50
Figure 2. Oscillator Frequency Change
versus Temperature
Figure 1. Oscillator Frequency versus
Timing Resistor
40
25
TA, AMBIENT TEMPERATURE (°C)
160
1.6
180
100 k
1.0 M
200
220
0.8
240
10 M
0
f, FREQUENCY (Hz)
Figure 3. Error Amp Open Loop Gain and
Phase versus Frequency
Gnd
0
1.0
Sink Saturation
(Load to Vref)
2.0
3.0
4.0
IO, OUTPUT LOAD CURRENT (mA)
Figure 4. Error Amp Output Saturation
Voltage versus Load Current
6
5.0
Vref , REFERENCE OUTPUT VOLTAGE (V)
- 4.0
- 8.0
- 12
- 16
VCC = 20 V
VC = 20 V
TA = 25°C
- 20
- 24
0
10
20
30
40
50
60
7.0
6.0
5.0
4.0
3.0
2.0
No Load
TA = 25°C
1.0
0
0
10
Iref, REFERENCE OUTPUT SOURCE CURRENT (mA)
20
30
40
VCC, SUPPLY VOLTAGE (V)
Figure 8. Reference Output Voltage
versus Supply Voltage
Figure 7. Reference Output Voltage Change
versus Output Source Current
100
VCC = 20 V
VC = 20 V
RT = 4.7 k
CT = 10 nF
TA = 25°C
OUTPUT DUTY CYCLE (%)
40
20
0
- 20
VCC = 20 V
VC = 20 V
No Load
- 40
80
60
40
20
0
- 55
- 25
0
25
50
75
100
125
0
2.0
3.0
4.0
PWM INPUT VOLTAGE (V)
Figure 9. Reference Output Voltage
versus Temperature
Figure 10. Output Duty Cycle versus
PWM Input Voltage
250
VCC = 20 V
VC = 20 V
RL = 1
CL = 1.0 nF
TA = 25°C
200
150
100
50
0
1.0
1.0
TA, AMBIENT TEMPERATURE (°C)
Vsat , OUTPUT SATURATION VOLTAGE (V)
t HL , BOTTOM DRIVE RESPONSE TIME (ns)
ΔVref , NORMALIZED REFERENCE VOLTAGE CHANGE (mV)
ΔVref , REFERENCE OUTPUT VOLTAGE CHANGE (mV)
XD33035 DIP-24 XL33035 SOP-24
0
2.0
3.0
4.0
5.0 6.0 7.0 8.0 9.0 10
CURRENT SENSE INPUT VOLTAGE (NORMALIZED TO Vth)
Figure 11. Bottom Drive Response Time versus
Current Sense Input Voltage
0.25
VCC = 20 V
VC = 20 V
TA = 25°C
0.2
0.15
0.1
0.05
0
0
4.0
8.0
12
ISink, SINK CURRENT (mA)
Figure 12. Fault Output Saturation
versus Sink Current
7
5.0
16
VCC = 20 V
VC = 20 V
TA = 25°C
100
OUTPUT VOLTAGE (%)
Vsat , OUTPUT SATURATION VOLTAGE (V)
XD33035 DIP-24 XL33035 SOP-24
1.2
0.8
0.4
0
0
0
10
20
30
ISink, SINK CURRENT (mA)
40
100 ns/DIV
Figure 14. Top Drive Output Waveform
VCC = 20 V
VC = 20 V
CL = 1.0 nF
TA = 25°C
100
0
50 ns/DIV
50 ns/DIV
Figure 15. Bottom Drive Output Waveform
Figure 16. Bottom Drive Output Waveform
0
16
VC
-1.0
I C , I CC, POWER SUPPLY CURRENT (mA)
Vsat, OUTPUT SATURATION VOLTAGE (V)
0
- 2.0
Source Saturation
(Load to Ground)
VCC = 20 V
VC = 20 V
TA = 25°C
2.0
1.0
0
0
VCC = 20 V
VC = 20 V
CL = 15 pF
TA = 25°C
OUTPUT VOLTAGE (%)
OUTPUT VOLTAGE (%)
Figure 13. Top Drive Output Saturation
Voltage versus Sink Current
100
VCC = 20 V
VC = 20 V
RL = 1.0 k
CL = 15 pF
TA = 25°C
Sink Saturation
(Load to VC)
Gnd
20
40
60
80
14
ICC
12
RT = 4.7 k
CT = 10 nF
Pins 3-6, 9, 15, 23 = Gnd
Pins 7, 22 = Open
TA = 25°C
10
8.0
6.0
4.0
IC
2.0
0
0
IO, OUTPUT LOAD CURRENT (mA)
5.0
10
15
20
25
VCC, SUPPLY VOLTAGE (V)
Figure 17. Bottom Drive Output Saturation
Voltage versus Load Current
Figure 18. Power and Bottom Drive Supply
Current versus Supply Voltage
8
30
XD33035 DIP-24 XL33035 SOP-24
PIN FUNCTION DESCRIPTION
Pin
Symbol
Description
1, 2, 24
BT, AT, CT
These three open collector Top Drive outputs are designed to drive the external
upper power switch transistors.
3
Fwd/Rev
The Forward/Reverse Input is used to change the direction of motor rotation.
4, 5, 6
SA, SB, SC
These three Sensor Inputs control the commutation sequence.
7
Output Enable
A logic high at this input causes the motor to run, while a low causes it to coast.
8
Reference Output
This output provides charging current for the oscillator timing capacitor CT and a
reference for the error amplifier. It may also serve to furnish sensor power.
9
Current Sense Noninverting Input
A 100 mV signal, with respect to Pin 15, at this input terminates output switch
conduction during a given oscillator cycle. This pin normally connects to the top
side of the current sense resistor.
10
Oscillator
The Oscillator frequency is programmed by the values selected for the timing
components, RT and CT.
11
Error Amp Noninverting Input
This input is normally connected to the speed set potentiometer.
12
Error Amp Inverting Input
This input is normally connected to the Error Amp Output in open loop
applications.
13
Error Amp Out/PWM Input
This pin is available for compensation in closed loop applications.
14
Fault Output
This open collector output is active low during one or more of the following
conditions: Invalid Sensor Input code, Enable Input at logic 0, Current Sense
Input greater than 100 mV (Pin 9 with respect to Pin 15), Undervoltage Lockout
activation, and Thermal Shutdown.
15
Current Sense Inverting Input
Reference pin for internal 100 mV threshold. This pin is normally connected to
the bottom side of the current sense resistor.
16
Gnd
This pin supplies a ground for the control circuit and should be referenced back
to the power source ground.
17
VCC
This pin is the positive supply of the control IC. The controller is functional over a
minimum VCC range of 10 to 30 V.
18
VC
The high state (VOH) of the Bottom Drive Outputs is set by the voltage applied to
this pin. The controller is operational over a minimum VC range of 10 to 30 V.
CB, BB, AB
These three totem pole Bottom Drive Outputs are designed for direct drive of the
external bottom power switch transistors.
22
60°/120° Select
The electrical state of this pin configures the control circuit operation for either
60° (high state) or 120° (low state) sensor electrical phasing inputs.
23
Brake
A logic low state at this input allows the motor to run, while a high state does not
allow motor operation and if operating causes rapid deceleration.
19, 20, 21
9
XD33035 DIP-24 XL33035 SOP-24
VM
4
SA
20 k
5
SB
Sensor
Inputs
CT
Lockout
18
VC
Reference
Regulator
9.1 V
Reference Output 8
Noninv. Input 11
PWM
13
Error Amp Out
PWM Input
10
21
AB
4.5 V
Error Amp
12
Faster
RT
24
Undervoltage
17
VCC
BT
25 μA
7
Output Enable
Vin
Top
Drive
Outputs
1
40 k
22
60°/120° Select
AT
Rotor
Position
Decoder
40 k
3
Forward/Reverse
Fault Output
2
20 k
6
SC
14
20 k
Oscillator
CT
Sink Only
= Positive True
Logic With
Hysteresis
20
Thermal
Shutdown
Latch
R
Q
S
Latch
S
Q
R
19
CB
40 k
9
100 mV
16
Bottom
Drive
Outputs
BB
Gnd
15
Current Sense Input
Current Sense
Reference Input
23
Brake Input
Figure 19. Representative Block Diagram
Inputs (Note 2)
Outputs (Note 3)
Sensor Electrical Phasing (Note 4)
Top Drives
Bottom Drives
SA
60°
SB
SA
120°
SB
SC
F/R
Enable
Brake
Current
Sense
AT
BT
CT
AB
BB
CB
Fault
1
1
1
0
0
0
0
1
1
1
0
0
0
0
1
1
1
0
1
1
0
0
0
1
0
1
1
1
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
0
1
0
0
1
1
1
1
1
1
0
0
1
0
0
1
1
0
0
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
1
1
(Note 5)
F/R = 1
1
1
1
0
0
0
0
1
1
1
0
0
0
0
1
1
1
0
1
1
0
0
0
1
0
1
1
1
0
0
0
0
0
1
1
1
0
0
0
0
0
0
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
1
1
0
0
1
1
1
1
1
1
0
0
0
0
1
1
1
1
1
0
0
0
0
1
0
1
1
0
0
0
0
0
0
1
1
0
1
1
1
1
1
1
(Note 5)
F/R = 0
1
0
0
1
1
0
1
0
1
0
1
0
X
X
X
X
0
0
X
X
1
1
1
1
1
1
0
0
0
0
0
0
0
0
(Note 6)
Brake = 0
1
0
0
1
1
0
1
0
1
0
1
0
X
X
X
X
1
1
X
X
1
1
1
1
1
1
1
1
1
1
1
1
0
0
(Note 7)
Brake = 1
V
V
V
V
V
V
X
1
1
X
1
1
1
1
1
1
1
(Note 8)
V
V
V
V
V
V
X
0
1
X
1
1
1
1
1
1
0
(Note 9)
V
V
V
V
V
V
X
0
0
X
1
1
1
0
0
0
0
(Note 10)
SC
10
XD33035 DIP-24 XL33035 SOP-24
V
V
V
V
V
V
X
1
0
1
1
1
1
0
0
0
0
(Note 11)
NOTES: 1. V = Any one of six valid sensor or drive combinations X = Don’t care.
2. The digital inputs (Pins 3, 4, 5, 6, 7, 22, 23) are all TTL compatible. The current sense input (Pin 9) has a 100 mV threshold with respect to Pin 15.
A logic 0 for this input is defined as < 85 mV, and a logic 1 is > 115 mV.
3. The fault and top drive outputs are open collector design and active in the low (0) state.
4. With 60°/120° select (Pin 22) in the high (1) state, configuration is for 60° sensor electrical phasing inputs. With Pin 22 in low (0) state, configuration
is for 120° sensor electrical phasing inputs.
5. Valid 60° or 120° sensor combinations for corresponding valid top and bottom drive outputs.
6. Invalid sensor inputs with brake = 0; All top and bottom drives off, Fault low.
7. Invalid sensor inputs with brake = 1; All top drives off, all bottom drives on, Fault low.
8. Valid 60° or 120° sensor inputs with brake = 1; All top drives off, all bottom drives on, Fault high.
9. Valid sensor inputs with brake = 1 and enable = 0; All top drives off, all bottom drives on, Fault low.
10. Valid sensor inputs with brake = 0 and enable = 0; All top and bottom drives off, Fault low.
11. All bottom drives off, Fault low.
Figure 20. Three Phase, Six Step Commutation Truth Table (Note 1)
Pulse Width Modulator
sensing an over current condition, immediately turning off
the switch and holding it off for the remaining duration of
oscillator ramp−up period. The stator current is converted to
a voltage by inserting a ground−referenced sense resistor RS
(Figure 36) in series with the three bottom switch transistors
(Q4, Q5, Q6). The voltage developed across the sense
resistor is monitored by the Current Sense Input (Pins 9 and
15), and compared to the internal 100 mV reference. The
current sense comparator inputs have an input common
mode range of approximately 3.0 V. If the 100 mV current
sense threshold is exceeded, the comparator resets the lower
sense latch and terminates output switch conduction. The
value for the current sense resistor is:
The use of pulse width modulation provides an energy
efficient method of controlling the motor speed by varying
the average voltage applied to each stator winding during the
commutation sequence. As CT discharges, the oscillator sets
both latches, allowing conduction of the top and bottom
drive outputs. The PWM comparator resets the upper latch,
terminating the bottom drive output conduction when the
positive−going ramp of CT becomes greater than the error
amplifier output. The pulse width modulator timing diagram
is shown in Figure 21. Pulse width modulation for speed
control appears only at the bottom drive outputs.
Current Limit
R +
S
I
Continuous operation of a motor that is severely
over−loaded results in overheating and eventual failure.
This destructive condition can best be prevented with the use
of cycle−by−cycle current limiting. That is, each on−cycle
is treated as a separate event. Cycle−by−cycle current
limiting is accomplished by monitoring the stator current
build−up each time an output switch conducts, and upon
0.1
stator(max)
The Fault output activates during an over current condition.
The dual−latch PWM configuration ensures that only one
single output conduction pulse occurs during any given
oscillator cycle, whether terminated by the output of the
error amp or the current limit comparator.
11
XD33035 DIP-24 XL33035 SOP-24
Undervoltage Lockout
Capacitor CT
A triple Undervoltage Lockout has been incorporated to
prevent damage to the IC and the external power switch
transistors. Under low power supply conditions, it
guarantees that the IC and sensors are fully functional, and
that there is sufficient bottom drive output voltage. The
positive power supplies to the IC (VCC) and the bottom
drives (VC) are each monitored by separate comparators that
have their thresholds at 9.1 V. This level ensures sufficient
gate drive necessary to attain low RDS(on) when driving
standard power MOSFET devices. When directly powering
the Hall sensors from the reference, improper sensor
operation can result if the reference output voltage falls
below 4.5 V. A third comparator is used to detect this
condition. If one or more of the comparators detects an
undervoltage condition, the Fault Output is activated, the top
drives are turned off and the bottom drive outputs are held
in a low state. Each of the comparators contain hysteresis to
prevent oscillations when crossing their respective
thresholds.
Error Amp
Out/PWM
Input
Current
Sense Input
Latch “Set"
Inputs
Top Drive
Outputs
Bottom Drive
Outputs
Fault Output
Figure 21. Pulse Width Modulator Timing Diagram
Reference
The on−chip 6.25 V regulator (Pin 8) provides charging
current for the oscillator timing capacitor, a reference for the
error amplifier, and can supply 20 mA of current suitable for
directly powering sensors in low voltage applications. In
higher voltage applications, it may become necessary to
transfer the power dissipated by the regulator off the IC. This
is easily accomplished with the addition of an external pass
transistor as shown in Figure 22. A 6.25 V reference level
was chosen to allow implementation of the simpler NPN
circuit, where Vref − VBE exceeds the minimum voltage
required by Hall Effect sensors over temperature. With
proper transistor selection and adequate heatsinking, up to
one amp of load current can be obtained.
The open collector Fault Output (Pin 14) was designed to
provide diagnostic information in the event of a system
malfunction. It has a sink current capability of 16 mA and
can directly drive a light emitting diode for visual indication.
Additionally, it is easily interfaced with TTL/CMOS logic
for use in a microprocessor controlled system. The Fault
Output is active low when one or more of the following
conditions occur:
1) Invalid Sensor Input code
2) Output Enable at logic [0]
3) Current Sense Input greater than 100 mV
4) Undervoltage Lockout, activation of one or more of
the comparators
5) Thermal Shutdown, maximum junction temperature
being exceeded
This unique output can also be used to distinguish between
motor start−up or sustained operation in an overloaded
condition. With the addition of an RC network between the
Fault Output and the enable input, it is possible to create a
time−delayed latched shutdown for overcurrent. The added
circuitry shown in Figure 23 makes easy starting of motor
systems which have high inertial loads by providing
additional starting torque, while still preserving overcurrent
protection. This task is accomplished by setting the current
limit to a higher than nominal value for a predetermined time.
During an excessively long overcurrent condition, capacitor
CDLY will charge, causing the enable input to cross its
threshold to a low state. A latch is then formed by the positive
feedback loop from the Fault Output to the Output Enable.
Once set, by the Current Sense Input, it can only be reset by
shorting CDLY or cycling the power supplies.
UVLO
17
Vin
Fault Output
18
REF
8
MPS
U01A
Vin
To
Sensor Control
Power Circuitry
≈5.6 V 6.25 V
39
17
UVLO
18
MPS
U51A
REF
0.1 8
To Control Circuitry
and Sensor Power
6.25 V
The NPN circuit is recommended for powering Hall or opto sensors, where
the output voltage temperature coefficient is not critical. The PNP circuit is
slightly more complex, but is also more accurate over temperature. Neither
circuit has current limiting.
Figure 22. Reference Output Buffers
12
XD33035 DIP-24 XL33035 SOP-24
Drive Outputs
of VCC. A zener clamp should be connected to this input
when driving power MOSFETs in systems where VCC is
greater than 20 V so as to prevent rupture of the MOSFET
gates.
The control circuitry ground (Pin 16) and current sense
inverting input (Pin 15) must return on separate paths to the
central input source ground.
The three top drive outputs (Pins 1, 2, 24) are open
collector NPN transistors capable of sinking 50 mA with a
minimum breakdown of 30 V. Interfacing into higher
voltage applications is easily accomplished with the circuits
shown in Figures 24 and 25.
The three totem pole bottom drive outputs (Pins 19, 20,
21) are particularly suited for direct drive of N−Channel
MOSFETs or NPN bipolar transistors (Figures 26, 27, 28
and 29). Each output is capable of sourcing and sinking up
to 100 mA. Power for the bottom drives is supplied from VC
(Pin 18). This separate supply input allows the designer
added flexibility in tailoring the drive voltage, independent
Thermal Shutdown
Internal thermal shutdown circuitry is provided to protect
the IC in the event the maximum junction temperature is
exceeded. When activated, typically at 170°C, the IC acts as
though the Output Enable was grounded.
14
14
4
2
5
2
6
RDLY
VM
POS
DEC
VCC
Rotor
Position
Decoder
1
1
Q2
Q1
Q3
3
24
24
22
UVLO
17
VM
Load
18
REF
21
Reset
21
8
20
CDLY
20
25 μA
7
Q4
19
t
DLY
[R
DLY
[R
DLY
C
DLY
ǒ
In
V
– (I enable R
)
ref
IL
DLY
V enable – (I enable R
)
th
IL
DLY
ǒ
6.25 – (20 x 10 –6 R
C
DLY
In
1.4 – (20 x 10 –6 R
)
DLY
)
DLY
Ǔ
Ǔ
Transistor Q1 is a common base stage used to level shift from VCC to the
high motor voltage, VM. The collector diode is required if VCC is present
while VM is low.
Figure 23. Timed Delayed Latched
Over Current Shutdown
Figure 24. High Voltage Interface with
NPN Power Transistors
13
XD33035 DIP-24 XL33035 SOP-24
14
VBoost VM = 170 V
VCC = 12 V
2
1.0 k
1
Rotor
Position
Decoder
1
24
2
5
6
4
1.0 M
4.7 k
21
1N4744
20
MOC8204
Optocoupler
Load
19
21
40 k
R
9
20
Q4
100 mV
23
C
15
RS
Brake Input
19
The addition of the RC filter will eliminate current−limit instability caused by the
leading edge spike on the current waveform. Resistor RS should be a low inductance type.
Figure 25. High Voltage Interface with
N−Channel Power MOSFETs
21
Figure 26. Current Waveform Spike Suppression
C
Rg
21
Rg
20
Rg
19
D
20
C
D
19
C
D
IB
9
100 mV
23
+
40 k
40 k
9
15
Brake Input
100 mV
D = 1N5819
23
Series gate resistor Rg will dampen any high frequency oscillations caused
by the MOSFET input capacitance and any series wiring induction in the
gate−source circuit. Diode D is required if the negative current into the Bottom Drive Outputs exceeds 50 mA.
15
0
-
Brake Input
t
Base Charge
Removal
The totem−pole output can furnish negative base current for enhanced transistor turn−off, with the addition of capacitor C.
Figure 27. MOSFET Drive Precautions
Figure 28. Bipolar Transistor Drive
14
XD33035 DIP-24 XL33035 SOP-24
D
21
SENSEFET
S
G
K
M
19
VCC = 12 V
Power Ground:
To Input Source Return
R @I @R
S pk
DS(on)
V 9 [
Pin
r
)R
DM(on)
S
If: SENSEFET = MPT10N10M
RS = 200 Ω, 1/4 W
Then : VPin 9 ≈ 0.75 Ipk
9
RS
15
100 mV
16
Gnd
Control Circuitry Ground (Pin 16) and Current Sense Inverting Input (Pin 15)
must return on separate paths to the Central Input Source Ground.
Virtually lossless current sensing can be achieved with the implementation of
SENSEFET power switches.
Figure 29. Current Sensing Power MOSFETs
4
8
7
6
R
Q
5
3
S
2
Boost Voltage (V)
20
VM + 12
VM + 8.0
VM + 4.0
0
20
40
60
Boost Current (mA)
1.0/200 V
*
1N5352A
1
0.001
VBoost
22
*
1555
* = MUR115
18 k
VM = 170 V
This circuit generates VBoost for Figure 25.
Figure 30. High Voltage Boost Supply
REF
8
REF
Enable
8
R1
25 μA
7
VA
VB
V
R1
R3
R4
Increase
Speed
11
R2
EA
EA
R2
12
13
PWM
PWM
13
ǒ
25 μA
11
C
12
7
Ǔ ǒ Ǔ
R3 ) R4
R2
Resistor R1 with capacitor C sets the acceleration time constant while R2
controls the deceleration. The values of R1 and R2 should be at least ten
times greater than the speed set potentiometer to minimize time constant
variations with different speed settings.
R4
+V
*
V
Pin 13
A R )R
R3 B
1
2 R3
Figure 32. Controlled Acceleration/Deceleration
Figure 31. Differential Input Speed Controller
15
XD33035 DIP-24 XL33035 SOP-24
13
BCD
Inputs
14
15
74LS145
12
5.0 V
16
11
VCC Q9
10
Q8
9
Q7
7
Q6
P3
6
Q5
P2
5
Q4
P1
4
Q3
P0
3
Q2
2
Q1
1
Gnd Q0
166 k
145 k
REF
8
100 k
126 k
25 μA
7
108 k
REF
8
11
92.3 k
77.6 k
To Sensor
Input (Pin 4)
EA
12
63.6 k
13
51.3 k
0.01
40.4 k
11
10 k
EA
100 k
0.1
8
The 74LS145 is an open collector BCD to One of Ten decoder. When connected as shown, input codes 0000 through 1001 steps the PWM in increments of approximately 10% from 0 to 90% on−time. Input codes 1010
through 1111 will produce 100% on−time or full motor speed.
ǒ
13
0.22 1.0 M
Figure 34. Closed Loop Speed Control
Ǔ ǒ Ǔ
R ) R4 R
R4
+V 3
2 *
V
Pin 3
ref R ) R
R
R3 B
3
1
2
REF
8
V +
B
ǒ
V
ref
R5
R6
) 1
Ǔ
R 3 §§ R 5 ø R 6
PWM
The rotor position sensors can be used as a tachometer. By differentiating
the positive−going edges and then integrating them over time, a voltage
proportional to speed can be generated. The error amp compares this voltage to that of the speed set to control the PWM.
Figure 33. Digital Speed Controller
V
12
1.0 M
10 k
25 μA
7
PWM
R1
T
R5
R2
R3
R6
R4
25 μA
7
11
EA
12
13
PWM
This circuit can control the speed of a cooling fan proportional to the difference
between the sensor and set temperatures. The control loop is closed as the
forced air cools the NTC thermistor. For controlled heating applications, exchange the positions of R1 and R2.
Figure 35. Closed Loop Temperature Control
16
XD33035 DIP-24 XL33035 SOP-24
SYSTEM APPLICATIONS
Three Phase Motor Commutation
spike reduction. Care must be taken in the selection of the
bottom power switch transistors so that the current during
braking does not exceed the device rating. During braking,
the peak current generated is limited only by the series
resistance of the conducting bottom switch and winding.
The three phase application shown in Figure 36 is a
full−featured open loop motor controller with full wave, six
step drive. The upper power switch transistors are
Darlingtons while the lower devices are power MOSFETs.
Each of these devices contains an internal parasitic catch
diode that is used to return the stator inductive energy back
to the power supply. The outputs are capable of driving a
delta or wye connected stator, and a grounded neutral wye
if split supplies are used. At any given rotor position, only
one top and one bottom power switch (of different totem
poles) is enabled. This configuration switches both ends of
the stator winding from supply to ground which causes the
current flow to be bidirectional or full wave. A leading edge
spike is usually present on the current waveform and can
cause a current−limit instability. The spike can be eliminated
by adding an RC filter in series with the Current Sense Input.
Using a low inductance type resistor for RS will also aid in
I
peak
+
V
R
) EMF
M
) R
switch
If the motor is running at maximum speed with no load, the
generated back EMF can be as high as the supply voltage,
and at the onset of braking, the peak current may approach
twice the motor stall current. Figure 37 shows the
commutation waveforms over two electrical cycles. The
first cycle (0° to 360°) depicts motor operation at full speed
while the second cycle (360° to 720°) shows a reduced speed
with about 50% pulse width modulation. The current
waveforms reflect a constant torque load and are shown
synchronous to the commutation frequency for clarity.
4
5
VM
Fault
Ind.
14
2
6
Q1
N
A
Rotor
Position
Decoder
3
winding
1
S
S
N
Q2
Fwd/Rev
60°/120°
Enable
24
22
7
Motor
Undervoltage
Lockout
18
Reference
Regulator
21
8
Speed
Set
Faster
RT
11
Error Amp
Q4
20
Thermal
Shutdown
12
13
C
25 μA
17
VM
B
Q3
Q5
PWM
R
19
Q
Q6
S
10
Oscillator
S
CT
ILimit
Q
15
Gnd
R
9
R
16
C
23
Brake
Figure 36. Three Phase, Six Step, Full Wave Motor Controller
17
RS
XD33035 DIP-24 XL33035 SOP-24
Rotor Electrical Position (Degrees)
0
60
120
180
240
300
360
420
480
540
600
660
720
SA
Sensor Inputs
60°/120°
Select Pin
Open
SB
SC
Code
100
110
111
011
001
000
100
110
111
011
001
000
100
110
010
011
001
101
100
110
010
011
001
101
Q1 + Q6
Q2 + Q6
SA
Sensor Inputs
60°/120°
Select Pin
Grounded
SB
SC
Code
AT
Top Drive
Outputs
BT
CT
AB
Bottom Drive
Outputs
BB
CB
Conducting
Power Switch
Transistors
Q2 + Q4 Q3 + Q4 Q3 + Q5
Q1 + Q5 Q1 + Q6
Q2 + Q6 Q2 + Q4 Q3 + Q4 Q3 + Q5 Q1 + Q5
+
A
O
−
+
Motor Drive
Current
B
O
−
+
C
O
−
Reduced Speed ( ≈ 50% PWM)
Full Speed (No PWM)
Fwd/Rev = 1
Figure 37. Three Phase, Six Step, Full Wave Commutation Waveforms
18
XD33035 DIP-24 XL33035 SOP-24
Figure 38 shows a three phase, three step, half wave motor
controller. This configuration is ideally suited for
automotive and other low voltage applications since there is
only one power switch voltage drop in series with a given
stator winding. Current flow is unidirectional or half wave
because only one end of each winding is switched.
Continuous braking with the typical half wave arrangement
presents a motor overheating problem since stator current is
limited only by the winding resistance. This is due to the lack
of upper power switch transistors, as in the full wave circuit,
used to disconnect the windings from the supply voltage
VM. A unique solution is to provide braking until the motor
stops and then turn off the bottom drives. This can be
accomplished by using the Fault Output in conjunction with
the Output Enable as an over current timer. Components
RDLY and CDLY are selected to give the motor sufficient time
to stop before latching the Output Enable and the top drive
AND gates low. When enabling the motor, the brake switch
is closed and the PNP transistor (along with resistors R1 and
RDLY) are used to reset the latch by discharging CDLY. The
stator flyback voltage is clamped by a single zener and three
diodes.
Motor
CDLY
R2
R1
14
RDLY
4
N
S
2
5
VM
Rotor
Position
Decoder
6
Fwd/Rev
60°/120°
22
7
24
25 μA
Undervoltage
17
VM
1
3
Lockout
18
Reference
Regulator
21
8
Speed
Set
Faster
RT
11
Error Amp
12
13
PWM
20
Thermal
Shutdown
R
19
Q
S
10
Oscillator
S
CT
Q
ILimit
R
Gnd
16
9
15
23
Brake
Figure 38. Three Phase, Three Step, Half Wave Motor Controller
19
S
N
XD33035 DIP-24 XL33035 SOP-24
Three Phase Closed Loop Controller
The 33035, by itself, is only capable of open loop
motor speed control. For closed loop motor speed control,
the 33035 requires an input voltage proportional to the
motor speed. Traditionally, this has been accomplished by
means of a tachometer to generate the motor speed feedback
voltage. Figure 39 shows an application whereby an
33039, powered from the 6.25 V reference (Pin 8) of the
33035, is used to generate the required feedback voltage
without the need of a costly tachometer. The same Hall
sensor signals used by the 33035 for rotor position
decoding are utilized by the 33039. Every positive or
negative going transition of the Hall sensor signals on any
of the sensor lines causes the 33039 to produce an output
pulse of defined amplitude and time duration, as determined
by the external resistor R1 and capacitor C1. The output train
1
8
2
3
of pulses at Pin 5 of the 33039 are integrated by the error
amplifier of the 33035 configured as an integrator to
produce a DC voltage level which is proportional to the
motor speed. This speed proportional voltage establishes the
PWM reference level at Pin 13 of the 33035 motor
controller and closes the feedback loop. The 33035
outputs drive a TMOS power MOSFET 3−phase bridge.
High currents can be expected during conditions of start−up,
breaking, and change of direction of the motor. The
system shown in Figure 39 is designed for a motorhaving
120/240 degrees Hall sensor electrical phasing. Thesystem
can easily be modified to accommodate 60/300degree
Hall sensor electrical phasing by removing thejumper
(J2) at Pin 22 of the 33035.
1.0 M
R1
7
33039
6
4
VM (18 to 30 V)
750 pF
C1
5
1.1 k
1.1 k
1.1 k
0.1
1000
TP1
1.0 k
F/R
1
1.0 k
24
2
23
Enable
5.1 k
Speed
0.01
Faster
N
3
22
4
21
5
20
J2
470
470
470
19
33035
7
18
8
17
9
16
10
15
11
14
12
13
Motor
1N5819
J1
330
1N5355B
18 V
Close Loop
TP2
Fault
100
0.05/1.0 W
0.1
100 k
2.2 k
0.1
1.0 M
10 k
S
S
Brake
N
6
4.7 k
1.0 k
1N4148
0.1
2.2 k
Reset
Latch On
Fault
33
47 μF
Figure 39. Closed Loop Brushless DC Motor Control
Using The 33035 and 33039
20
XD33035 DIP-24 XL33035 SOP-24
Sensor Phasing Comparison
There are four conventions used to establish the relative
phasing of the sensor signals in three phase motors. With six
step drive, an input signal change must occur every 60
electrical degrees; however, the relative signal phasing is
dependent upon the mechanical sensor placement. A
comparison of the conventions in electrical degrees is shown
in Figure 40. From the sensor phasing table in Figure 41,
note that the order of input codes for 60° phasing is the
reverse of 300°. This means the 33035, when configured
for 60° sensor electrical phasing, will operate a motor with
either 60° or 300° sensor electrical phasing, but resulting in
opposite directions of rotation. The same is true for the part
when it is configured for 120° sensor electrical phasing; the
motor will operate equally, but will result in opposite
directions of rotation for 120° for 240° conventions.
In this data sheet, the rotor position is always given in
electrical degrees since the mechanical position is a function
of the number of rotating magnetic poles. The relationship
between the electrical and mechanical position is:
Electrical Degrees + Mechanical Degrees
An increase in the number of magnetic poles causes more
electrical revolutions for a given mechanical revolution.
General purpose three phase motors typically contain a four
pole rotor which yields two electrical revolutions for one
mechanical.
Two and Four Phase Motor Commutation
The 33035 is also capable of providing a four step
output that can be used to drive two or four phase motors.
The truth table in Figure 42 shows that by connecting sensor
inputs SB and SC together, it is possible to truncate the
number of drive output states from six to four. The output
power switches are connected to BT, CT, BB, and CB.
Figure 43 shows a four phase, four step, full wave motor
control application. Power switch transistors Q1 through Q8
are Darlington type, each with an internal parasitic catch
diode. With four step drive, only two rotor position sensors
spaced at 90 electrical degrees are required. The
commutation waveforms are shown in Figure 44.
Figure 45 shows a four phase, four step, half wave motor
controller. It has the same features as the circuit in Figure 38,
except for the deletion of speed control and braking.
Rotor Electrical Position (Degrees)
0
60 120 180 240 300 360 420 480 540 600 660 720
SA
Sensor Electrical Phasing
60°
SB
SC
SA
120°
SB
SC
SA
240°
SB
33035 (60°/120° Select Pin Open)
SC
Inputs
SA
300°
ǒ#Rotor2 PolesǓ
Sensor Electrical
Spacing* = 90°
SA
SB
SB
SC
Figure 40. Sensor Phasing Comparison
Sensor Electrical Phasing (Degrees)
60°
120°
240°
300°
SA
SB
SC
SA
SB
SC
SA
SB
SC
SA
SB
SC
1
0
0
1
0
1
1
1
0
1
1
1
1
1
0
1
0
0
1
0
0
1
1
0
1
1
1
1
1
0
1
0
1
1
0
0
0
1
1
0
1
0
0
0
1
0
0
0
0
0
1
0
1
1
0
1
1
0
0
1
0
0
0
0
0
1
0
1
0
0
1
1
Outputs
Top Drives
Bottom Drives
F/R
BT
CT
BB
CB
1
1
0
0
0
1
1
0
1
1
1
1
1
0
1
1
1
1
0
1
0
0
0
1
1
0
0
0
1
1
0
0
0
1
1
0
0
0
0
0
1
1
1
0
0
1
1
1
0
1
0
0
0
0
1
0
*With 33035 sensor input SB connected to SC.
Figure 42. Two and Four Phase, Four Step,
Commutation Truth Table
Figure 41. Sensor Phasing Table
21
Figure 43. Four Phase, Four Step, Full Wave Motor Controller
22
CT
RT
VM
Enable
Fwd/Rev
10
13
12
11
8
18
17
7
22
3
6
5
4
Oscillator
Gnd
R
S
S
R
Thermal
Shutdown
Lockout
Undervoltage
PWM
Error Amp
Reference
Regulator
25 μA
16
Q
Q
Rotor
Position
Decoder
23
ILimit
9
15
19
20
21
24
1
2
14
C
Fault
Ind.
R
Q8
Q4
RS
Q7
Q3
VM
Q6
Q2
Q5
D
C
B
A
Q1
N
S
Motor
S
N
XD33035 DIP-24 XL33035 SOP-24
XD33035 DIP-24 XL33035 SOP-24
Rotor Electrical Position (Degrees)
0
90
180
270
360
450
540
630
720
SA
Sensor Inputs
60°/120°
Select Pin
Open
SB
Code
Top Drive
Outputs
10
10
01
00
10
11
01
00
Q3 + Q5
Q4 + Q6
Q1 + Q7
Q2 + Q8
Q 3 + Q5
Q 4 + Q6
Q1 + Q7
Q2 + Q 8
BT
CT
BB
Bottom Drive
Outputs
CB
Conducting
Power Switch
Transistors
+
A O
−
+
B O
Motor Drive
Current
+
C O
−
+
D O
−
Full Speed (No PWM)
Fwd/Rev = 1
Figure 44. Four Phase, Four Step, Full Wave Motor Controller
23
Figure 45. Four Phase, Four Step, Half Wave Motor Controller
24
CT
RT
VM
Enable
Fwd/Rev
10
13
12
11
8
18
17
7
22
3
6
5
4
Oscillator
Gnd
R
S
S
R
Thermal
Shutdown
Lockout
Undervoltage
PWM
Error Amp
Reference
Regulator
25 μ A
16
Q
Q
Rotor
Position
Decoder
Brake
23
ILimit
9
15
19
20
21
24
1
2
14
C
Fault
Ind.
R
RS
VM
Motor
S
N
N
S
XD33035 DIP-24 XL33035 SOP-24
XD33035 DIP-24 XL33035 SOP-24
Brush Motor Control
makes it possible to reverse the direction of the motor, using
the normal forward/reverse switch, on the fly and not have
to completely stop before reversing.
Though the 33035 was designed to control brushless
DC motors, it may also be used to control DC brush type
motors. Figure 46 shows an application of the 33035
driving a MOSFET H−bridge affording minimal parts count
to operate a brush−type motor. Key to the operation is the
input sensor code [100] which produces a top−left (Q1) and
a bottom−right (Q3) drive when the controller’s
forward/reverse pin is at logic [1]; top−right (Q4), bottom−left
(Q2) drive is realized when the Forward/Reverse pin is at
logic [0]. This code supports the requirements necessary for
H−bridge drive accomplishing both direction and speed
control.
The controller functions in a normal manner with a pulse
width modulated frequency of approximately 25 kHz.
Motor speed is controlled by adjusting the voltage presented
to the noninverting input of the error amplifier establishing
the PWM’s slice or reference level. Cycle−by−cycle current
limiting of the motor current is accomplished by sensing the
voltage (100 mV) across the RS resistor to ground of the
H−bridge motor current. The over current sense circuit
LAYOUT CONSIDERATIONS
Do not attempt to construct any of the brushless motor
control circuits on wire−wrap or plug−in prototype
boards. High frequency printed circuit layout techniques
are imperative to prevent pulse jitter. This is usually caused
by excessive noise pick−up imposed on the current sense or
error amp inputs. The printed circuit layout should contain
a ground plane with low current signal and high drive and
output buffer grounds returning on separate paths back to the
power supply input filter capacitor VM. Ceramic bypass
capacitors (0.1 μF) connected close to the integrated circuit
at VCC, VC, Vref and the error amp noninverting input may
be required depending upon circuit layout. This provides a
low impedance path for filtering any high frequency noise.
All high current loops should be kept as short as possible
using heavy copper runs to minimize radiated EMI.
25
XD33035 DIP-24 XL33035 SOP-24
Fault
Ind.
14
4
+12 V
20 k
2
5
1.0 k
Rotor
Position
Decoder
6
Fwd/Rev
1.0 k
1
Q1*
3
22
Enable
7
Q4*
Undervoltage
17
+12 V
24
25 μA
Lockout
18
DC Brush
Motor
Reference
Regulator
Q2*
21
8
M
22
11
10 k
Faster
10 k
Error Amp
12
13
PWM
20
Thermal
Shutdown
R
Q3*
19
Q
22
S
10
Oscillator
S
0.005
Q
ILimit
R
Gnd
16
23
Brake
Figure 46. H−Bridge Brush−Type Controller
26
9
15
1.0 k
0.001
RS
XD33035 DIP-24 XL33035 SOP-24
27
26
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