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XL33035

XL33035

  • 厂商:

    XINLUDA(信路达)

  • 封装:

    SOP-24

  • 描述:

    电机驱动器及控制器 SOP-24

  • 数据手册
  • 价格&库存
XL33035 数据手册
XD33035 DIP-24 XL33035 SOP-24 Brushless DC Motor Controller DIP/SOP The 33035 is a high performance second generation monolithic brushless DC motor controller containing all of the active functions required to implement a full featured open loop, three or four phase motor control system. This device consists of a rotor position decoder for proper commutation sequencing, temperature compensated reference capable of supplying sensor power, frequency programmable sawtooth oscillator, three open collector top drivers, and three high current totem pole bottom drivers ideally suited for driving power MOSFETs. Also included are protective features consisting of undervoltage lockout, cycle−by−cycle current limiting with a selectable time delayed latched shutdown mode, internal thermal shutdown, and a unique fault output that can be interfaced into microprocessor controlled systems. Typical motor control functions include open loop speed, forward or reverse direction, run enable, and dynamic braking. The 33035 is designed to operate with electrical sensor phasings of 60°/300° or 120°/240°, and can also efficiently control brush DC motors. Features • • • • • • • • • • • 10 to 30 V Operation Undervoltage Lockout 6.25 V Reference Capable of Supplying Sensor Power Fully Accessible Error Amplifier for Closed Loop Servo Applications High Current Drivers Can Control External 3−Phase MOSFET Bridge Cycle−By−Cycle Current Limiting Pinned−Out Current Sense Reference Internal Thermal Shutdown Selectable 60°/300° or 120°/240° Sensor Phasings Can Efficiently Control Brush DC Motors with External MOSFET H−Bridge NCV Prefix for Automotive and Other Applications Requiring Unique Site and Control Change Requirements; AEC−Q100 Qualified and PPAP Capable 1 Top Drive Output BT 1 24 CT AT 2 23 Brake Fwd/Rev 3 22 60°/120° Select SA 4 21 AB SB 5 20 BB SC 6 19 CB Output Enable 7 18 VC Reference Output 8 17 VCC Current Sense Noninverting Input 9 16 Gnd Sensor Inputs Oscillator 10 Error Amp 11 Noninverting Input Error Amp Inverting Input 12 15 Bottom Drive Outputs Current Sense Inverting Input 14 Fault Output 13 Error Amp Out/ PWM Input XD33035 DIP-24 XL33035 SOP-24 Representative Schematic Diagram VM Fault N 14 4 S S N 5 2 Rotor Position Decoder 6 Fwd/Rev 60°/120° Enable Vin 1 3 22 24 7 Undervoltage 17 Lockout Motor Output Buffers 18 Reference Regulator 8 Speed Set Faster 21 11 Error Amp Thermal Shutdown 12 20 PWM RT R 13 Q S CT 10 Oscillator S 19 Q 9 R 15 16 23 Brake This device contains 285 active transistors. 2 Current Sense Reference XD33035 DIP-24 XL33035 SOP-24 MAXIMUM RATINGS Rating Symbol Value Unit VCC 40 V − Vref V IOSC 30 mA Error Amp Input Voltage Range (Pins 11, 12, Note 1) VIR −0.3 to Vref V Error Amp Output Current (Source or Sink, Note 2) IOut 10 mA VSense −0.3 to 5.0 V Fault Output Voltage VCE(Fault) 20 V Fault Output Sink Current ISink(Fault) 20 mA Top Drive Voltage (Pins 1, 2, 24) VCE(top) 40 V Top Drive Sink Current (Pins 1, 2, 24) ISink(top) 50 mA Bottom Drive Supply Voltage (Pin 18) VC 30 V IDRV 100 mA − − − 2000 200 2000 V V V PD RθJA 867 75 mW °C/W PD RθJA 650 100 mW °C/W TJ 150 °C TA −40 to + 85 −40 to +125 °C Tstg −65 to +150 °C Power Supply Voltage Digital Inputs (Pins 3, 4, 5, 6, 22, 23) Oscillator Input Current (Source or Sink) Current Sense Input Voltage Range (Pins 9, 15) Bottom Drive Output Current (Source or Sink, Pins 19, 20, 21) Electrostatic Discharge Sensitivity (ESD) Human Body Model (HBM) Class 2, JESD22 A114−C Machine Model (MM) Class A, JESD22 A115−A Charged Device Model (CDM), JESD22 C101−C Power Dissipation and Thermal Characteristics P Suffix, Dual In Line, Case 724 Maximum Power Dissipation @ TA = 85°C Thermal Resistance, Junction−to−Air DW Suffix, Surface Mount, Case 751E Maximum Power Dissipation @ TA = 85°C Thermal Resistance, Junction−to−Air Operating Junction Temperature Operating Ambient Temperature Range (Note 3) 33035 33035 Storage Temperature Range Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect device reliability. 3 XD33035 DIP-24 XL33035 SOP-24 ELECTRICAL CHARACTERISTICS (VCC = VC = 20 V, RT = 4.7 k, CT = 10 nF, TA = 25°C, unless otherwise noted.) Characteristic Symbol Min Typ Max Unit 5.9 5.82 6.24 − 6.5 6.57 − 1.5 30 mV REFERENCE SECTION Reference Output Voltage (Iref = 1.0 mA) TA = 25°C (Note 4) Vref V Line Regulation (VCC = 10 to 30 V, Iref = 1.0 mA) Regline Load Regulation (Iref = 1.0 to 20 mA) Regload − 16 30 mV Output Short Circuit Current (Note 5) ISC 40 75 − mA Reference Under Voltage Lockout Threshold Vth 4.0 4.5 5.0 V Input Offset Voltage (Note 4) VIO − 0.4 10 mV Input Offset Current (Note 4) IIO − 8.0 500 nA Input Bias Current (Note 4) IIB − −46 −1000 nA 80 − dB ERROR AMPLIFIER Input Common Mode Voltage Range VICR Open Loop Voltage Gain (VO = 3.0 V, RL = 15 k) AVOL 70 Input Common Mode Rejection Ratio CMRR 55 86 − dB Power Supply Rejection Ratio (VCC = VC = 10 to 30 V) PSRR 65 105 − dB VOH VOL 4.6 − 5.3 0.5 − 1.0 fOSC 22 25 28 kHz Output Voltage Swing High State (RL = 15 k to Gnd) Low State (RL = 15 k to Vref) (0 V to Vref) V V OSCILLATOR SECTION Oscillator Frequency ΔfOSC/ΔV − 0.01 5.0 % Sawtooth Peak Voltage VOSC(P) − 4.1 4.5 V Sawtooth Valley Voltage VOSC(V) 1.2 1.5 − V Input Threshold Voltage (Pins 3, 4, 5, 6, 7, 22, 23) High State Low State VIH VIL 3.0 − 2.2 1.7 − 0.8 Sensor Inputs (Pins 4, 5, 6) High State Input Current (VIH = 5.0 V) Low State Input Current (VIL = 0 V) IIH IIL −150 −600 −70 −337 −20 −150 Forward/Reverse, 60°/120° Select (Pins 3, 22, 23) High State Input Current (VIH = 5.0 V) Low State Input Current (VIL = 0 V) IIH IIL −75 −300 −36 −175 −10 −75 Output Enable High State Input Current (VIH = 5.0 V) Low State Input Current (VIL = 0 V) IIH IIL −60 −60 −29 −29 −10 −10 Vth 85 101 115 VICR − 3.0 − V IIB − −0.9 −5.0 μA Top Drive Output Sink Saturation (Isink = 25 mA) VCE(sat) − 0.5 1.5 V Top Drive Output Off−State Leakage (VCE = 30 V) IDRV(leak) − 0.06 100 μA tr tf − − 107 26 300 300 VOH VOL (VCC −2.0) − (VCC −1.1) 1.5 − 2.0 Frequency Change with Voltage (VCC = 10 to 30 V) LOGIC INPUTS V μA μA μA CURRENT−LIMIT COMPARATOR Threshold Voltage Input Common Mode Voltage Range Input Bias Current mV OUTPUTS AND POWER SECTIONS Top Drive Output Switching Time (CL = 47 pF, RL = 1.0 k) Rise Time Fall Time ns Bottom Drive Output Voltage High State (VCC = 20 V, VC = 30 V, Isource = 50 mA) Low State (VCC = 20 V, VC = 30 V, Isink = 50 mA) V 4 XD33035 DIP-24 XL33035 SOP-24 ELECTRICAL CHARACTERISTICS (VCC = VC = 20 V, RT = 4.7 k, CT = 10 nF, TA = 25°C, unless otherwise noted.) Characteristic Symbol Min Typ Max Unit tr tf − − 38 30 200 200 Fault Output Sink Saturation (Isink = 16 mA) VCE(sat) − 225 500 mV Fault Output Off−State Leakage (VCE = 20 V) IFLT(leak) − 1.0 100 μA Vth(on) VH 8.2 0.1 8.9 0.2 10 0.3 ICC − − − − 12 14 3.5 5.0 16 20 6.0 10 OUTPUTS AND POWER SECTIONS Bottom Drive Output Switching Time (CL = 1000 pF) Rise Time Fall Time Under Voltage Lockout Drive Output Enabled (VCC or VC Increasing) Hysteresis ns V Power Supply Current Pin 17 (VCC = VC = 20 V) Pin 17 (VCC = 20 V, VC = 30 V) Pin 18 (VCC = VC = 20 V) Pin 18 (VCC = 20 V, VC = 30 V) 1. 2. 3. 4. mA IC The input common mode voltage or input signal voltage should not be allowed to go negative by more than 0.3 V. The compliance voltage must not exceed the range of − 0.3 to Vref. 33035: T A = −40°C to +85°C; Maximum package power dissipation limits must be observed. 5 XD33035 DIP-24 XL33035 SOP-24 4.0 , OSC OSCILLATOR FREQUENCY CHANGE (%) VCC = 20 V TA = 25°C CT = 1.0 nF 10 VCC = 20 V VC = 20 V RT = 4.7 k CT = 10 nF 2.0 0 - 2.0 1.0 1.0 CT = 100 nF CT = 10 nF 10 100 Δf f OSC , OSCILLATOR FREQUENCY (kHz) 100 - 4.0 - 55 - 25 0 RT, TIMING RESISTOR (kΩ) 40 48 60 80 Phase 0 100 125 VCC = 20 V VC = 20 V TA = 25°C Source Saturation (Load to Ground) -1.6 140 VCC = 20 V VC = 20 V VO = 3.0 V RL = 15 k CL = 100 pF TA = 25°C 10 k 75 Vref - 0.8 120 Gain 16 - 8.0 -16 - 24 1.0 k 0 100 24 8.0 φ, EXCESS PHASE (DEGREES) Vsat , OUTPUT SATURATION VOLTAGE (V) A VOL, OPEN LOOP VOLTAGE GAIN (dB) 56 32 50 Figure 2. Oscillator Frequency Change versus Temperature Figure 1. Oscillator Frequency versus Timing Resistor 40 25 TA, AMBIENT TEMPERATURE (°C) 160 1.6 180 100 k 1.0 M 200 220 0.8 240 10 M 0 f, FREQUENCY (Hz) Figure 3. Error Amp Open Loop Gain and Phase versus Frequency Gnd 0 1.0 Sink Saturation (Load to Vref) 2.0 3.0 4.0 IO, OUTPUT LOAD CURRENT (mA) Figure 4. Error Amp Output Saturation Voltage versus Load Current 6 5.0 Vref , REFERENCE OUTPUT VOLTAGE (V) - 4.0 - 8.0 - 12 - 16 VCC = 20 V VC = 20 V TA = 25°C - 20 - 24 0 10 20 30 40 50 60 7.0 6.0 5.0 4.0 3.0 2.0 No Load TA = 25°C 1.0 0 0 10 Iref, REFERENCE OUTPUT SOURCE CURRENT (mA) 20 30 40 VCC, SUPPLY VOLTAGE (V) Figure 8. Reference Output Voltage versus Supply Voltage Figure 7. Reference Output Voltage Change versus Output Source Current 100 VCC = 20 V VC = 20 V RT = 4.7 k CT = 10 nF TA = 25°C OUTPUT DUTY CYCLE (%) 40 20 0 - 20 VCC = 20 V VC = 20 V No Load - 40 80 60 40 20 0 - 55 - 25 0 25 50 75 100 125 0 2.0 3.0 4.0 PWM INPUT VOLTAGE (V) Figure 9. Reference Output Voltage versus Temperature Figure 10. Output Duty Cycle versus PWM Input Voltage 250 VCC = 20 V VC = 20 V RL = 1 CL = 1.0 nF TA = 25°C 200 150 100 50 0 1.0 1.0 TA, AMBIENT TEMPERATURE (°C) Vsat , OUTPUT SATURATION VOLTAGE (V) t HL , BOTTOM DRIVE RESPONSE TIME (ns) ΔVref , NORMALIZED REFERENCE VOLTAGE CHANGE (mV) ΔVref , REFERENCE OUTPUT VOLTAGE CHANGE (mV) XD33035 DIP-24 XL33035 SOP-24 0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10 CURRENT SENSE INPUT VOLTAGE (NORMALIZED TO Vth) Figure 11. Bottom Drive Response Time versus Current Sense Input Voltage 0.25 VCC = 20 V VC = 20 V TA = 25°C 0.2 0.15 0.1 0.05 0 0 4.0 8.0 12 ISink, SINK CURRENT (mA) Figure 12. Fault Output Saturation versus Sink Current 7 5.0 16 VCC = 20 V VC = 20 V TA = 25°C 100 OUTPUT VOLTAGE (%) Vsat , OUTPUT SATURATION VOLTAGE (V) XD33035 DIP-24 XL33035 SOP-24 1.2 0.8 0.4 0 0 0 10 20 30 ISink, SINK CURRENT (mA) 40 100 ns/DIV Figure 14. Top Drive Output Waveform VCC = 20 V VC = 20 V CL = 1.0 nF TA = 25°C 100 0 50 ns/DIV 50 ns/DIV Figure 15. Bottom Drive Output Waveform Figure 16. Bottom Drive Output Waveform 0 16 VC -1.0 I C , I CC, POWER SUPPLY CURRENT (mA) Vsat, OUTPUT SATURATION VOLTAGE (V) 0 - 2.0 Source Saturation (Load to Ground) VCC = 20 V VC = 20 V TA = 25°C 2.0 1.0 0 0 VCC = 20 V VC = 20 V CL = 15 pF TA = 25°C OUTPUT VOLTAGE (%) OUTPUT VOLTAGE (%) Figure 13. Top Drive Output Saturation Voltage versus Sink Current 100 VCC = 20 V VC = 20 V RL = 1.0 k CL = 15 pF TA = 25°C Sink Saturation (Load to VC) Gnd 20 40 60 80 14 ICC 12 RT = 4.7 k CT = 10 nF Pins 3-6, 9, 15, 23 = Gnd Pins 7, 22 = Open TA = 25°C 10 8.0 6.0 4.0 IC 2.0 0 0 IO, OUTPUT LOAD CURRENT (mA) 5.0 10 15 20 25 VCC, SUPPLY VOLTAGE (V) Figure 17. Bottom Drive Output Saturation Voltage versus Load Current Figure 18. Power and Bottom Drive Supply Current versus Supply Voltage 8 30 XD33035 DIP-24 XL33035 SOP-24 PIN FUNCTION DESCRIPTION Pin Symbol Description 1, 2, 24 BT, AT, CT These three open collector Top Drive outputs are designed to drive the external upper power switch transistors. 3 Fwd/Rev The Forward/Reverse Input is used to change the direction of motor rotation. 4, 5, 6 SA, SB, SC These three Sensor Inputs control the commutation sequence. 7 Output Enable A logic high at this input causes the motor to run, while a low causes it to coast. 8 Reference Output This output provides charging current for the oscillator timing capacitor CT and a reference for the error amplifier. It may also serve to furnish sensor power. 9 Current Sense Noninverting Input A 100 mV signal, with respect to Pin 15, at this input terminates output switch conduction during a given oscillator cycle. This pin normally connects to the top side of the current sense resistor. 10 Oscillator The Oscillator frequency is programmed by the values selected for the timing components, RT and CT. 11 Error Amp Noninverting Input This input is normally connected to the speed set potentiometer. 12 Error Amp Inverting Input This input is normally connected to the Error Amp Output in open loop applications. 13 Error Amp Out/PWM Input This pin is available for compensation in closed loop applications. 14 Fault Output This open collector output is active low during one or more of the following conditions: Invalid Sensor Input code, Enable Input at logic 0, Current Sense Input greater than 100 mV (Pin 9 with respect to Pin 15), Undervoltage Lockout activation, and Thermal Shutdown. 15 Current Sense Inverting Input Reference pin for internal 100 mV threshold. This pin is normally connected to the bottom side of the current sense resistor. 16 Gnd This pin supplies a ground for the control circuit and should be referenced back to the power source ground. 17 VCC This pin is the positive supply of the control IC. The controller is functional over a minimum VCC range of 10 to 30 V. 18 VC The high state (VOH) of the Bottom Drive Outputs is set by the voltage applied to this pin. The controller is operational over a minimum VC range of 10 to 30 V. CB, BB, AB These three totem pole Bottom Drive Outputs are designed for direct drive of the external bottom power switch transistors. 22 60°/120° Select The electrical state of this pin configures the control circuit operation for either 60° (high state) or 120° (low state) sensor electrical phasing inputs. 23 Brake A logic low state at this input allows the motor to run, while a high state does not allow motor operation and if operating causes rapid deceleration. 19, 20, 21 9 XD33035 DIP-24 XL33035 SOP-24 VM 4 SA 20 k 5 SB Sensor Inputs CT Lockout 18 VC Reference Regulator 9.1 V Reference Output 8 Noninv. Input 11 PWM 13 Error Amp Out PWM Input 10 21 AB 4.5 V Error Amp 12 Faster RT 24 Undervoltage 17 VCC BT 25 μA 7 Output Enable Vin Top Drive Outputs 1 40 k 22 60°/120° Select AT Rotor Position Decoder 40 k 3 Forward/Reverse Fault Output 2 20 k 6 SC 14 20 k Oscillator CT Sink Only = Positive True Logic With Hysteresis 20 Thermal Shutdown Latch R Q S Latch S Q R 19 CB 40 k 9 100 mV 16 Bottom Drive Outputs BB Gnd 15 Current Sense Input Current Sense Reference Input 23 Brake Input Figure 19. Representative Block Diagram Inputs (Note 2) Outputs (Note 3) Sensor Electrical Phasing (Note 4) Top Drives Bottom Drives SA 60° SB SA 120° SB SC F/R Enable Brake Current Sense AT BT CT AB BB CB Fault 1 1 1 0 0 0 0 1 1 1 0 0 0 0 1 1 1 0 1 1 0 0 0 1 0 1 1 1 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 0 1 0 0 1 1 1 1 1 1 0 0 1 0 0 1 1 0 0 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 1 1 (Note 5) F/R = 1 1 1 1 0 0 0 0 1 1 1 0 0 0 0 1 1 1 0 1 1 0 0 0 1 0 1 1 1 0 0 0 0 0 1 1 1 0 0 0 0 0 0 1 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 0 1 1 0 0 1 1 1 1 1 1 0 0 0 0 1 1 1 1 1 0 0 0 0 1 0 1 1 0 0 0 0 0 0 1 1 0 1 1 1 1 1 1 (Note 5) F/R = 0 1 0 0 1 1 0 1 0 1 0 1 0 X X X X 0 0 X X 1 1 1 1 1 1 0 0 0 0 0 0 0 0 (Note 6) Brake = 0 1 0 0 1 1 0 1 0 1 0 1 0 X X X X 1 1 X X 1 1 1 1 1 1 1 1 1 1 1 1 0 0 (Note 7) Brake = 1 V V V V V V X 1 1 X 1 1 1 1 1 1 1 (Note 8) V V V V V V X 0 1 X 1 1 1 1 1 1 0 (Note 9) V V V V V V X 0 0 X 1 1 1 0 0 0 0 (Note 10) SC 10 XD33035 DIP-24 XL33035 SOP-24 V V V V V V X 1 0 1 1 1 1 0 0 0 0 (Note 11) NOTES: 1. V = Any one of six valid sensor or drive combinations X = Don’t care. 2. The digital inputs (Pins 3, 4, 5, 6, 7, 22, 23) are all TTL compatible. The current sense input (Pin 9) has a 100 mV threshold with respect to Pin 15. A logic 0 for this input is defined as < 85 mV, and a logic 1 is > 115 mV. 3. The fault and top drive outputs are open collector design and active in the low (0) state. 4. With 60°/120° select (Pin 22) in the high (1) state, configuration is for 60° sensor electrical phasing inputs. With Pin 22 in low (0) state, configuration is for 120° sensor electrical phasing inputs. 5. Valid 60° or 120° sensor combinations for corresponding valid top and bottom drive outputs. 6. Invalid sensor inputs with brake = 0; All top and bottom drives off, Fault low. 7. Invalid sensor inputs with brake = 1; All top drives off, all bottom drives on, Fault low. 8. Valid 60° or 120° sensor inputs with brake = 1; All top drives off, all bottom drives on, Fault high. 9. Valid sensor inputs with brake = 1 and enable = 0; All top drives off, all bottom drives on, Fault low. 10. Valid sensor inputs with brake = 0 and enable = 0; All top and bottom drives off, Fault low. 11. All bottom drives off, Fault low. Figure 20. Three Phase, Six Step Commutation Truth Table (Note 1) Pulse Width Modulator sensing an over current condition, immediately turning off the switch and holding it off for the remaining duration of oscillator ramp−up period. The stator current is converted to a voltage by inserting a ground−referenced sense resistor RS (Figure 36) in series with the three bottom switch transistors (Q4, Q5, Q6). The voltage developed across the sense resistor is monitored by the Current Sense Input (Pins 9 and 15), and compared to the internal 100 mV reference. The current sense comparator inputs have an input common mode range of approximately 3.0 V. If the 100 mV current sense threshold is exceeded, the comparator resets the lower sense latch and terminates output switch conduction. The value for the current sense resistor is: The use of pulse width modulation provides an energy efficient method of controlling the motor speed by varying the average voltage applied to each stator winding during the commutation sequence. As CT discharges, the oscillator sets both latches, allowing conduction of the top and bottom drive outputs. The PWM comparator resets the upper latch, terminating the bottom drive output conduction when the positive−going ramp of CT becomes greater than the error amplifier output. The pulse width modulator timing diagram is shown in Figure 21. Pulse width modulation for speed control appears only at the bottom drive outputs. Current Limit R + S I Continuous operation of a motor that is severely over−loaded results in overheating and eventual failure. This destructive condition can best be prevented with the use of cycle−by−cycle current limiting. That is, each on−cycle is treated as a separate event. Cycle−by−cycle current limiting is accomplished by monitoring the stator current build−up each time an output switch conducts, and upon 0.1 stator(max) The Fault output activates during an over current condition. The dual−latch PWM configuration ensures that only one single output conduction pulse occurs during any given oscillator cycle, whether terminated by the output of the error amp or the current limit comparator. 11 XD33035 DIP-24 XL33035 SOP-24 Undervoltage Lockout Capacitor CT A triple Undervoltage Lockout has been incorporated to prevent damage to the IC and the external power switch transistors. Under low power supply conditions, it guarantees that the IC and sensors are fully functional, and that there is sufficient bottom drive output voltage. The positive power supplies to the IC (VCC) and the bottom drives (VC) are each monitored by separate comparators that have their thresholds at 9.1 V. This level ensures sufficient gate drive necessary to attain low RDS(on) when driving standard power MOSFET devices. When directly powering the Hall sensors from the reference, improper sensor operation can result if the reference output voltage falls below 4.5 V. A third comparator is used to detect this condition. If one or more of the comparators detects an undervoltage condition, the Fault Output is activated, the top drives are turned off and the bottom drive outputs are held in a low state. Each of the comparators contain hysteresis to prevent oscillations when crossing their respective thresholds. Error Amp Out/PWM Input Current Sense Input Latch “Set" Inputs Top Drive Outputs Bottom Drive Outputs Fault Output Figure 21. Pulse Width Modulator Timing Diagram Reference The on−chip 6.25 V regulator (Pin 8) provides charging current for the oscillator timing capacitor, a reference for the error amplifier, and can supply 20 mA of current suitable for directly powering sensors in low voltage applications. In higher voltage applications, it may become necessary to transfer the power dissipated by the regulator off the IC. This is easily accomplished with the addition of an external pass transistor as shown in Figure 22. A 6.25 V reference level was chosen to allow implementation of the simpler NPN circuit, where Vref − VBE exceeds the minimum voltage required by Hall Effect sensors over temperature. With proper transistor selection and adequate heatsinking, up to one amp of load current can be obtained. The open collector Fault Output (Pin 14) was designed to provide diagnostic information in the event of a system malfunction. It has a sink current capability of 16 mA and can directly drive a light emitting diode for visual indication. Additionally, it is easily interfaced with TTL/CMOS logic for use in a microprocessor controlled system. The Fault Output is active low when one or more of the following conditions occur: 1) Invalid Sensor Input code 2) Output Enable at logic [0] 3) Current Sense Input greater than 100 mV 4) Undervoltage Lockout, activation of one or more of the comparators 5) Thermal Shutdown, maximum junction temperature being exceeded This unique output can also be used to distinguish between motor start−up or sustained operation in an overloaded condition. With the addition of an RC network between the Fault Output and the enable input, it is possible to create a time−delayed latched shutdown for overcurrent. The added circuitry shown in Figure 23 makes easy starting of motor systems which have high inertial loads by providing additional starting torque, while still preserving overcurrent protection. This task is accomplished by setting the current limit to a higher than nominal value for a predetermined time. During an excessively long overcurrent condition, capacitor CDLY will charge, causing the enable input to cross its threshold to a low state. A latch is then formed by the positive feedback loop from the Fault Output to the Output Enable. Once set, by the Current Sense Input, it can only be reset by shorting CDLY or cycling the power supplies. UVLO 17 Vin Fault Output 18 REF 8 MPS U01A Vin To Sensor Control Power Circuitry ≈5.6 V 6.25 V 39 17 UVLO 18 MPS U51A REF 0.1 8 To Control Circuitry and Sensor Power 6.25 V The NPN circuit is recommended for powering Hall or opto sensors, where the output voltage temperature coefficient is not critical. The PNP circuit is slightly more complex, but is also more accurate over temperature. Neither circuit has current limiting. Figure 22. Reference Output Buffers 12 XD33035 DIP-24 XL33035 SOP-24 Drive Outputs of VCC. A zener clamp should be connected to this input when driving power MOSFETs in systems where VCC is greater than 20 V so as to prevent rupture of the MOSFET gates. The control circuitry ground (Pin 16) and current sense inverting input (Pin 15) must return on separate paths to the central input source ground. The three top drive outputs (Pins 1, 2, 24) are open collector NPN transistors capable of sinking 50 mA with a minimum breakdown of 30 V. Interfacing into higher voltage applications is easily accomplished with the circuits shown in Figures 24 and 25. The three totem pole bottom drive outputs (Pins 19, 20, 21) are particularly suited for direct drive of N−Channel MOSFETs or NPN bipolar transistors (Figures 26, 27, 28 and 29). Each output is capable of sourcing and sinking up to 100 mA. Power for the bottom drives is supplied from VC (Pin 18). This separate supply input allows the designer added flexibility in tailoring the drive voltage, independent Thermal Shutdown Internal thermal shutdown circuitry is provided to protect the IC in the event the maximum junction temperature is exceeded. When activated, typically at 170°C, the IC acts as though the Output Enable was grounded. 14 14 4 2 5 2 6 RDLY VM POS DEC VCC Rotor Position Decoder 1 1 Q2 Q1 Q3 3 24 24 22 UVLO 17 VM Load 18 REF 21 Reset 21 8 20 CDLY 20 25 μA 7 Q4 19 t DLY [R DLY [R DLY C DLY ǒ In V – (I enable R ) ref IL DLY V enable – (I enable R ) th IL DLY ǒ 6.25 – (20 x 10 –6 R C DLY In 1.4 – (20 x 10 –6 R ) DLY ) DLY Ǔ Ǔ Transistor Q1 is a common base stage used to level shift from VCC to the high motor voltage, VM. The collector diode is required if VCC is present while VM is low. Figure 23. Timed Delayed Latched Over Current Shutdown Figure 24. High Voltage Interface with NPN Power Transistors 13 XD33035 DIP-24 XL33035 SOP-24 14 VBoost VM = 170 V VCC = 12 V 2 1.0 k 1 Rotor Position Decoder 1 24 2 5 6 4 1.0 M 4.7 k 21 1N4744 20 MOC8204 Optocoupler Load 19 21 40 k R 9 20 Q4 100 mV 23 C 15 RS Brake Input 19 The addition of the RC filter will eliminate current−limit instability caused by the leading edge spike on the current waveform. Resistor RS should be a low inductance type. Figure 25. High Voltage Interface with N−Channel Power MOSFETs 21 Figure 26. Current Waveform Spike Suppression C Rg 21 Rg 20 Rg 19 D 20 C D 19 C D IB 9 100 mV 23 + 40 k 40 k 9 15 Brake Input 100 mV D = 1N5819 23 Series gate resistor Rg will dampen any high frequency oscillations caused by the MOSFET input capacitance and any series wiring induction in the gate−source circuit. Diode D is required if the negative current into the Bottom Drive Outputs exceeds 50 mA. 15 0 - Brake Input t Base Charge Removal The totem−pole output can furnish negative base current for enhanced transistor turn−off, with the addition of capacitor C. Figure 27. MOSFET Drive Precautions Figure 28. Bipolar Transistor Drive 14 XD33035 DIP-24 XL33035 SOP-24 D 21 SENSEFET S G K M 19 VCC = 12 V Power Ground: To Input Source Return R @I @R S pk DS(on) V 9 [ Pin r )R DM(on) S If: SENSEFET = MPT10N10M RS = 200 Ω, 1/4 W Then : VPin 9 ≈ 0.75 Ipk 9 RS 15 100 mV 16 Gnd Control Circuitry Ground (Pin 16) and Current Sense Inverting Input (Pin 15) must return on separate paths to the Central Input Source Ground. Virtually lossless current sensing can be achieved with the implementation of SENSEFET power switches. Figure 29. Current Sensing Power MOSFETs 4 8 7 6 R Q 5 3 S 2 Boost Voltage (V) 20 VM + 12 VM + 8.0 VM + 4.0 0 20 40 60 Boost Current (mA) 1.0/200 V * 1N5352A 1 0.001 VBoost 22 * 1555 * = MUR115 18 k VM = 170 V This circuit generates VBoost for Figure 25. Figure 30. High Voltage Boost Supply REF 8 REF Enable 8 R1 25 μA 7 VA VB V R1 R3 R4 Increase Speed 11 R2 EA EA R2 12 13 PWM PWM 13 ǒ 25 μA 11 C 12 7 Ǔ ǒ Ǔ R3 ) R4 R2 Resistor R1 with capacitor C sets the acceleration time constant while R2 controls the deceleration. The values of R1 and R2 should be at least ten times greater than the speed set potentiometer to minimize time constant variations with different speed settings. R4  +V   * V Pin 13 A R )R R3 B 1 2 R3 Figure 32. Controlled Acceleration/Deceleration Figure 31. Differential Input Speed Controller 15 XD33035 DIP-24 XL33035 SOP-24 13 BCD Inputs 14 15 74LS145 12 5.0 V 16 11 VCC Q9 10 Q8 9 Q7 7 Q6 P3 6 Q5 P2 5 Q4 P1 4 Q3 P0 3 Q2 2 Q1 1 Gnd Q0 166 k 145 k REF 8 100 k 126 k 25 μA 7 108 k REF 8 11 92.3 k 77.6 k To Sensor Input (Pin 4) EA 12 63.6 k 13 51.3 k 0.01 40.4 k 11 10 k EA 100 k 0.1 8 The 74LS145 is an open collector BCD to One of Ten decoder. When connected as shown, input codes 0000 through 1001 steps the PWM in increments of approximately 10% from 0 to 90% on−time. Input codes 1010 through 1111 will produce 100% on−time or full motor speed. ǒ 13 0.22 1.0 M Figure 34. Closed Loop Speed Control Ǔ ǒ Ǔ R ) R4 R R4  +V  3  2 * V Pin 3 ref R ) R R R3 B 3 1 2 REF 8 V + B ǒ V  ref R5 R6 ) 1 Ǔ R 3 §§ R 5 ø R 6 PWM The rotor position sensors can be used as a tachometer. By differentiating the positive−going edges and then integrating them over time, a voltage proportional to speed can be generated. The error amp compares this voltage to that of the speed set to control the PWM. Figure 33. Digital Speed Controller V 12 1.0 M 10 k 25 μA 7 PWM R1 T R5 R2 R3 R6 R4 25 μA 7 11 EA 12 13 PWM This circuit can control the speed of a cooling fan proportional to the difference between the sensor and set temperatures. The control loop is closed as the forced air cools the NTC thermistor. For controlled heating applications, exchange the positions of R1 and R2. Figure 35. Closed Loop Temperature Control 16 XD33035 DIP-24 XL33035 SOP-24 SYSTEM APPLICATIONS Three Phase Motor Commutation spike reduction. Care must be taken in the selection of the bottom power switch transistors so that the current during braking does not exceed the device rating. During braking, the peak current generated is limited only by the series resistance of the conducting bottom switch and winding. The three phase application shown in Figure 36 is a full−featured open loop motor controller with full wave, six step drive. The upper power switch transistors are Darlingtons while the lower devices are power MOSFETs. Each of these devices contains an internal parasitic catch diode that is used to return the stator inductive energy back to the power supply. The outputs are capable of driving a delta or wye connected stator, and a grounded neutral wye if split supplies are used. At any given rotor position, only one top and one bottom power switch (of different totem poles) is enabled. This configuration switches both ends of the stator winding from supply to ground which causes the current flow to be bidirectional or full wave. A leading edge spike is usually present on the current waveform and can cause a current−limit instability. The spike can be eliminated by adding an RC filter in series with the Current Sense Input. Using a low inductance type resistor for RS will also aid in I peak + V R ) EMF M ) R switch If the motor is running at maximum speed with no load, the generated back EMF can be as high as the supply voltage, and at the onset of braking, the peak current may approach twice the motor stall current. Figure 37 shows the commutation waveforms over two electrical cycles. The first cycle (0° to 360°) depicts motor operation at full speed while the second cycle (360° to 720°) shows a reduced speed with about 50% pulse width modulation. The current waveforms reflect a constant torque load and are shown synchronous to the commutation frequency for clarity. 4 5 VM Fault Ind. 14 2 6 Q1 N A Rotor Position Decoder 3 winding 1 S S N Q2 Fwd/Rev 60°/120° Enable 24 22 7 Motor Undervoltage Lockout 18 Reference Regulator 21 8 Speed Set Faster RT 11 Error Amp Q4 20 Thermal Shutdown 12 13 C 25 μA 17 VM B Q3 Q5 PWM R 19 Q Q6 S 10 Oscillator S CT ILimit Q 15 Gnd R 9 R 16 C 23 Brake Figure 36. Three Phase, Six Step, Full Wave Motor Controller 17 RS XD33035 DIP-24 XL33035 SOP-24 Rotor Electrical Position (Degrees) 0 60 120 180 240 300 360 420 480 540 600 660 720 SA Sensor Inputs 60°/120° Select Pin Open SB SC Code 100 110 111 011 001 000 100 110 111 011 001 000 100 110 010 011 001 101 100 110 010 011 001 101 Q1 + Q6 Q2 + Q6 SA Sensor Inputs 60°/120° Select Pin Grounded SB SC Code AT Top Drive Outputs BT CT AB Bottom Drive Outputs BB CB Conducting Power Switch Transistors Q2 + Q4 Q3 + Q4 Q3 + Q5 Q1 + Q5 Q1 + Q6 Q2 + Q6 Q2 + Q4 Q3 + Q4 Q3 + Q5 Q1 + Q5 + A O − + Motor Drive Current B O − + C O − Reduced Speed ( ≈ 50% PWM) Full Speed (No PWM) Fwd/Rev = 1 Figure 37. Three Phase, Six Step, Full Wave Commutation Waveforms 18 XD33035 DIP-24 XL33035 SOP-24 Figure 38 shows a three phase, three step, half wave motor controller. This configuration is ideally suited for automotive and other low voltage applications since there is only one power switch voltage drop in series with a given stator winding. Current flow is unidirectional or half wave because only one end of each winding is switched. Continuous braking with the typical half wave arrangement presents a motor overheating problem since stator current is limited only by the winding resistance. This is due to the lack of upper power switch transistors, as in the full wave circuit, used to disconnect the windings from the supply voltage VM. A unique solution is to provide braking until the motor stops and then turn off the bottom drives. This can be accomplished by using the Fault Output in conjunction with the Output Enable as an over current timer. Components RDLY and CDLY are selected to give the motor sufficient time to stop before latching the Output Enable and the top drive AND gates low. When enabling the motor, the brake switch is closed and the PNP transistor (along with resistors R1 and RDLY) are used to reset the latch by discharging CDLY. The stator flyback voltage is clamped by a single zener and three diodes. Motor CDLY R2 R1 14 RDLY 4 N S 2 5 VM Rotor Position Decoder 6 Fwd/Rev 60°/120° 22 7 24 25 μA Undervoltage 17 VM 1 3 Lockout 18 Reference Regulator 21 8 Speed Set Faster RT 11 Error Amp 12 13 PWM 20 Thermal Shutdown R 19 Q S 10 Oscillator S CT Q ILimit R Gnd 16 9 15 23 Brake Figure 38. Three Phase, Three Step, Half Wave Motor Controller 19 S N XD33035 DIP-24 XL33035 SOP-24 Three Phase Closed Loop Controller The 33035, by itself, is only capable of open loop motor speed control. For closed loop motor speed control, the 33035 requires an input voltage proportional to the motor speed. Traditionally, this has been accomplished by means of a tachometer to generate the motor speed feedback voltage. Figure 39 shows an application whereby an 33039, powered from the 6.25 V reference (Pin 8) of the 33035, is used to generate the required feedback voltage without the need of a costly tachometer. The same Hall sensor signals used by the 33035 for rotor position decoding are utilized by the 33039. Every positive or negative going transition of the Hall sensor signals on any of the sensor lines causes the 33039 to produce an output pulse of defined amplitude and time duration, as determined by the external resistor R1 and capacitor C1. The output train 1 8 2 3 of pulses at Pin 5 of the 33039 are integrated by the error amplifier of the 33035 configured as an integrator to produce a DC voltage level which is proportional to the motor speed. This speed proportional voltage establishes the PWM reference level at Pin 13 of the 33035 motor controller and closes the feedback loop. The 33035 outputs drive a TMOS power MOSFET 3−phase bridge. High currents can be expected during conditions of start−up, breaking, and change of direction of the motor. The system shown in Figure 39 is designed for a motorhaving 120/240 degrees Hall sensor electrical phasing. Thesystem can easily be modified to accommodate 60/300degree Hall sensor electrical phasing by removing thejumper (J2) at Pin 22 of the 33035. 1.0 M R1 7 33039 6 4 VM (18 to 30 V) 750 pF C1 5 1.1 k 1.1 k 1.1 k 0.1 1000 TP1 1.0 k F/R 1 1.0 k 24 2 23 Enable 5.1 k Speed 0.01 Faster N 3 22 4 21 5 20 J2 470 470 470 19 33035 7 18 8 17 9 16 10 15 11 14 12 13 Motor 1N5819 J1 330 1N5355B 18 V Close Loop TP2 Fault 100 0.05/1.0 W 0.1 100 k 2.2 k 0.1 1.0 M 10 k S S Brake N 6 4.7 k 1.0 k 1N4148 0.1 2.2 k Reset Latch On Fault 33 47 μF Figure 39. Closed Loop Brushless DC Motor Control Using The 33035 and 33039 20 XD33035 DIP-24 XL33035 SOP-24 Sensor Phasing Comparison There are four conventions used to establish the relative phasing of the sensor signals in three phase motors. With six step drive, an input signal change must occur every 60 electrical degrees; however, the relative signal phasing is dependent upon the mechanical sensor placement. A comparison of the conventions in electrical degrees is shown in Figure 40. From the sensor phasing table in Figure 41, note that the order of input codes for 60° phasing is the reverse of 300°. This means the 33035, when configured for 60° sensor electrical phasing, will operate a motor with either 60° or 300° sensor electrical phasing, but resulting in opposite directions of rotation. The same is true for the part when it is configured for 120° sensor electrical phasing; the motor will operate equally, but will result in opposite directions of rotation for 120° for 240° conventions. In this data sheet, the rotor position is always given in electrical degrees since the mechanical position is a function of the number of rotating magnetic poles. The relationship between the electrical and mechanical position is: Electrical Degrees + Mechanical Degrees An increase in the number of magnetic poles causes more electrical revolutions for a given mechanical revolution. General purpose three phase motors typically contain a four pole rotor which yields two electrical revolutions for one mechanical. Two and Four Phase Motor Commutation The 33035 is also capable of providing a four step output that can be used to drive two or four phase motors. The truth table in Figure 42 shows that by connecting sensor inputs SB and SC together, it is possible to truncate the number of drive output states from six to four. The output power switches are connected to BT, CT, BB, and CB. Figure 43 shows a four phase, four step, full wave motor control application. Power switch transistors Q1 through Q8 are Darlington type, each with an internal parasitic catch diode. With four step drive, only two rotor position sensors spaced at 90 electrical degrees are required. The commutation waveforms are shown in Figure 44. Figure 45 shows a four phase, four step, half wave motor controller. It has the same features as the circuit in Figure 38, except for the deletion of speed control and braking. Rotor Electrical Position (Degrees) 0 60 120 180 240 300 360 420 480 540 600 660 720 SA Sensor Electrical Phasing 60° SB SC SA 120° SB SC SA 240° SB 33035 (60°/120° Select Pin Open) SC Inputs SA 300° ǒ#Rotor2 PolesǓ Sensor Electrical Spacing* = 90° SA SB SB SC Figure 40. Sensor Phasing Comparison Sensor Electrical Phasing (Degrees) 60° 120° 240° 300° SA SB SC SA SB SC SA SB SC SA SB SC 1 0 0 1 0 1 1 1 0 1 1 1 1 1 0 1 0 0 1 0 0 1 1 0 1 1 1 1 1 0 1 0 1 1 0 0 0 1 1 0 1 0 0 0 1 0 0 0 0 0 1 0 1 1 0 1 1 0 0 1 0 0 0 0 0 1 0 1 0 0 1 1 Outputs Top Drives Bottom Drives F/R BT CT BB CB 1 1 0 0 0 1 1 0 1 1 1 1 1 0 1 1 1 1 0 1 0 0 0 1 1 0 0 0 1 1 0 0 0 1 1 0 0 0 0 0 1 1 1 0 0 1 1 1 0 1 0 0 0 0 1 0 *With 33035 sensor input SB connected to SC. Figure 42. Two and Four Phase, Four Step, Commutation Truth Table Figure 41. Sensor Phasing Table 21 Figure 43. Four Phase, Four Step, Full Wave Motor Controller 22 CT RT VM Enable Fwd/Rev 10 13 12 11 8 18 17 7 22 3 6 5 4 Oscillator Gnd R S S R Thermal Shutdown Lockout Undervoltage PWM Error Amp Reference Regulator 25 μA 16 Q Q Rotor Position Decoder 23 ILimit 9 15 19 20 21 24 1 2 14 C Fault Ind. R Q8 Q4 RS Q7 Q3 VM Q6 Q2 Q5 D C B A Q1 N S Motor S N XD33035 DIP-24 XL33035 SOP-24 XD33035 DIP-24 XL33035 SOP-24 Rotor Electrical Position (Degrees) 0 90 180 270 360 450 540 630 720 SA Sensor Inputs 60°/120° Select Pin Open SB Code Top Drive Outputs 10 10 01 00 10 11 01 00 Q3 + Q5 Q4 + Q6 Q1 + Q7 Q2 + Q8 Q 3 + Q5 Q 4 + Q6 Q1 + Q7 Q2 + Q 8 BT CT BB Bottom Drive Outputs CB Conducting Power Switch Transistors + A O − + B O Motor Drive Current + C O − + D O − Full Speed (No PWM) Fwd/Rev = 1 Figure 44. Four Phase, Four Step, Full Wave Motor Controller 23 Figure 45. Four Phase, Four Step, Half Wave Motor Controller 24 CT RT VM Enable Fwd/Rev 10 13 12 11 8 18 17 7 22 3 6 5 4 Oscillator Gnd R S S R Thermal Shutdown Lockout Undervoltage PWM Error Amp Reference Regulator 25 μ A 16 Q Q Rotor Position Decoder Brake 23 ILimit 9 15 19 20 21 24 1 2 14 C Fault Ind. R RS VM Motor S N N S XD33035 DIP-24 XL33035 SOP-24 XD33035 DIP-24 XL33035 SOP-24 Brush Motor Control makes it possible to reverse the direction of the motor, using the normal forward/reverse switch, on the fly and not have to completely stop before reversing. Though the 33035 was designed to control brushless DC motors, it may also be used to control DC brush type motors. Figure 46 shows an application of the 33035 driving a MOSFET H−bridge affording minimal parts count to operate a brush−type motor. Key to the operation is the input sensor code [100] which produces a top−left (Q1) and a bottom−right (Q3) drive when the controller’s forward/reverse pin is at logic [1]; top−right (Q4), bottom−left (Q2) drive is realized when the Forward/Reverse pin is at logic [0]. This code supports the requirements necessary for H−bridge drive accomplishing both direction and speed control. The controller functions in a normal manner with a pulse width modulated frequency of approximately 25 kHz. Motor speed is controlled by adjusting the voltage presented to the noninverting input of the error amplifier establishing the PWM’s slice or reference level. Cycle−by−cycle current limiting of the motor current is accomplished by sensing the voltage (100 mV) across the RS resistor to ground of the H−bridge motor current. The over current sense circuit LAYOUT CONSIDERATIONS Do not attempt to construct any of the brushless motor control circuits on wire−wrap or plug−in prototype boards. High frequency printed circuit layout techniques are imperative to prevent pulse jitter. This is usually caused by excessive noise pick−up imposed on the current sense or error amp inputs. The printed circuit layout should contain a ground plane with low current signal and high drive and output buffer grounds returning on separate paths back to the power supply input filter capacitor VM. Ceramic bypass capacitors (0.1 μF) connected close to the integrated circuit at VCC, VC, Vref and the error amp noninverting input may be required depending upon circuit layout. This provides a low impedance path for filtering any high frequency noise. All high current loops should be kept as short as possible using heavy copper runs to minimize radiated EMI. 25 XD33035 DIP-24 XL33035 SOP-24 Fault Ind. 14 4 +12 V 20 k 2 5 1.0 k Rotor Position Decoder 6 Fwd/Rev 1.0 k 1 Q1* 3 22 Enable 7 Q4* Undervoltage 17 +12 V 24 25 μA Lockout 18 DC Brush Motor Reference Regulator Q2* 21 8 M 22 11 10 k Faster 10 k Error Amp 12 13 PWM 20 Thermal Shutdown R Q3* 19 Q 22 S 10 Oscillator S 0.005 Q ILimit R Gnd 16 23 Brake Figure 46. H−Bridge Brush−Type Controller 26 9 15 1.0 k 0.001 RS XD33035 DIP-24 XL33035 SOP-24 27 26
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