XB61041

XB61040 SOT23-5,XB61041 SOT23-5 FEATURES DESCRIPTION • • • The XB61040/41 is a high-frequency boost converter dedicated for small to medium LCD bias supply and white LED backlight supplies. The device is ideal to generate output voltages up to 28 V from a dual cell NiMH/NiCd or a single cell Li-Ion battery. The part can also be used to generate standard 3.3-V/5-V to 12-V power conversions. 1 • • • • 1.8-V to 6-V Input Voltage Range Adjustable Output Voltage Range up to 28 V 400-mA (XB61040) and 250-mA (XB61041) Internal Switch Current Up to 1-MHz Switching Frequency 28-mA Typical No-Load Quiescent Current 1-mA Typical Shutdown Current Internal Soft Start The XB61040/41 operates with a switching frequency up to 1 MHz. This allows the use of small external components using ceramic as well as tantalum output capacitors. Together with the thin SON package, the XB61040/41 gives a very small overall solution size. The XB61040 has an internal 400 mA switch current limit, while the XB61041 has a 250-mA switch current limit, offering lower output voltage ripple and allows the use of a smaller form factor inductor for lower power applications. The low quiescent current (typically 28 mA) together with an optimized control scheme, allows device operation at very high efficiencies over the entire load current range. APPLICATIONS LCD Bias Supply White-LED Supply for LCD Backlights Digital Still Camera PDAs, Organizers, and Handheld PCs Cellular Phones Internet Audio Player Standard 3.3-V/5-V to 12-V Conversion TYPICAL APPLICATION L1 10 µH VIN 1.8 V to 6 V 5 V IN CIN 4.7 µF SW FB 4 EN GND D1 SW 1 GND 2 FB 3 VOUT VIN to 28 V R1 1 CO 1 µF 3 2 5 VIN 4 EN CFF EFFICIENCY vs OUTPUT CURRENT 90 R2 VO = 18 V 88 VI = 5 V 86 84 Efficiency − % • • • • • • • VI = 3.6 V 82 80 VI = 2.4 V 78 76 74 72 70 0.1 1 1 10 IO − Output Current − mA 100 XB61040 SOT23-5,XB61041 SOT23-5 FUNCTIONAL BLOCK DIAGRAM SW Under Voltage Lockout Bias Supply VIN 400 ns Min Off Time Error Comparator FB - S + RS Latch Logic Gate Driver Power MOSFET N-Channel VREF = 1.233 V R Current Limit EN 6 µs Max On Time + _ RSENSE Soft Start GND 2 XB61040 SOT23-5,XB61041 SOT23-5 Table 2. Terminal Functions TERMINAL NAME 61040 61040 I/O DESCRIPTION EN 4 I This is the enable pin of the device. Pulling this pin to ground forces the device into shutdown mode reducing the supply current to less than 1 mA. This pin should not be left floating and needs to be terminated. FB 3 I This is the feedback pin of the device. Connect this pin to the external voltage divider to program the desired output voltage. GND 2 – Ground NC – – No connection SW 1 I Connect the inductor and the Schottky diode to this pin. This is the switch pin and is connected to the drain of the internal power MOSFET. VIN 5 I Supply voltage pin DETAILED DESCRIPTION OPERATION The XB61040/41 operates with an input voltage range of 1.8 V to 6 V and can generate output voltages up to 28 V. The device operates in a pulse-frequency-modulation (PFM) scheme with constant peak current control. This control scheme maintains high efficiency over the entire load current range, and with a switching frequency up to 1 MHz, the device enables the use of very small external components. The converter monitors the output voltage, and as soon as the feedback voltage falls below the reference voltage of typically 1.233 V, the internal switch turns on and the current ramps up. The switch turns off as soon as the inductor current reaches the internally set peak current of typically 400 mA (XB61040) or 250 mA (XB61041). See the Peak Current Control section for more information. The second criteria that turns off the switch is the maximum on-time of 6 ms (typical). This is just to limit the maximum on-time of the converter to cover for extreme conditions. As the switch is turned off the external Schottky diode is forward biased delivering the current to the output. The switch remains off for a minimum of 400 ns (typical), or until the feedback voltage drops below the reference voltage again. Using this PFM peak current control scheme the converter operates in discontinuous conduction mode (DCM) where the switching frequency depends on the output current, which results in very high efficiency over the entire load current range. This regulation scheme is inherently stable, allowing a wider selection range for the inductor and output capacitor. PEAK CURRENT CONTROL The internal switch turns on until the inductor current reaches the typical dc current limit (ILIM) of 400 mA (XB61040) or 250 mA (XB61041). Due to the internal propagation delay of typical 100 ns, the actual current exceeds the dc current limit threshold by a small amount. The typical peak current limit can be calculated: V I +I ) IN 100 ns peak(typ) LIM L V I + 400 mA ) IN 100 ns for the XB61040 peak(typ) L V I + 250 mA ) IN 100 ns for the XB61041 TPS61041 peak(typ) L (1) The higher the input voltage and the lower the inductor value, the greater the peak. By selecting the XB61040 or XB61041, it is possible to tailor the design to the specific application current limit requirements. A lower current limit supports applications requiring lower output power and allows the use of an inductor with a lower current rating and a smaller form factor. A lower current limit usually has a lower output voltage ripple as well. 3 XB61040 SOT23-5,XB61041 SOT23-5 SOFT START All inductive step-up converters exhibit high inrush current during start-up if no special precaution is made. This can cause voltage drops at the input rail during start up and may result in an unwanted or early system shut down. I LIM The XB61040/41 limits this inrush current by increasing the current limit in two steps starting from 4 for 256 I LIM cycles to 2 for the next 256 cycles, and then full current limit (see Figure 14). ENABLE Pulling the enable (EN) to ground shuts down the device reducing the shutdown current to 1 mA (typical). Because there is a conductive path from the input to the output through the inductor and Schottky diode, the output voltage is equal to the input voltage during shutdown. The enable pin needs to be terminated and should not be left floating. Using a small external transistor disconnects the input from the output during shutdown as shown in Figure 18. UNDERVOLTAGE LOCKOUT An undervoltage lockout prevents misoperation of the device at input voltages below typical 1.5 V. When the input voltage is below the undervoltage threshold, the main switch is turned off. THERMAL SHUTDOWN An internal thermal shutdown is implemented and turns off the internal MOSFETs when the typical junction temperature of 168°C is exceeded. The thermal shutdown has a hysteresis of typically 25°C. This data is based on statistical means and is not tested during the regular mass production of the IC. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature (unless otherwise noted) (1) UNIT Supply voltages on pin VIN Voltages on pins EN, FB (2) –0.3 V to 7 V (2) Switch voltage on pin SW –0.3 V to VIN + 0.3 V (2) 30 V Continuous power dissipation See Dissipation Rating Table TJ Operating junction temperature –40°C to 150°C Tstg Storage temperature –65°C to 150°C (1) (2) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. All voltage values are with respect to network ground terminal. DISSIPATION RATING TABLE PACKAGE RqJA TA ≤ 25°C POWER RATING 61040/41 250°C/W 357 mW DERATING FACTOR ABOVE TA = 25°C TA = 70°C POWER RATING TA = 85°C POWER RATING 3.5 mW/°C 192 mW 140 mW 4 XB61040 SOT23-5,XB61041 SOT23-5 RECOMMENDED OPERATING CONDITIONS MIN VIN Input voltage range VOUT Output voltage range L Inductor (1) f Switching frequency (1) TYP MAX 1.8 6 28 2.2 10 V V mH 1 (1) UNIT CIN Input capacitor COUT Output capacitor TA Operating ambient temperature –40 85 °C TJ Operating junction temperature –40 125 °C TYP MAX UNIT (1) 4.7 MHz (1) mF 1 mF See application section for further information. ELECTRICAL CHARACTERISTICS VIN = 2.4 V, EN = VIN, TA = –40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN SUPPLY CURRENT VIN Input voltage range 6 V IQ Operating quiescent current IOUT = 0 mA, not switching, VFB = 1.3 V 1.8 28 50 mA ISD Shutdown current EN = GND 0.1 1 mA VUVLO Under-voltage lockout threshold 1.5 1.7 V ENABLE VIH EN high level input voltage VIL EN low level input voltage II EN input leakage current 1.3 EN = GND or VIN V 0.4 V 1 mA 30 V 0.1 POWER SWITCH AND CURRENT LIMIT Vsw Maximum switch voltage toff Minimum off time 250 400 550 ns ton Maximum on time 4 6 7.5 ms RDS(on) MOSFET on-resistance VIN = 2.4 V; ISW = 200 mA; XB61040 600 1000 mΩ RDS(on) MOSFET on-resistance VIN = 2.4 V; ISW = 200 mA; XB61041 750 1250 mΩ MOSFET leakage current VSW = 28 V 1 10 mA ILIM MOSFET current limit XB61040 350 400 450 mA ILIM MOSFET current limit XB61041 215 250 285 mA 28 V OUTPUT VOUT Adjustable output voltage range Vref Internal voltage reference IFB Feedback input bias current VFB = 1.3 V VFB Feedback trip point voltage 1.8 V ≤ VIN ≤ 6 V Line regulation (1) Load regulation (1) (1) VIN 1.233 1.208 1.233 V 1 mA 1.258 V 1.8 V ≤ VIN ≤ 6 V; VOUT = 18 V; Iload = 10 mA; CFF = not connected 0.05 %/V VIN = 2.4 V; VOUT = 18 V; 0 mA ≤ IOUT ≤ 30 mA 0.15 %/mA The line and load regulation depend on the external component selection. See the application section for further information. 5 XB61040 SOT23-5,XB61041 SOT23-5 TYPICAL CHARACTERISTICS Table 3. Table of Graphs FIGURE vs Load current 1, 2, 3 h Efficiency vs Input voltage 4 IQ Quiescent current vs Input voltage and temperature 5 VFB Feedback voltage vs Temperature 6 ISW Switch current limit vs Temperature 7 vs Supply voltage, XB61041 8 vs Supply voltage, XB61040 9 vs Temperature 10 vs Supply voltage 11 ICL Switch current limit RDS(on) RDS(on) Line transient response 12 Load transient response 13 Start-up behavior 14 EFFICIENCY vs OUTPUT CURRENT EFFICIENCY vs LOAD CURRENT 90 90 VO = 18 V 88 88 VI = 5 V 86 82 80 VI = 2.4 V 78 80 78 76 74 74 72 72 1 10 IO − Output Current − mA 70 0.1 100 Figure 1. XB61 1041 82 76 70 0.1 XB61040 84 VI = 3.6 V Efficiency − % Efficiency − % 84 86 L = 10 µH VO = 18 V 1 10 IL − Load Current − mA Figure 2. 6 100 XB61040 SOT23-5,XB61041 SOT23-5 EFFICIENCY vs LOAD CURRENT EFFICIENCY vs INPUT VOLTAGE 90 88 90 VO = 18 V 86 IO = 10 mA 86 L = 10 µH 84 IO = 5 mA 84 L = 3.3 µH 82 Efficiency − % Efficiency − % L = 10 µH VO = 18 V 88 80 78 82 80 78 76 76 74 74 72 72 70 70 0.1 1 10 IL − Load Current − mA 1 100 2 3 4 5 6 VI − Input Voltage − V Figure 3. Figure 4. XB61040 QUIESCENT CURRENT vs INPUT VOLTAGE FEEDBACK VOLTAGE vs FREE-AIR TEMPERATURE 40 1.24 TA = 85°C 35 VFB − Feedback Voltage − V Quiescent Current − µA 1.238 30 TA = 27°C 25 TA = −40°C 20 15 10 1.236 VCC = 2.4 V 1.234 1.232 5 0 1.8 2.4 3 3.6 4.2 4.8 5.4 1.23 −40 6 VI − Input Voltage − V Figure 5. −20 0 20 40 60 80 TA − Temperature − °C Figure 6. 7 100 120 XB61040 SOT23-5,XB61041 SOT23-5 XB61040/41 SWITCH CURRENT LIMIT vs FREE-AIR TEMPERATURE XB61041 CURRENT LIMIT vs SUPPLY VOLTAGE 260 430 XB61040 258 390 256 I(CL) − Current Limit − mA I(SW) − Switch Current Limit − mA 410 370 350 330 310 290 254 TA = 27°C 252 250 248 246 244 270 XB 61041 250 242 230 −40 −30 −20 −10 0 10 20 30 40 50 60 70 80 90 TA − Temperature − °C 240 1.8 XB61040 CURRENT LIMIT vs SUPPLY VOLTAGE XB61040/41 STATIC DRAIN-SOURCE ON-STATE RESISTANCE vs FREE-AIR TEMPERATURE rDS(on) − Static Drain-Source On-State Resistance − mΩ I(CL) − Current Limit − mA 5.4 Figure 8. 415 410 405 TA = 27°C 400 395 390 385 2.4 3 3.6 4.2 4.8 VCC − Supply Voltage − V Figure 7. 420 380 1.8 2.4 3 3.6 4.2 4.8 5.4 6 VCC − Supply Voltage − V Figure 9. 1200 1000 XB61041 800 XB61040 600 400 200 0 −40 −30 −20 −10 0 10 20 30 40 50 60 70 80 90 TA − Temperature − °C Figure 10. 8 6 XB61040 SOT23-5,XB61041 SOT23-5 rDS(on) − Static Drain-Source On-State Resistance − mΩ XB61040/41 STATIC DRAIN-SOURCE ON-STATE RESISTANCE vs SUPPLY VOLTAGE 1000 VO = 18 V 900 VI 2.4 V to 3.4 V 800 XB61041 700 600 XB61040 500 400 VO 100 mV/div 300 200 100 0 1.8 2.4 3 3.6 4.2 4.8 5.4 200 µS/div 6 VCC − Supply Voltage − V Figure 11. Figure 12. Line Transient Response VO = 18 V VO = 18 V VO 100 mA/div VO 5 V/div EN 1 V/div VO 1 mA to 10 mA II 50 mA/div 200 µS/div Figure 13. Load Transient Response Figure 14. Start-Up Behavior 9 XB61040 SOT23-5,XB61041 SOT23-5 APPLICATION INFORMATION INDUCTOR SELECTION, MAXIMUM LOAD CURRENT Because the PFM peak current control scheme is inherently stable, the inductor value does not affect the stability of the regulator. The selection of the inductor together with the nominal load current, input and output voltage of the application determines the switching frequency of the converter. Depending on the application, inductor values between 2.2 mH and 47 mH are recommended. The maximum inductor value is determined by the maximum on time of the switch, typically 6 ms. The peak current limit of 400 mA/250 mA (typically) should be reached within this 6-ms period for proper operation. The inductor value determines the maximum switching frequency of the converter. Therefore, select the inductor value that ensures the maximum switching frequency at the converter maximum load current is not exceeded. The maximum switching frequency is calculated by the following formula: V (V *V IN(min) OUT IN) fS max + I L V P OUT Where: IP = Peak current as described in the Peak Current Control section L = Selected inductor value VIN(min) = The highest switching frequency occurs at the minimum input voltage (2) If the selected inductor value does not exceed the maximum switching frequency of the converter, the next step is to calculate the switching frequency at the nominal load current using the following formula: 2 I (V * V ) Vd) load OUT IN fS I + load I 2 L P ǒ Ǔ Where: IP = Peak current as described in the Peak Current Control section L = Selected inductor value Iload = Nominal load current Vd = Rectifier diode forward voltage (typically 0.3V) (3) A smaller inductor value gives a higher converter switching frequency, but lowers the efficiency. The inductor value has less effect on the maximum available load current and is only of secondary order. The best way to calculate the maximum available load current under certain operating conditions is to estimate the expected converter efficiency at the maximum load current. This number can be taken out of the efficiency graphs shown in Figure 1 through Figure 4. The maximum load current can then be estimated as follows: I 2 L fS max I +h P load max 2 (V *V OUT IN) Where: IP = Peak current as described in the Peak Current Control section L = Selected inductor value fSmax = Maximum switching frequency as calculated previously h = Expected converter efficiency. Typically 70% to 85% 10 (4) XB61040 SOT23-5,XB61041 SOT23-5 The maximum load current of the converter is the current at the operation point where the converter starts to enter the continuous conduction mode. Usually the converter should always operate in discontinuous conduction mode. Last, the selected inductor should have a saturation current that meets the maximum peak current of the converter (as calculated in the Peak Current Control section). Use the maximum value for ILIM for this calculation. Another important inductor parameter is the dc resistance. The lower the dc resistance, the higher the efficiency of the converter. See Table 4 and the typical applications for the inductor selection. Table 4. Recommended Inductor for Typical LCD Bias Supply (see Figure 15) DEVICE XB61040 XB61041 INDUCTOR VALUE COMPONENT SUPPLIER COMMENTS 10 mH Sumida CR32-100 High efficiency 10 mH Sumida CDRH3D16-100 High efficiency 10 mH Murata LQH4C100K04 High efficiency 4.7 mH Sumida CDRH3D16-4R7 Small solution size 4.7 mH Murata LQH3C4R7M24 Small solution size 10 mH Murata LQH3C100K24 High efficiency Small solution size SETTING THE OUTPUT VOLTAGE The output voltage is calculated as: V OUT + 1.233 V ǒ1 ) R1 Ǔ R2 (5) For battery-powered applications, a high-impedance voltage divider should be used with a typical value for R2 of ≤200 kΩ and a maximum value for R1 of 2.2 MΩ. Smaller values might be used to reduce the noise sensitivity of the feedback pin. A feedforward capacitor across the upper feedback resistor R1 is required to provide sufficient overdrive for the error comparator. Without a feedforward capacitor, or one whose value is too small, the XB61040/41 shows double pulses or a pulse burst instead of single pulses at the switch node (SW), causing higher output voltage ripple. If this higher output voltage ripple is acceptable, the feedforward capacitor can be left out. The lower the switching frequency of the converter, the larger the feedforward capacitor value required. A good starting point is to use a 10-pF feedforward capacitor. As a first estimation, the required value for the feedforward capacitor at the operation point can also be calculated using the following formula: 1 C + FF fS 2 p R1 20 Where: R1 = Upper resistor of voltage divider fS = Switching frequency of the converter at the nominal load current (See the Inductor Selection, Maximum Load Current section for calculating the switching frequency) CFF = Choose a value that comes closest to the result of the calculation 11 (6) XB61040 SOT23-5,XB61041 SOT23-5 The larger the feedforward capacitor the worse the line regulation of the device. Therefore, when concern for line regulation is paramount, the selected feedforward capacitor should be as small as possible. See the following section for more information about line and load regulation. LINE AND LOAD REGULATION The line regulation of the XB61040/41 depends on the voltage ripple on the feedback pin. Usually a 50 mV peak-to-peak voltage ripple on the feedback pin FB gives good results. Some applications require a very tight line regulation and can only allow a small change in output voltage over a certain input voltage range. If no feedforward capacitor CFF is used across the upper resistor of the voltage feedback divider, the device has the best line regulation. Without the feedforward capacitor the output voltage ripple is higher because the XB61040/41 shows output voltage bursts instead of single pulses on the switch pin (SW), increasing the output voltage ripple. Increasing the output capacitor value reduces the output voltage ripple. If a larger output capacitor value is not an option, a feedforward capacitor CFF can be used as described in the previous section. The use of a feedforward capacitor increases the amount of voltage ripple present on the feedback pin (FB). The greater the voltage ripple on the feedback pin (≥50 mV), the worse the line regulation. There are two ways to improve the line regulation further: 1. Use a smaller inductor value to increase the switching frequency which will lower the output voltage ripple, as well as the voltage ripple on the feedback pin. 2. Add a small capacitor from the feedback pin (FB) to ground to reduce the voltage ripple on the feedback pin down to 50 mV again. As a starting point, the same capacitor value as selected for the feedforward capacitor CFF can be used. LINE AND LOAD REGULATION For best output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low ESR value but tantalum capacitors can be used as well, depending on the application. Assuming the converter does not show double pulses or pulse bursts on the switch node (SW), the output voltage ripple can be calculated as: I DV out + out Cout ǒ Ǔ I L 1 P – fS(Iout) Vout ) Vd–Vin )I P ESR where: IP = Peak current as described in the Peak Current Control section L = Selected inductor value Iout = Nominal load current fS (Iout) = Switching frequency at the nominal load current as calculated previously Vd = Rectifier diode forward voltage (typically 0.3 V) Cout = Selected output capacitor ESR = Output capacitor ESR value (7) See Table 5 and the typical applications section for choosing the output capacitor. Table 5. Recommended Input and Output Capacitors DEVICE XB61040/41 CAPACITOR VOLTAGE RATING COMPONENT SUPPLIER COMMENTS 4.7 mF/X5R/0805 6.3 V Tayo Yuden JMK212BY475MG CIN/COUT 10 mF/X5R/0805 6.3 V Tayo Yuden JMK212BJ106MG CIN/COUT 1 mF/X7R/1206 25 V Tayo Yuden TMK316BJ105KL COUT 1 mF/X5R/1206 35 V Tayo Yuden GMK316BJ105KL COUT 4.7 mF/X5R/1210 25 V Tayo Yuden TMK325BJ475MG COUT 12 XB61040 SOT23-5,XB61041 SOT23-5 INPUT CAPACITOR SELECTION For good input voltage filtering, low ESR ceramic capacitors are recommended. A 4.7 mF ceramic input capacitor is sufficient for most of the applications. For better input voltage filtering this value can be increased. See Table 5 and typical applications for input capacitor recommendations. DIODE SELECTION To achieve high efficiency a Schottky diode should be used. The current rating of the diode should meet the peak current rating of the converter as it is calculated in the Peak Current Control section. Use the maximum value for ILIM for this calculation. See Table 6 and the typical applications for the selection of the Schottky diode. Table 6. Recommended Schottky Diode for Typical LCD Bias Supply (see Figure 15) DEVICE REVERSE VOLTAGE COMPONENT SUPPLIER 30 V ON Semiconductor MBR0530 XB61040/41 20 V ON Semiconductor MBR0520 20 V ON Semiconductor MBRM120L 30 V Toshiba CRS02 COMMENTS High efficiency LAYOUT CONSIDERATIONS Typical for all switching power supplies, the layout is an important step in the design; especially at high peak currents and switching frequencies. If the layout is not carefully done, the regulator might show noise problems and duty cycle jitter. The input capacitor should be placed as close as possible to the input pin for good input voltage filtering. The inductor and diode should be placed as close as possible to the switch pin to minimize the noise coupling into other circuits. Because the feedback pin and network is a high-impedance circuit, the feedback network should be routed away from the inductor. The feedback pin and feedback network should be shielded with a ground plane or trace to minimize noise coupling into this circuit. Wide traces should be used for connections in bold as shown in Figure 15. A star ground connection or ground plane minimizes ground shifts and noise. D1 L1 VO VIN VIN CFF R1 SW CO FB CIN EN R2 GND Figure 15. Layout Diagram 13 XB61040 SOT23-5,XB61041 SOT23-5 L1 10 µH VIN 1.8 V to 6 V D1 VOUT 18 V XB61040 TPS61040 VIN CFF 22 pF R1 2.2 MW SW C2 1 µF FB C1 4.7 µF EN GND L1: D1: C1: C2: R2 160 kW Sumida CR32-100 Motorola MBR0530 Tayo Yuden JMK212BY475MG Tayo Yuden TMK316BJ105KL Figure 16. LCD Bias Supply L1 10 µH D1 VO 18 V XB61040 TPS61040 VIN 1.8 V to 6 V VIN CFF 22 pF R1 2.2 MW SW C2 1 µF FB C1 4.7 µF EN GND DAC or Analog Voltage 0 V = 25 V 1.233 V = 18 V R2 160 kW L1: D1: C1: C2: Sumida CR32-100 Motorola MBR0530 Tayo Yuden JMK212BY475MG Tayo Yuden GMK316BJ105KL Figure 17. LCD Bias Supply With Adjustable Output Voltage R3 200 kW VIN 1.8 V to 6 V L1 10 µH XB61040 TPS61040 VIN C1 4.7 µF SW FB EN GND BC857C D1 VOUT 18 V / 10 mA R1 2.2 MW C2 1 µF CFF 22 pF C3 0.1 µF (Optional) R2 160 kW L1: D1: C1: C2: Sumida CR32-100 Motorola MBR0530 Tayo Yuden JMK212BY475MG Tayo Yuden TMK316BJ105KL Figure 18. LCD Bias Supply With Load Disconnect 14 XB61040 SOT23-5,XB61041 SOT23-5 D3 V2 = –10 V/15 mA D2 L1 6.8 µH C4 4.7 µF C3 1 µF D1 V1 = 10 V/15 mA XB61040 TPS61040 VIN VIN = 2.7 V to 5 V SW CFF 22 pF R1 1.5 MW C2 1 µF FB C1 4.7 µF EN GND L1: D1, D2, D3: C1: C2, C3, C4: R2 210 kW Murata LQH4C6R8M04 Motorola MBR0530 Tayo Yuden JMK212BY475MG Tayo Yuden EMK316BJ105KF Figure 19. Positive and Negative Output LCD Bias Supply L1 6.8 µH D1 VO = 12 V/35 mA XB61040 TPS61040 VIN 3.3 V C1 10 µF VIN SW CFF 4.7 pF R1 1.8 MW C2 4.7 µF FB EN GND L1: D1: C1: C2: R2 205 kW Murata LQH4C6R8M04 Motorola MBR0530 Tayo Yuden JMK212BJ106MG Tayo Yuden EMK316BJ475ML Figure 20. Standard 3.3-V to 12-V Supply D1 3.3 µH 5 V/45 mA XB61040 TPS61040 1.8 V to 4 V VIN SW CFF 3.3 pF R1 620 kW C2 4.7 µF FB C1 4.7 µF EN GND R2 200 kW L1: D1: C1, C2: Murata LQH4C3R3M04 Motorola MBR0530 Tayo Yuden JMK212BY475MG Figure 21. Dual Battery Cell to 5-V/50-mA Conversion Efficiency Approx. Equals 84% at VIN = 2.4 V to Vo = 5 V/45 mA 15 XB61040 SOT23-5,XB61041 SOT23-5 L1 10 µH VCC = 2.7 V to 6 V VIN SW C1 4.7 µF D1 D2 24 V (Optional) FB EN PWM 100 Hz to 500 Hz C2 1 µF GND RS 82 Ω L1: D1: C1: C2: Murata LQH4C100K04 Motorola MBR0530 Tayo Yuden JMK212BY475MG Tayo Yuden TMK316BJ105KL Figure 22. White LED Supply With Adjustable Brightness Control Using a PWM Signal on the Enable Pin, Efficiency Approx. Equals 86% at VIN = 3 V, ILED = 15 mA L1 10 µH VCC = 2.7 V to 6 V C1 4.7 µF VIN SW C2 100 nF (See Note A) D2 24 V (Optional) FB EN R1 120 kΩ GND Analog Brightness Control 3.3 V ≅ Led Off 0 V ≅ Iled = 20 mA A. D1 MBRM120L R2 160 kΩ RS 110 Ω L1: D1: C1: C2: Murata LQH4C3R3M04 Motorola MBR0530 Tayo Yuden JMK212BY475MG Standard Ceramic Capacitor A smaller output capacitor value for C2 causes a larger LED ripple. Figure 23. White LED Supply With Adjustable Brightness Control Using an Analog Signal on the Feedback Pin 16 XB61040 SOT23-5,XB61041 SOT23-5 SOT23-5封装尺寸图 17