SILICONCONTENT
TECHNOLOGY
SCT2321
3.8V-32V Vin, 2A Synchronous Step-down DCDC Converter with EMI Reduction
FEATURES
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DESCRIPTION
EMI Reduction with Switching Node Ringing-free
500kHz Switching Frequency
Force Pulse Width Modulation (FPWM) Mode
3.8V-32V Wide Input Voltage Range
Up to 2A Continuous Output Load Current
0.8V ±1% Feedback Reference Voltage
Fully Integrated 130mΩ Rdson High Side MOSFET
and 70mΩ Rdson Low Side MOSFET
1uA Shut-down Current
80ns Minimum On-time
Precision Enable Threshold for Programmable
UVLO Threshold and Hysteresis
Low Dropout Mode Operation
4ms Built-in Soft Start Time
Output Over Voltage Protection
Thermal Shutdown Protection at 160°C
Available in TSOT23-6L Package
The SCT2321 is an Electromagnetic Interference
(EMI) friendly buck converter with implementing
optimized design for EMI reduction. The converter
has proprietary designed gate driver scheme to resist
switching node ringing without sacrificing MOSFET
turn-on and turn-off time, which erases high
frequency radiation EMI noise caused by the
MOSFETs hard switching.
The SCT2321 offers output over-voltage protection,
cycle-by-cycle peak current limit, and thermal
shutdown protection. The device is available in a lowprofile TSOT23-6 package.
APPLICATIONS
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The SCT2321 is a 2A synchronous buck converter
with up to 32V wide input voltage range, which fully
integrates a 130mΩ high-side MOSFET and a 70mΩ
low-side MOSFET to provide high efficiency stepdown DCDC conversion. The SCT2321 adopts peak
current mode control with the integrated
compensation network, which makes SCT2321 easily
to be used by minimizing the off-chip component
count. The SCT2321 supports Force Pulse Width
Modulation (FPWM) Mode to achieve the small output
ripple at light load condition.
White Goods, Home Appliance
Surveillance
Audio, WiFi Speaker
Printer, Charging Station
DTV, STB, Monitor/LCD Display
TYPICAL APPLICATION
R2
ON
OFF
VIN
C1
FB
BST
EN
SW
VIN
GND
C2
L1
VOUT
C3
Efficiency (%)
R1
100
90
80
70
60
50
40
30
20
10
0
SCT2321, VOUT=3.3V
SCT2321, VOUT=5V
1
10
100
1000
Output Current (mA)
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1
SCT2321
REVISION HISTORY
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Revision 1.1 Released to Production. Complete orderable information.
Revision 1.2 Released to Production. Updated application waveforms.
DEVICE ORDER INFORMATION
PART NUMBER
PACKAGE MARKING
PACKAGE DISCRIPTION
SCT2321TVB
2321
TSOT23-6L
1）For Tape & Reel, Add Suffix R (e.g. SCT2321TVBR)
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
Over operating free-air temperature unless otherwise noted(1)
(1)
(2)
DESCRIPTION
MIN
MAX
UNIT
BST
-0.3
40
V
VIN, SW, EN
-0.3
34
VS, FB
-0.3
Operating junction temperature(2)
Storage temperature TSTG
FB
1
6
BST
V
EN
2
5
SW
5.5
V
VIN
3
4
GND
-40
125
C
-65
150
C
Top View: TSOT23-6L, Plastic
Stresses beyond those listed under Absolute Maximum Rating may cause device permanent damage. The device is not guaranteed to
function outside of its Recommended Operation Conditions.
The IC includes over temperature protection to protect the device during overload conditions. Junction temperature will exceed 150°C
when over temperature protection is active. Continuous operation above the specified maximum operating junction temperature will
reduce lifetime
PIN FUNCTIONS
2
NAME
NO.
PIN FUNCTION
FB
1
Buck converter output feedback sensing voltage. Connect a resistor divider from
VOUT to FB to set up output voltage. The device regulates FB to the internal
reference of 0.8V typical.
EN
2
Enable logic input. Floating the pin enables the device. This pin supports high
voltage input up to VIN supply to be connected VIN directly to enable the device
automatically. The device has precision enable thresholds 1.18V rising / 1,1V
falling for programmable UVLO threshold and hysteresis.
VIN
3
Power supply input. Must be locally bypassed.
GND
4
Power ground. Must be soldered directly to ground plane.
SW
5
Switching node of the buck converter.
BST
6
Power supply for the high-side power MOSFET gate driver. Must connect a 0.1uF
or greater ceramic capacitor between BST pin and SW node.
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SCT2321
RECOMMENDED OPERATING CONDITIONS
Over operating free-air temperature range unless otherwise noted
PARAMETER
DEFINITION
VIN
TJ
Input voltage range
Operating junction temperature
MIN
MAX
UNIT
3.8
-40
32
125
V
°C
MIN
MAX
UNIT
-2
+2
kV
-0.5
+0.5
kV
ESD RATINGS
PARAMETER
VESD
DEFINITION
Human Body Model(HBM), per ANSI-JEDEC-JS-0012014 specification, all pins(1)
Charged Device Model(CDM), per ANSI-JEDEC-JS-0022014specification, all pins(1)
(1) HBM and CDM stressing are done in accordance with the ANSI/ESDA/JEDEC JS-001-2014 specification
THERMAL INFORMATION
PARAMETER
RθJA
RθJC
THERMAL METRIC
Junction to ambient thermal resistance(1)
Junction to case thermal
resistance(1)
TSOT23-6L
89
39
UNIT
°C/W
(1) SCT provides RθJA and RθJC numbers only as reference to estimate junction temperatures of the devices. RθJA and RθJC are not a
characteristic of package itself, but of many other system level characteristics such as the design and layout of the printed circuit
board (PCB) on which the SCT2321 is mounted, and external environmental factors. The PCB board is a heat sink that is soldered to
the leads and thermal pad of the SCT2321. Changing the design or configuration of the PCB board changes the efficiency of the heat
sink and therefore the actual RθJA and RθJC.
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SCT2321
ELECTRICAL CHARACTERISTICS
VIN=12V, TJ=-40°C~125°C, typical values are tested under 25°C.
SYMBOL
PARAMETER
TEST CONDITION
Power Supply and Output
VIN
Operating input voltage
ISD
Input UVLO
Hysteresis
Shutdown current
IQ
Quiescent current
VIN_UVLO
MIN
TYP
MAX
32
V
3.5
420
1
3.7
V
mV
uA
3.8
VIN rising
EN=0, No load, VIN=12V
EN=floating, No load, No
switching. VIN=12V. BSTSW=5V
250
Enable, Soft Start and Working Modes
VEN_H
Enable high threshold
1.18
VEN_L
Enable low threshold
IEN
Enable pin input current
EN=1V
IEN_HYS
Enable pin hysteresis current
EN=1.5V
3
1.03
1.1
1
1.5
UNIT
uA
1.25
V
V
2
uA
4
uA
Power MOSFETs
RDSON_H
High side FET on-resistance
130
mΩ
RDSON_L
70
mΩ
Low side FET on-resistance
Feedback and Error Amplifier
VFB
Feedback Voltage
0.792
0.8
0.808
V
Current Limit
ILIM_HSD
HSD peak current limit
2.5
2.8
3.1
A
ILIM_LSD
2.8
3.2
3.6
A
450
500
550
kHz
LSD valley current limit
Switching Frequency
FSW
Switching frequency
tON_MIN
VIN=12V, VOUT=5V
Minimum on-time
80
ns
4
ms
Soft Start Time
tSS
Internal soft-start time
Protection
VOVP
THIC_W
THIC_R
TSD
4
Output OVP threshold
Hysteresis
OCP hiccup wait time
OCP hiccup restart time
Thermal shutdown threshold
Hysteresis
VOUT rising
TJ rising
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110
5
512
8192
160
25
All Rights Reserved
%
%
Cycles
Cycles
°C
SCT2321
100
90
80
70
60
50
40
30
20
10
0
Efficiency (%)
Efficiency (%)
TYPICAL CHARACTERISTICS
VIN=12V, VOUT=3.3V
VIN=12V, VOUT=5V
1
10
100
100
90
80
70
60
50
40
30
20
10
0
1000
VIN=24V, VOUT=3.3V
VIN=24V, VOUT=5V
1
10
Output Current (mA)
Figure 1. SCT2321 Efficiency, Vin=12V
1000
Figure 2. SCT2321 Efficiency, Vin=24V
2
0.9
1.5
0.85
VREF (V)
I_SD (uA)
100
Output Current (mA)
1
0.8
0.75
0.5
VREF
I_SD
0.7
0
-50
-50
0
50
100
0
150
50
100
150
Temperature (°C)
Temperature (°C)
Figure 4. Reference Voltage vs Temperature
Figure 3. Shut-down Current vs Temperature
3.6
4
3.5
3.4
VIN (V)
Current (A)
3.5
3
3.3
UVLO RISING
3.2
UVLO FALLING
3.1
2.5
ILIM_LSD
3
ILIM_HSD
2.9
2
-50
0
50
100
150
-50
0
Figure 5. Peak Current Limit vs Temperature
50
100
Temperature(°C)
Temperature (°C)
Figure 6. VIN UVLO vs Temperature
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150
5.07
5.06
5.05
5.04
5.03
5.02
5.01
5
4.99
4.98
4.97
4.96
5.08
5.06
Output Voltage (v)
Output Voltage (v)
SCT2321
VIN=12V
5.02
5
4.98
4.96
VIN=24V
4.94
0
500
1000
1500
Output Current (mA)
2000
Figure 7. Load Regulation, (VOUT=5V)
6
SCT2321
5.04
0
10
Figure 8. Line Regulation (IOUT=2A)
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20
30
Input Voltage (V)
All Rights Reserved
40
SCT2321
FUNCTIONAL BLOCK DIAGRAM
VIN
3
4uA
1.5uA
UVLO
20K
EN
2
+
VIN UVLO
and LDO
EN
VCC
1.21V
VCC
HS MOSFET
Current Limit
BOOT
UVLO
Ramp
SS/4ms
+
+ GM
0.8V
FB
1
BOOT
Strap
PWM
+
COMP
6
BST
5
SW
4
GND
Q1
18k
PWM and Dead
Time Control
Logic
7.6nF
+
OVP
0.88V
Q2
Oscillator
with PLL
Thermal
Protection
CLK
LS MOSFET
Current Limit
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SCT2321
OPERATION
Overview
The SCT2321 device are 3.8V-32V input, 2A output, EMI friendly, fully integrated synchronous buck converters.
The device employs fixed frequency peak current mode control. An internal clock with 500kHz frequency initiates
turning on the integrated high-side power MOSFET Q1 in each cycle, then inductor current rises linearly and the
converter charges output cap. When sensed voltage on high-side MOSFET peak current rising above the voltage
of internal COMP (see functional block diagram), the device turns off high-side MOSFET Q1 and turns on low-side
MOSFET Q2. The inductor current decreases when MOSFET Q2 is ON. In the next rising edge of clock cycle, the
low-side MOSFET Q2 turns off. This repeats on cycle-by-cycle based.
The peak current mode control with the internal loop compensation network and the built-in 4ms soft-start simplify
the SCT2321 footprints and minimize the off-chip component counts.
The error amplifier serves the COMP node by comparing the voltage on the FB pin with an internal 0.8V reference
voltage. When the load current increases, a reduction in the feedback voltage relative to the reference raises COMP
voltage till the average inductor current matches the increased load current. This feedback loop well regulates the
output voltage. The device also integrates an internal slope compensation circuitry to prevent sub-harmonic
oscillation when duty cycle is greater than 50% for a fixed frequency peak current mode control.
The SCT2321 converter has optimized gate driver scheme to achieve switching node voltage ringing-free without
sacrificing the MOSFET switching time to further damping high frequency radiation EMI noise.
To provide the lower output ripple in light load condition, the SCT2321 offers the fixed 500kHz switching frequency
and works at the Force Pulse Width Modulation (FPWM) mode.
The hiccup mode minimizes power dissipation during prolonged output overcurrent or short conditions. The hiccup
wait time is 512 cycles and the hiccup restart time is 8192 cycles. The SCT2321 device also feature full protections
including cycle-by-cycle high-side MOSFET peak current limit, over-voltage protection, and over-temperature
protection.
VIN Power
The SCT2321 is designed to operate from an input voltage supply range between 3.8V to 32V, at least 0.1uF
decoupling ceramic cap is recommended to bypass the supply noise. If the input supply locates more than a few
inches from the converter, an additional electrolytic or tantalum bulk capacitor or with recommended 22uF may be
required in addition to the local ceramic bypass capacitors.
Under Voltage Lockout UVLO
The SCT2321 Under Voltage Lock Out (UVLO) default startup threshold is typical 3.5V with VIN rising and shutdown
threshold is 3.1V with VIN falling. The more accurate UVLO threshold can be programmed through the precision
enable threshold of EN pin.
Enable and Start up
When applying a voltage higher than the EN high threshold (typical 1.18V/rise), the SCT2321 enable all functions
and the device starts soft-start phase. The SCT2321 have the built in 4ms soft-start time to prevent the output
overshoot and inrush current. When EN pin is pulled low, the internal SS net will be discharged to ground. Buck
operation is disabled when EN voltage falls below its lower threshold (typically 1.1V/fall).
An internal 1.5uA pull up current source connected from internal LDO power rail to EN pin guarantees that floating
8
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EN pin automatically enables the device. For the application requiring higher VIN UVLO voltage than the default
setup, there is a 4uA hysteresis pull up current source on EN pin which configures the VIN UVLO voltage with an
off-chip resistor divider R3 and R4, shown in Figure 9. The resistor divider R3 and R4 are calculated by equation
(1) and (2).
EN pin is a high voltage pin, and can be directly connected to VIN to automatically start up the device with VIN
rising to its internal UVLO threshold.
VIN
I2
4uA
I1
1.5uA
R3
20K
EN
+
EN
1.21V
R4
Figure 9. Adjustable VIN UVLO
𝑅3 =
𝑅4 =
𝑉𝑆𝑡𝑎𝑟𝑡 (
𝑉𝐸𝑁𝐹
) − 𝑉𝑆𝑡𝑜𝑝
𝑉𝐸𝑁𝑅
𝑉𝐸𝑁𝐹
𝐼1 (1 −
𝑉𝐸𝑁𝑅
) + 𝐼2
𝑅3 × 𝑉𝐸𝑁𝐹
𝑉𝑆𝑡𝑜𝑝 − 𝑉𝐸𝑁𝐹 + 𝑅3 (𝐼1 + 𝐼2 )
(1)
(2)
Where:
Vstart: Vin rise threshold to enable the device
Vstop: Vin fall threshold to disable the device
I1=1.5uA
I2=4uA
VENR=1.18V
VEMF=1.1V
EMI Reduction with Switching Node Ringing-free
In buck converter, the switching node ringing amplitude and cycles are critical especially related to the high
frequency radiation EMI noise. The SCT2321 implements the multi-level gate driver speed technique to achieve the
switching node ringing-free without scarifying the switching node rise/fall slew rate and power efficiency of the
converter. The switching node ringing amplitude and cycles are damped by the built-in MOSFETs gate driving
technique (SCT Patented Proprietary Design). The switching node zoomed in wave form is shown in Figure 10.
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Figure 10. SCT2321 Switching Node Waveform
Peak Current Limit and Hiccup Mode
The SCT2321 have cycle-by-cycle peak current limit with sensing the internal high side MOSFET Q1 current during
overcurrent condition. While the Q1 turns on, its conduction current is monitored by the internal sensing circuitry.
Once the high-side MOSFET Q1 current exceeds the limit, it turns off immediately. If the Q1 over current time
exceeds 512 switching cycles (hiccup waiting time), the buck converter enters hiccup mode and shuts down. After
8192 cycles off, the buck converter restarts to power up. The hiccup modes reduce the power dissipation in over
current condition.
Over Voltage Protection and Minimum On-time
The SCT2321 feature buck converter output over voltage protection (OVP). If the output feedback pin voltage
exceeds110% of feedback reference voltage (0.8V), the converter stops switching immediately. When the output
feedback pin voltage drops below 105% of feedback reference voltage, the converter resumes to switching. The
OVP function prevents the connected output circuitry damaged from un-predictive overvoltage. Featured feedback
overvoltage protection also prevents dynamic voltage spike to damage the circuitry at load during fast loading
transient.
The high-side MOSFET Q1 has minimum on-time 80ns typical limitation. While the device operates at minimum ontime, further increasing VIN results in pushing output voltage beyond regulation point. With output feedback over
voltage protection, the converter skips pulse by turning off high-side MOSFET Q1 and prevents output running away
higher to damage the load.
Force Pulse Width Modulation (FPWM) Working Modes
To provide the lower output ripple in light load condition, the SCT2321 offers the fixed 500kHz switching frequency
and works at the Force Pulse Width Modulation (FPWM) mode.
Bootstrap Voltage Regulator
An external bootstrap capacitor between BST and SW pin powers floating high-side power MOSFET gate driver.
The bootstrap capacitor voltage is charged from an integrated voltage regulator when high-side power MOSFET is
off and low-side power MOSFET is on.
The floating supply (BST to SW) UVLO threshold is 2.7V rising and hysteresis of 350mV. When the converter
operates with high duty cycle or prolongs in sleep mode for certain long time, the required time interval to recharging
bootstrap capacitor is too long to keep the voltage at bootstrap capacitor sufficient. When the voltage across
bootstrap capacitor drops below 2.35V, BST UVLO occurs. The SCT2321 intervenes to turn on low side MOSFET
periodically to refresh the voltage of bootstrap capacitor to guarantee operation over a wide duty range.
10
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Thermal Shutdown
Once the junction temperature in the SCT2321 exceeds 160°C, the thermal sensing circuit stops converter
switching and restarts with the junction temperature falling below 125°C. Thermal shutdown prevents the damage
on device during excessive heat and power dissipation condition.
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SCT2321
APPLICATION INFORMATION
Typical Application
C5
68pF
R1
158k
C4
0.1uF
R2
30k
ON
OFF
VIN=3.8V~32V
C1
10uF
FB
BST
EN
SW
VIN
GND
L1
10uH
VOUT=5V
C3
3 x 22uF
C2
0.1uF
Figure 11. 24V Input, 5V/2A Output
Design Parameters
12
Design Parameters
Example Value
Input Voltage
24V
Output Voltage
5V
Output Current
2A
Output voltage ripple (peak to peak)
±0.3V
Switching Frequency
500kHz
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Input Capacitor Selection
For good input voltage filtering, choose low-ESR ceramic capacitors. A ceramic capacitor 10μF is recommended
for the decoupling capacitor anda0.1μF ceramic bypass capacitor is recommended to be placed as close as
possible to the VIN pin of the SCT2321.
Use Equation (3) to calculate the input voltage ripple:
∆𝑉𝐼𝑁 =
𝐼𝑂𝑈𝑇
VOUT
𝑉𝑂𝑈𝑇
×
× (1 −
)
𝐶𝐼𝑁 × 𝑓𝑆𝑊
VIN
𝑉𝐼𝑁
(3)
Where:
•
CIN is the input capacitor value
•
fsw is the converter switching frequency
•
IOUT is the maximum load current
Due to the inductor current ripple, the input voltage changes if there is parasitic inductance and resistance between
the power supply and the VIN pin. It is recommended to have enough input capacitance to make the input voltage
ripple less than 100mV. Generally, a 35V/10uF input ceramic capacitor is recommended for most of applications.
Choose the right capacitor value carefully with considering high-capacitance ceramic capacitors DC bias effect,
which has a strong influence on the final effective capacitance.
Inductor Selection
The performance of inductor affects the power supply’s steady state operation, transient behavior, loop stability,
and buck converter efficiency. The inductor value, DC resistance (DCR), and saturation current influences both
efficiency and the magnitude of the output voltage ripple. Larger inductance value reduces inductor current ripple
and therefore leads to lower output voltage ripple. For a fixed DCR, a larger value inductor yields higher efficiency
via reduced RMS and core losses. However, a larger inductor within a given inductor family will generally have a
greater series resistance, thereby counteracting this efficiency advantage.
Inductor values can have ±20% or even ±30% tolerance with no current bias. When the inductor current approaches
saturation level, its inductance can decrease 20% to 35% from the value at 0-A current depending on how the
inductor vendor defines saturation. When selecting an inductor, choose its rated current especially the saturation
current larger than its peak current during the operation.
To calculate the current in the worst case, use the maximum input voltage, minimum output voltage, maxim load
current and minimum switching frequency of the application, while considering the inductance with -30% tolerance
and low-power conversion efficiency.
For a buck converter, calculate the inductor minimum value as shown in equation (4).
𝐿𝐼𝑁𝐷𝑀𝐼𝑁 =
(4)
𝑉𝑂𝑈𝑇 × (𝑉𝐼𝑁𝑀𝐴𝑋 − 𝑉𝑂𝑈𝑇 )
𝑉𝐼𝑁𝑀𝐴𝑋 × 𝐾𝐼𝑁𝐷 × 𝐼𝑂𝑈𝑇 × 𝑓𝑆𝑊
Where:
•
KIND is the coefficient of inductor ripple current relative to the maximum output current.
Therefore, the peak switching current of inductor, ILPEAK, is calculated as in equation (5).
𝐼𝐿𝑃𝐸𝐴𝐾 = 𝐼𝑂𝑈𝑇 + 𝐾𝐼𝑁𝐷 ×
𝐼𝑂𝑈𝑇
2
(5)
Set the current limit of the SCT2321 higher than the peak current ILPEAK and select the inductor with the saturation
current higher than the current limit. The inductor’s DC resistance (DCR) and the core loss significantly affect the
efficiency of power conversion. Core loss is related to the core material and different inductors have different core
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SCT2321
loss. For a certain inductor, larger current ripple generates higher DCR and ESR conduction losses and higher core
loss.
Table 1 lists recommended inductors for the SCT2321. Verify whether the recommended inductor can support the
user's target application with the previous calculations and bench evaluation. In this application, the WE's inductor
744314101 is used on SCT2321 evaluation board.
Table 1. Recommended Inductors
Part Number
L
(uH)
DCR Max
(mΩ)
Saturation Current/Heat
Rating Current (A)
Size Max
(LxWxH mm)
Vendor
744314101
10
33
3.5
7x7x5
Wurth Electronik
Output Capacitor Selection
For buck converter, the output capacitor value determines the regulator pole, the output voltage ripple, and how the
regulator responds to a large change in load current. The output capacitance needs to be selected based on the
most stringent of these three criteria.
For small output voltage ripple, choose a low-ESR output capacitor like a ceramic capacitor, for example, X5R and
X7R family. Typically, 1～3x 22μF ceramic output capacitors work for most applications. Higher capacitor values
can be used to improve the load transient response. Due to a capacitor’s de-rating under DC bias, the bias can
significantly reduce capacitance. Ceramic capacitors can lose most of their capacitance at rated voltage. Therefore,
leave margin on the voltage rating to ensure adequate effective capacitance.
From the required output voltage ripple, use the equation (6) to calculate the minimum required effective
capacitance, COUT.
𝐶𝑂𝑈𝑇 =
∆𝐼𝐿𝑃𝑃
8 × 𝑉𝑂𝑈𝑇𝑅𝑖𝑝𝑝𝑙𝑒 × 𝑓𝑆𝑊
(6)
Where
• VOUTRipple is output voltage ripple caused by charging and discharging of the output capacitor.
• ΔILPP is the inductor peak to peak ripple current, equal to kIND * IOUT.
• ƒSW is the converter switching frequency.
The allowed maximum ESR of the output capacitor is calculated by the equation (7).
𝑅𝐸𝑆𝑅 =
𝑉𝑂𝑈𝑇𝑅𝑖𝑝𝑝𝑙𝑒
∆𝐼𝐿𝑃𝑃
(7)
The output capacitor affects the crossover frequency ƒC. Considering the loop stability and effect of the internal loop
1
compensation parameters, choose the crossover frequency less than 55 kHz ( × fSW ) without considering the
10
feed-forward capacitor. A simple estimation for the crossover frequency without feed forward capacitor is shown in
equation (8), assuming COUT has small ESR.
𝐶𝑂𝑈𝑇 >
18𝑘 × 𝐺𝑀 × 𝐺𝑀𝑃 × 0.8𝑉
2𝜋 × 𝑉𝑂𝑈𝑇 × 𝑓𝐶
(8)
Where
• GM is the transfer conductance of the error amplifier (300uS).
• GMP is the gain from internal COMP to inductor current, which is 5A/V.
14
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•
fC is the cross over frequency.
Additional capacitance de-rating for aging, temperature and DC bias should be factored in which increases this
minimum value. Capacitors generally have limits to the amount of ripple current they can handle without failing or
producing excess heat. An output capacitor that can support the inductor ripple current must be specified. The
capacitor data sheets specify the RMS (Root Mean Square) value of the maximum ripple current. Equation (9) can
be used to calculate the RMS ripple current the output capacitor needs to support.
𝐼𝐶𝑂𝑈𝑇𝑅𝑀𝑆 =
𝑉𝑂𝑈𝑇 ∙ (𝑉𝐼𝑁 − 𝑉𝑂𝑈𝑇 )
(9)
√12 ∙ 𝑉𝐼𝑁 ∙ 𝐿𝐼𝑁𝐷 ∙ 𝑓𝑆𝑊
Output Feed-Forward Capacitor Selection
The SCT2321 has the internal integrated loop compensation as shown in the function block diagram. The
compensation network includes a 18k resistor and a 7.6nF capacitor. Usually, the type II compensation network
has a phase margin between 60 and 90 degree. However, if the output capacitor has ultra-low ESR, the converter
results in low phase margin. To increase the converter phase margin, a feed-forward cap Cff is used to boost the
phase margin at the converter cross-over frequency fc. Equation (10) is used to calculate the feed-forward capacitor.
𝐶𝑓𝑓 =
1
2𝜋 ∙ 𝑓𝐶 × 𝑅1
(10)
Output Feedback Resistor Divider Selection
The SCT2321 features external programmable output voltage by using a resistor divider network R1 and R2 as
shown in the typical application circuit Figure17. Use equation (11) to calculate the resistor divider values.
𝑅1 =
(𝑉𝑂𝑈𝑇 − 𝑉𝑟𝑒𝑓 ) × 𝑅2
𝑉𝑟𝑒𝑓
(11)
Set the resistor R2 value to be approximately 30k. Slightly increasing or decreasing R1 can result in closer output
voltage matching when using standard value resistors.
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SCT2321
Application Waveforms
16
Figure 12. SW node waveform and Output Ripple
VIN=24V, IOUT=2A
Figure 13. SW node Waveform and Output Ripple
VIN=24V, IOUT=10mA
Figure 14. Power Up
VIN=24V, VOUT=5V, IOUT=2A
Figure 15. Power Down
VIN=24V, VOUT=5V, IOUT=2A
Figure 16. Load Transient
VOUT=5V, IOUT=0.2A to 1.8A, SR=250mA/us
Figure 17. Load Transient
VOUT=5V, IOUT=0.5A to 1.5A, SR=250mA/us
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Layout Guideline
The regulator could suffer from instability and noise problems without carefully layout of PCB. Radiation of highfrequency noise induces EMI, so proper layout of the high-frequency switching path is essential. Minimize the length
and area of all traces connected to the SW pin, and always use a ground plane under the switching regulator to
minimize coupling. The input capacitor needs to be very close to the VIN pin and GND pin to reduce the input supply
ripple. Place the capacitor as close to VIN pin as possible to reduce high frequency ringing voltage on SW pin as
well. Figure 18 is the recommended PCB layout of SCT2321.
The layout needs be done with well consideration of the thermal. A large top layer ground plate using multiple
thermal vias is used to improve the thermal dissipation. The bottom layer is a large ground plane connected to the
top layer ground by vias.
FB
BST
EN
SW
VIN
GND
VOUT
Figure 18. PCB Layout Example
Thermal Considerations
The maximum IC junction temperature should be restricted to 125°C under normal operating conditions. Calculate
the maximum allowable dissipation, PD(max), and keep the actual power dissipation less than or equal to PD(max) . The
maximum-power-dissipation limit is determined using Equation (12).
𝑃𝐷(𝑀𝐴𝑋) =
125 − 𝑇𝐶𝐴
𝑅θJA
(12)
where
• TA is the maximum ambient temperature for the application.
• RθJA is the junction-to-ambient thermal resistance given in the Thermal Information table.
The real junction-to-ambient thermal resistance RθJA of the package greatly depends on the PCB type, layout,
thermal pad connection and environmental factor. Using thick PCB copper and soldering the GND to a large ground
plate enhance the thermal performance. Using more vias connects the ground plate on the top layer and bottom
layer around the IC without solder mask also enhance the thermal capability.
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SCT2321
PACKAGE INFORMATION
TOP VIEW
BOTTOM VIEW
SYMBOL
SIDE VIEW
NOTE:
1.
2.
3.
4.
5.
6.
18
Drawing proposed to be made a JEDEC package outline MO220 variation.
Drawing not to scale.
All linear dimensions are in millimeters.
Thermal pad shall be soldered on the board.
Dimensions of exposed pad on bottom of package do not
include mold flash.
Contact PCB board fabrication for minimum solder mask web
tolerances between the pins.
A
A1
A2
D
E
E1
b
c
e
L
ɵ
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Unit: Millimeter
MIN
TYP
MAX
------1.10
0.000
0.10
0.70
1.00
2.85
2.95
2.65
2.95
1.55
1.65
0.30
0.50
0.08
0.20
0.95(BSC)
0.30
0.60
0º
8º
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SCT2321
TAPE AND REEL INFORMATION
Feeding Direction
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SCT2321
TYPICAL APPLICATION
24V Vin, 5V Vout, 1.1MHz Synchronous Buck Converter
Power Efficiency
100
BST
EN
SW
VIN
GND
C2
0.1uF
90
L1
10uH
VOUT=5V
VIN=12V
C1
10uF
C3
22uF
Efficiency (%)
ON
OFF
VS
80
70
VIN=12V, SCT2323
VIN=24V, SCT2323
VIN=12V, SCT2325
60
C2
0.1uF
VIN=24V, SCT2325
50
1
10
100
1000
Output Current (mA)
RELATED PARTS
PN
DESCRIPTION
COMMENTS
SCT2325
SCT2323
3.8V-32V Vin, Fixed Vout, 2A
Synchronous Buck Converter with
EMI Reduction
SCT2330
SCT2331
3.8V-32V Vin, 3A Synchronous
Step-down DCDC Converter with
EMI Reduction
•
•
•
•
•
•
•
•
•
•
•
•
ON
OFF
VS
BST
EN
SW
VIN
GND
1.1MHz switching frequency with ±6% FSS
EMI reduction with switching node ringing-free.
SW anti-ringing in discontinuous current mode
20uA ultra-low quiescent current
Minimum external components. Easy-to-use
Fixed 5V Vout (SCT2325) and 3.3V Vout (SCT2323)
500KHz switching frequency
3A Continuous output current
EMI reduction with switching node ringing-free.
Ultra-low quiescent current. High efficiency PFM at
light load (SCT2330)
Frequency Spread Spectrum (SCT2330)
Fixed PWM mode for lower output ripple (SCT2331)
C2
0.1uF
L1
10uH
VOUT=5V
VIN=4V-28V
C1
10uF
C3
22uF
C2
0.1uF
Figure 25. SCT2325 Typical Application
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee the third
party Intellectual Property rights are not infringed upon when integrating Silicon Content Technology (SCT) products into any
application. SCT will not assume any legal responsibility for any said applications.
20
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