TIPD135

Ian Williams TI Precision Designs: Verified Design 10 µA-100 mA, 0.05% Error, High-Side Current Sensing TI Precision Designs Circuit Description TI Precision Designs are analog solutions created by TI’s analog experts. Verified Designs offer the theory, component selection, simulation, complete PCB schematic & layout, bill of materials, and measured performance of useful circuits. Circuit modifications that help to meet alternate design goals are also discussed. This split-supply, high-side current sensing solution accurately detects load currents from 10 µA to 100 mA. The linear range of the differential voltage output is from -4.9V to +4.9 V. An instrumentation amplifier with pin-programmable gain was selected in order to measure the four-decade load current range. Design Resources Design Archive TINA-TI™ PGA281 TPS7A4101 Ask The Analog Experts WEBENCH® Design Center TI Precision Designs Library All Design files SPICE Simulator Product Folder Product Folder +15V VIN+ +5V VOUT+ + VSP VSOP VDD VOP VBUS ILOAD RSH To ADC PGA281 VON VIN- ILOAD RLOAD -15V VOCM VSON VOUT- VSN +2.5V An IMPORTANT NOTICE at the end of this TI reference design addresses authorized use, intellectual property matters and other important disclaimers and information. TINA-TI is a trademark of Texas Instruments WEBENCH is a registered trademark of Texas Instruments TIDU033-September 2013-Revised September 2013 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution Copyright © 2013, Texas Instruments Incorporated 1 www.ti.com 1 Design Summary The design requirements are as follows:  Supply Voltage: ±15 V  Input: 10 µA – 100 mA (bidirectional)  Output: ± 4.9 V (differential)  Maximum Shunt Voltage: 700 mV The design goals and performance are summarized in Table 1. Figure 1 depicts the measured transfer function of the design. Table 1. Comparison of Design Goals, Simulation, and Measured Performance Goal Simulated Measured Error (%FSRerror) 0.05% 0.002% 0.0481% Relative Error (ILOAD = 10 µA, G = 176) 35.0% 30.47% 33.16% Relative Error (ILOAD = 100 µA, G = 176) 5% 3.13% 3.15% Relative Error (ILOAD = 1 mA, G = 176) 1% 0.31% 0.31% Relative Error (ILOAD = 10 mA, G = 5.5) 0.1% 0.09% 0.09% Relative Error (ILOAD = 100 mA, G = 5.5) 0.01% 0.01% 0.01% 5.0 4.5 4.0 Output Voltage (V) 3.5 3.0 2.5 2.0 G = 5.5 V/V G = 176 V/V 1.5 1.0 0.5 0.0 0.00 0.01 0.02 0.03 0.04 0.05 0.06 Load Current (A) 0.07 0.08 0.09 0.10 Figure 1: Measured Transfer Function 2 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution TIDU033-September 2013-Revised September 2013 Copyright © 2013, Texas Instruments Incorporated www.ti.com 2 Theory of Operation A more complete schematic for this design is shown in Figure 2. The transfer function of the circuit is based on the relationship between the load current ILOAD, the shunt resistance RSH, and the gain blocks inside the PGA281 which are controlled by the switch SW 1. In this high-side current sense application, a power supply VBUS is connected to the load represented by RLOAD. RLOAD draws current ILOAD from VBUS which flows through RSH, developing a voltage drop across RSH as defined by Ohm’s Law. This differential voltage is filtered and applied to the input of the PGA, where it is amplified and output differentially with a common-mode voltage equal to half of its output stage supply voltage, or VSOP/2. VSP VSOP C2 C1 C3 VIN+ + VSP G4 C4 R11 VBUS C6 R13 G1 RLOAD PGA281 VOCM VSON G0 VIN- R14 VON G1 C8 R12 C5 VOP G2 G0 ILOAD VOUT+ VDD G3 G2 RSH VSOP G4 G3 C7 To ADC VOUTC11 VSN C9 R15 C10 R16 VSN VSOP VSOP (+5V) VSP (+15V) IN +15V SW1 TPS7A4101 R17 R2 R3 R4 R5 R6 R7 R8 R9 R10 OUT EN D1 G4 R1 G3 C12 R19 C13 FB C14 GND R20 G2 G1 G0 -15V D2 R21 C15 D3 Open à GN = 0 Closed à GN = 1 VSN (-15V) Figure 2: Complete Circuit Schematic The transfer function for this design is defined by the following equation: (1) TIDU033-September 2013-Revised September 2013 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution Copyright © 2013, Texas Instruments Incorporated 3 www.ti.com 2.1 High-Side Current Sensing In high-side current sensing applications, the sense resistor is placed between the supply voltage V BUS and the load RLOAD. High-side sensing is desirable in that it directly monitors the current delivered by the supply, which allows for the detection of load shorts. The most common challenge when using this topology is that the amplifier’s input common-mode voltage range must include the load’s supply voltage, or VBUS. With ±15 V supplies the PGA281 can accept common-mode voltages of ±12.5 V, making it a suitable choice for high-side current sensing as long as VBUS falls within the common-mode voltage range. Figure 3 depicts a typical high-side current sensing scenario. VCM ≈ VBUS Differential Amplifier VBUS + – + VSH - RSH + + VOUT - System Load Figure 3: High-Side Current Sensing Topology As shown in Figure 3, the value of VSH is the voltage drop across the shunt resistor RSH. If the value of VSH is too large, then the voltage actually delivered to the load may not meet the load’s minimum requirements. Therefore it is important to limit the voltage drop across the shunt resistor. Equation 2 can be used to calculate the maximum value of RSH. (2) It is recommended to use the maximum shunt resistance to minimize relative error at minimum load current. Relative error is discussed in Section 4.4. Based on the availability of resistors at major distributors, a value of 6.8 Ω was chosen for RSH. The gain(s) required for this design depend on the maximum output swing of the amplifier, shunt resistor, and the load current range. It is recommended to use the maximum gain to ensure full utilization of the linear operating range of the device. Equation 3 shows how to calculate the maximum gain for the maximum load current. (3) The closest available gain setting on the PGA281 (without exceeding the maximum) is 5.5 V/V. To determine if the design requires more than one gain, the minimum load current that can be measured given GI_LOAD(max) and VOUT(min) must be calculated as shown in Equation 4 and Equation 5. 4 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution TIDU033-September 2013-Revised September 2013 Copyright © 2013, Texas Instruments Incorporated www.ti.com (4) (5) Since ILOAD(min) does not meet the minimum load current specification of 10 µA, a second gain is required. It is recommended to now calculate the maximum gain for the minimum load current (G I_LOAD(min)) as shown in Equation 6. (6) The closest available gain setting on the PGA281 is 176 V/V. Equation 7 and Equation 8 show how to calculate the maximum load current that can be measured given GI_LOAD(min). (7) (8) Since the minimum load current that can be measured when G = 5.5 V/V overlaps with the maximum load current for G = 176 V/V, only two gains are required to measure the entire load current range. 2.2 Input and Output Filtering In order to minimize noise at the input of the PGA281, a low-pass filter network is placed between the sense resistor and the PGA281 input pins. Figure 4 shows the input filter schematic. R11 VSH VIN+ C4 + – C6 R13 C8 VIN- Figure 4: Input Filter Schematic The input filter has both a common-mode component and a differential component. Since the shunt voltage signal is dc, the cutoff frequencies of this filter can be set very low in order to attenuate any ac noise which may be present. However, this system can also be used as a universal differential gain block for signals up to 10 kHz, so the differential cutoff frequency is set to 10 kHz. TIDU033-September 2013-Revised September 2013 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution Copyright © 2013, Texas Instruments Incorporated 5 www.ti.com The cutoff frequency of the filter is defined by the following equations: (9) (10) (11) The desired cutoff frequencies for this filter are f C_DIFF = 10 kHz and fC_CM = 200 kHz. A simple way to calculate the required passive component values is to pick a common value for C4 and solve for R11. By rearranging the terms of the equations above, the required value of R 11 for each filter is defined as: (12) (13) Substituting C4 = 10 nF, fC_DIFF = 10 kHz and fC_CM = 200 kHz in the equations above yields the following values for R1: Given these ideal values, 75 Ω was selected as the nearest 1% standard value which satisfies both requirements. Using the final circuit values of R11 = 75 Ω and C4 = 10 nF, the final cutoff frequencies of the filter are: The filter at the output of the PGA281 follows the same topology, however the desired cutoff frequencies are one decade higher (fC_DIFF = 100 kHz and fC_CM = 2 MHz). The same design equations as above can be used to calculate the required resistor and capacitor values, yielding the following result: 6 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution TIDU033-September 2013-Revised September 2013 Copyright © 2013, Texas Instruments Incorporated www.ti.com 3 Component Selection 3.1 Instrumentation Amplifier Selection In high-side current sensing applications, the sense resistance is placed between the supply voltage V BUS and the load. In this case the instrumentation amplifier’s input common-mode voltage range must include VBUS, which can often be greater than the supply voltage range of most single-supply amplifiers. The PGA281 was chosen for this design since it features a wide input voltage range, made possible by its dedicated high-voltage input stage power supplies. In order to accurately measure a wide range of currents, multiple instrumentation amplifier gain settings must be used. The PGA281 is ideal for this application as it can be easily programmed to multiple gains, from 0.125 V/V to 176 V/V, by applying a logic-level voltage to its gain set pins. 3.2 Power Management Selection The PGA281 requires three separate power connections: a high-voltage, split-supply for the input stage and single supplies for the output stage and digital logic. In order to simplify the power connections required on the PCB for this design, a linear regulator was used to generate a +5 V rail from the +15 V rail of the split supply. The TPS7A4101 was chosen for this design to meet the input voltage range and output current requirements of this design while allowing for adjustable output voltage and consuming low quiescent current. 3.3 Passive Component Selection The passive component with the greatest impact on this design is the shunt resistor RSH. This component must accurately convert load current to a differential voltage, while also potentially dissipating a significant amount of power. Selecting an ideal shunt resistor can be difficult, since resistors with very low tolerances typically can’t dissipate significant power, and resistors with high power handling typically aren’t available in low tolerances. A compromise was found for this design by selecting a 6.8 Ω resistor with ±1% tolerance in a large 2512 package that can dissipate up to 16 W of power - well within the maximum requirement of 1 W. The resistors and capacitors in the PGA281 input and output filter network also have an impact on the design as they are located within the signal path. However, the cutoff frequencies of the filters do not need to be extremely accurate, so resistor tolerance of ±1% and capacitor tolerance of ±10% are chosen. Select tighter tolerances if required by your application. Other passive components in this design may be selected for ±1% or greater tolerance as they will not directly affect the transfer function of the system. Ensure that all capacitors selected have sufficient voltage ratings. 3.4 Protection Component Selection Several additional circuit components provide protection for the system against ESD (electrostatic discharge), EFT (electrical fast transients), and surge (simulates a lightning strike). This protection is provided by a Schottky diode and two TVS (transient voltage suppressor) diodes. The BAT54-V-GS08 Schottky diode ensures that no current flows through the split supply when the power terminals are connected in reverse polarity. This diode protects against reverse voltages up to 30V, and the small SOT-23 package takes up a minimal amount of PCB area. Since the split supplies to the PGA281 can reach up to ±18 V, the TVS diodes should have a breakdown voltage slightly higher than 18V. The diodes must also be bidirectional and should have a very fast response time in order to provide sufficient protection against fast transients. Based on these requirements the SMBJ20CA was chosen to provide up to 600 W of protection. TIDU033-September 2013-Revised September 2013 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution Copyright © 2013, Texas Instruments Incorporated 7 www.ti.com Simulation schematic shown in Figure 5 includes the circuit values obtained in the design process. VSOP R13 75 Iload 100m VSON C10 10n V VOUT VON R14 75 EF 1 - High 0 - Low VOCM SW_G0 SW_G1 SW_G2 SW_G3 SW_G4 + C11 10n VS3 5 R15 20k AM2 + AM1 + + VS2 15 + VSOP R16 20k VSOP VOP VOCM VON VSN VSP C5 1n C3 10n INN VSN R12 75 VOP G0 G1 G2 G3 G4 EF Vin C6 100n V VSOP VOCM C8 10n Rsh 6.8 DVDD INP + Vbus 10 U1 PGA281 VSP C7 10n R11 75 C4 10n C2 10n VSP C9 1n TM The TINA-TI C1 10n 4 Vhi 5 VS5 15 VSN TM Figure 5: TINA-TI Schematic Note that a series of single pole, double throw (SPDT) switches are used to control the gain of the PGA281. Refer to the gain control table in the PGA281 data sheet for all the possible gain settings. Also note that the power management circuitry in the real system is replaced with discrete power supplies in the simulation schematic. 8 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution TIDU033-September 2013-Revised September 2013 Copyright © 2013, Texas Instruments Incorporated www.ti.com 4.1 Transfer Function The result of the dc transfer function is shown in Figure 6. 5.0 4.5 4.0 Output Voltage (V) 3.5 3.0 2.5 2.0 G = 5.5 V/V G = 176 V/V 1.5 1.0 0.5 0.0 0.00 0.01 0.02 0.03 0.04 0.05 0.06 Load Current (A) 0.07 0.08 0.09 0.10 Figure 6: Simulated Transfer Function TIDU033-September 2013-Revised September 2013 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution Copyright © 2013, Texas Instruments Incorporated 9 www.ti.com 4.2 Frequency Response The circuit shown in Figure 7 was utilized to perform an ac analysis for each gain of the circuit. The simulation results are shown in Figure 8. U1 PGA281 VSP DVDD VON VSON C10 10n 0 - Low SW_G0 VOCM SW_G1 SW_G2 SW_G3 SW_G4 + C11 10n VS3 5 + VOUT 1 - High R15 20k AM2 + AM1 + VS2 15 V VSOP R16 20k VSOP + EF VSN VSP R14 75 C7 10n INN VOCM C9 1n VOCM VSN R12 75 VOP G0 G1 G2 G3 G4 EF R13 75 C6 100n VG1 C8 10n + INP VSOP C5 1n C3 10n C2 10n C4 10n R11 75 C1 10n VSOP VSP Vhi 5 VS5 15 VSN Figure 7: AC Analysis Schematic 100.00 Gain (dB) 50.00 0.00 G = 44.91 dB = 176 V/V BW = 9.90 kHz G = 14.81 dB = 5.5 V/V BW = 10.00 kHz -50.00 -100.00 1 10 100 1k Frequency (Hz) 10k 100k 1M Figure 8: Simulated AC Analysis 10 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution TIDU033-September 2013-Revised September 2013 Copyright © 2013, Texas Instruments Incorporated www.ti.com The dc gains of the simulation were found to be 44.91 dB and 14.81 dB, which equate to 176.0 V/V and 5.5 V/V, respectively. The -3dB bandwidth of each gain configuration was 9.90 kHz and 10.00 kHz, respectively. 4.3 Full Scale Error The data from Figure 6 was exported to a spreadsheet in order to calculate the error as a percent of the full-scale range (%FSRerror). Equation 14 was used to calculate %FSRerror. (14) In order to accurately calculate the error, the gain was switched from 176 V/V to 5.5 V/V once the load current reached 4 mA. Figure 9 shows %FSRerror as a function of load current. 0.10 0.08 0.06 FSRerror (%) 0.04 0.02 0.00 -0.02 -0.04 -0.06 -0.08 -0.10 0 0.01 0.02 0.03 0.04 0.05 0.06 Load Current (A) 0.07 0.08 0.09 0.1 Figure 9: Simulated Full Scale Error The maximum simulated %FSRerror was found to be 0.002%, which meets our design goal of 0.05%. TIDU033-September 2013-Revised September 2013 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution Copyright © 2013, Texas Instruments Incorporated 11 www.ti.com 4.4 Relative Error The error relative to the load current (relative error) was also calculated for each load current decade. Equation 15 shows how to calculate relative error. (15) The maximum offset voltage of the PGA281 is calculated using Equation 16. (16) The maximum relative error for each decade of load current occurs at the minimum load current for each range (10 µA, 100 µA, 1 mA, 10 mA, and 100 mA). For example, the maximum relative error for a load current of 10 µA is calculated in Equation 17. (17) The maximum relative error for the other load currents can be calculated in a similar manner. Table 2 summarizes the results of the calculations. Equation 18 calculates the relative error using the simulation data. The output voltage was referred to the input by dividing by the ideal gain (176 V/V or 5.5 V/V). (18) 4.5 Simulated Results Summary Table 2 summarizes the simulated performance of the design. Table 2. Comparison of Design Goals and Simulated Performance 12 Goal Simulated Error (%FSRerror) 0.05% 0.002% Relative Error (ILOAD = 10 µA, G = 176) 35.0% 30.47% Relative Error (ILOAD = 100 µA, G = 176) 5% 3.13% Relative Error (ILOAD = 1 mA, G = 176) 1% 0.31% Relative Error (ILOAD = 10 mA, G = 5.5) 0.1% 0.09% Relative Error (ILOAD = 100 mA, G = 5.5) 0.01% 0.01% 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution TIDU033-September 2013-Revised September 2013 Copyright © 2013, Texas Instruments Incorporated www.ti.com 5 PCB Design The PCB schematic and bill of materials can be found in the Appendix. 5.1 PCB Layout The PCB used in this design is a 3” by 3” square. This generous size allows for efficient routing of critical components and the use of larger BNC and banana plug connectors. The high-level approach to this layout was to place the analog signal path components on the top layer, with input connections on the left and output connections on the right, and to place the power management components on the bottom layer. The load current source is connected at J2 and J4. Large copper areas are used to connect the load current to RSH, ensuring that parasitic trace resistance is minimized. Narrow copper traces running under RSH connect the induced current sense voltage to the PGA281 input filter while minimizing load current leakage. All passive components in the analog signal path are placed and routed very tightly in order to minimize parasitics, and all decoupling capacitors are located very close to their associated power pins. Solid copper areas on the bottom layer provide low-impedance paths for the various power supplies. Solid copper planes on both layers provide an excellent low-impedance path for return currents to ground. Connections to the split power supply are made at J7, J8, and J9. Connections to the differential output voltage are made at J3 and J5. If a voltage input is used, connections to the input voltage source are made at J1 and J6. The PCB layout for both layers is shown in Figure 10. Figure 10: PCB Layout TIDU033-September 2013-Revised September 2013 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution Copyright © 2013, Texas Instruments Incorporated 13 www.ti.com 6 Verification & Measured Performance 6.1 Bench Test Hardware Setup The circuit defined by this reference design is intended to represent a functional block which, in a real application, would only be part of a greater complete system. Even so, the convenient input and output connectors on the PCB allow the circuit to be easily tested on a bench using standard lab equipment. The test setup used consists of the components listed below. Figure 11 shows the full bench test setup. 1. Precision Current Source/Meter: Provides the load current to the system and sets the load compliance voltage. 2. Digital Multimeter, 6 ½ Digits: Measures the differential output voltage of the system. 3. Triple Output Power Supply: Provides ±15 V power supply rails to the system. 4. Digital Multimeter, 8 ½ Digits: Measures the load current input to the system. Figure 11: Bench Test Setup Once the test hardware was connected, a LabVIEW program was used to communicate with the equipment over a GPIB interface and automate the data collection process. This allowed quick and thorough characterization of the system’s performance by automatically sweeping the load current and logging the load current and voltage output. This process was repeated for each gain setting of the PGA281. 14 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution TIDU033-September 2013-Revised September 2013 Copyright © 2013, Texas Instruments Incorporated www.ti.com 6.2 Full Scale Error The logged test data was exported to a spreadsheet in order to calculate the error as a percent of the fullscale range (%FSRerror). Equation 14 was used to calculate %FSRerror. In order to accurately calculate the error, the gain was switched from 176 V/V to 5.5 V/V once the load current reached 4 mA. Figure 12 shows %FSRerror as a function of load current. 0.10 0.08 0.06 FSRerror (%) 0.04 0.02 0.00 -0.02 -0.04 -0.06 -0.08 -0.10 0 0.01 0.02 0.03 0.04 0.05 0.06 Load Current (A) 0.07 0.08 0.09 0.1 Figure 12: Measured Full Scale Error The maximum %FSRerror was found to be 0.0481%, which meets our design goal of 0.05%. TIDU033-September 2013-Revised September 2013 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution Copyright © 2013, Texas Instruments Incorporated 15 www.ti.com 6.3 Relative Error The error relative to the load current (relative error) was also calculated for each load current decade. Equation 18 calculates the measured relative error using the logged test data. The output voltage was referred to the input by dividing by the ideal gain (176 V/V or 5.5 V/V). Table 3 compares the measured and ideal output voltages at the minimum load current of each decade. Table 3. Measured Output Voltage for Various Load Currents 6.4 Load Current Specified Gain (V/V) Ideal Output Voltage Measured Output Voltage 10 µA 176 11.968 mV 11.323 mV 100 µA 176 119.68 mV 119.107 mV 1 mA 176 1.1968 V 1.1974 V 10 mA 5.5 374.00 mV 374.15 mV 100 mA 5.5 3.74 V 3.7419 V Measured Results Summary Table 4 summarizes the measured performance of the design. Table 4. Comparison of Design Goals and Measured Performance 16 Load Current Goal Measured Error (%FSRerror) 0.05% 0.0481% Relative Error (ILOAD = 10 µA, G = 176) 35.0% 33.16% Relative Error (ILOAD = 100 µA, G = 176) 5% 3.15% Relative Error (ILOAD = 1 mA, G = 176) 1% 0.31% Relative Error (ILOAD = 10 mA, G = 5.5) 0.1% 0.09% Relative Error (ILOAD = 100 mA, G = 5.5) 0.01% 0.01% 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution TIDU033-September 2013-Revised September 2013 Copyright © 2013, Texas Instruments Incorporated www.ti.com 7 Modifications The components selected for this design were based on the design goals outlined at the beginning of the design process. The type of resistor selected for RSH was the best overall choice for the 10µA to 100mA current range and the gain settings offered by the PGA281. However, applications which measure lower currents should consider a resistor with lower tolerance in a smaller package. Conversely, applications dealing with higher currents could benefit from specialized current sense resistors in a larger package (such as D2PAK) or a parallel combination of resistors. Some applications may require or benefit from multiplexing of multiple inputs or more robust diagnostics. The PGA280 is a programmable-gain instrumentation amplifier with the same analog core as the PGA281 TM but with these additional features included, as well as digital control via SPI and seven GPIO pins. Table 5 compares the PGA280 to the PGA281 as a potential PGA for this design. Table 5. Brief Comparison of PGA281 and PGA280 Instrumentation Amplifier Error Diagnostics Number of Differential Inputs Gain Control Approx. Price (US$) PGA281 Error Flag pin, no advanced error reporting 1 Dedicated gain set pins 2.55 / 1ku PGA280 Error Flag pin, detailed error reporting via registers 2, more possible with GPIO mux control SPI 2.90 / 1ku Other programmable-gain instrumentation amplifiers with a wide power supply and analog input range and low offset voltage could also be used in a high-side current sensing application. The PGA204 and PGA205 are suitable choices for this system, with the differences being different gain options, higher quiescent current, lower noise, and higher cost. Table 6 summarizes other potential PGAs for this design as compared to the PGA281. Table 6. Brief Comparison of Programmable-gain Instrumentation Amplifiers Instrumentation Amplifier Gain Options Offset Voltage (µV, RTI) Voltage Noise (RTI, 1 kHz) Quiescent Current Approx. Price (US$) PGA281 1/8 to 176 ±5 + 45/G 22 nV/√Hz 3 mA 2.55 / 1ku PGA204 1, 10, 100, 1000 ±10 + 20/G 13 nV/√Hz 5.2 mA 8.35 / 1ku PGA205 1, 2, 4, 8 ±10 + 20/G 13 nV/√Hz 5.2 mA 7.25 / 1ku TIDU033-September 2013-Revised September 2013 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution Copyright © 2013, Texas Instruments Incorporated 17 www.ti.com 8 About the Author Ian Williams (ian@ti.com) is an applications engineer in the Precision Analog – Linear team at Texas Instruments where he supports industrial products and applications. Ian graduated from the University of Texas, Dallas, where he earned a Bachelor of Science in Electrical Engineering with a concentration in Microelectronics. 9 Acknowledgements & References 9.1 Acknowledgements The author wishes to acknowledge Peter Semig for his guidance in the completion of this design, and Michael Mock for providing LabVIEW software which automated the data collection process. 9.2 18 References 1. P. Semig. (2013, June 18). 10uA-10mA, Single-Supply, Low-Side, Current Sensing Solution. Available: http://www.ti.com/lit/ug/slau504/slau504.pdf 2. P. Semig and C. Wells. (2012, February 8). A Current Sensing Tutorial Parts I-IV. Available: http://www.eetimes.com/design/industrial-control 3. PGA281EVM User’s Guide (SBOU130). Available: http://www.ti.com/lit/ug/sbou130/sbou130.pdf 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution TIDU033-September 2013-Revised September 2013 Copyright © 2013, Texas Instruments Incorporated www.ti.com Appendix A. A.1 Electrical Schematic Figure A-1: Electrical Schematic A.2 Bill of Materials Figure A-2: Bill of Materials TIDU033-September 2013-Revised September 2013 10µA-100mA, 0.05% Error, High-Side Current Sensing Solution Copyright © 2013, Texas Instruments Incorporated 19 IMPORTANT NOTICE FOR TI REFERENCE DESIGNS Texas Instruments Incorporated ("TI") reference designs are solely intended to assist designers (“Buyers”) who are developing systems that incorporate TI semiconductor products (also referred to herein as “components”). Buyer understands and agrees that Buyer remains responsible for using its independent analysis, evaluation and judgment in designing Buyer’s systems and products. TI reference designs have been created using standard laboratory conditions and engineering practices. TI has not conducted any testing other than that specifically described in the published documentation for a particular reference design. 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With such components, TI’s goal is to help enable customers to design and create their own end-product solutions that meet applicable functional safety standards and requirements. Nonetheless, such components are subject to these terms. No TI components are authorized for use in FDA Class III (or similar life-critical medical equipment) unless authorized officers of the parties have executed an agreement specifically governing such use. Only those TI components that TI has specifically designated as military grade or “enhanced plastic” are designed and intended for use in military/aerospace applications or environments. Buyer acknowledges and agrees that any military or aerospace use of TI components that have not been so designated is solely at Buyer's risk, and Buyer is solely responsible for compliance with all legal and regulatory requirements in connection with such use. TI has specifically designated certain components as meeting ISO/TS16949 requirements, mainly for automotive use. In any case of use of non-designated products, TI will not be responsible for any failure to meet ISO/TS16949. Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265 Copyright © 2013, Texas Instruments Incorporated