HT1541ARTZ

HT1541A 1.3MHz Boost Converter DESCRIPTION FEATURES The HT1541 is a current mode step up converter intended for small, low power applications. The HT1541 switches at 1.3MHz and allows the use of tiny, low cost capacitors and inductors 2mm or less in height. Internal soft start results in small inrush current and extends battery life. The HT1541 operates from an input voltage as low as 2.5V and can generate 12V at up to 300mA from a 5V supply.       The HT1541 includes under-voltage lockout, current limiting, and thermal overload protection to prevent damage in the event of an output overload. The HT1541 is available in a small 5-pin TSOT23 package or QFN-8 (2mmX2mm) package. TYPICAL APPLICATION     On Board Power MOSFET Uses Tiny Capacitors and Inductors 1.3MHz Fixed Switching Frequency Internally Compensated Internal Soft-Start Operates with Input Voltage as Low as 2.5V and Output Voltage as High as 22V 12V at 300mA from 5V Input UVLO, Thermal Shutdown Internal Current Limit Available in a TSOT23-5 Package or QFN-8 (2mmX2mm) Package APPLICATIONS       Camera Phone Flash Handheld Computers and PDAs Digital Still and Video Cameras External Modems Small LCD Displays White LED Driver Efficiency vs Load Current 100 95 VIN = 5V HT1541A EFFICIENCY (%) 90 85 80 VIN = 3.3V 75 70 VIN = 4.2V 65 60 55 50 VOUT = 12V 0 75 150 225 300 375 450 LOAD CURRENT (mA) Rev. 01 HT1541A PACKAGE REFERENCE TOP VIEW SW 1 GND 2 FB 3 5 IN 4 EN TSOT23-5 QFN-8 (2mmX2mm) (4) ABSOLUTE MAXIMUM RATINGS (1) Thermal Resistance SW Pin ........................................... –0.3V to 25V All Other Pins ................................ –0.3V to 6.5V Junction Temperature ...............................150°C (2) Continuous Power Dissipation (TA = +25°C) TSOT23-5 ................................................ 0.47W QFN-8 (2mmx2mm) ................................. 1.56W Lead Temperature ....................................260°C Storage Temperature .............. –65°C to +150°C TSOT25 ................................. 220 .... 110.. C/W QFN-8 (2mmX2mm) ............... 80 ...... 16... C/W Recommended Operating Conditions (3) Supply Voltage VIN ............................. 2.5V to 6V Output Voltage VOUT ............................ 3V to 22V Operating Temperature............. –40°C to +85°C Maximum Junction Temp. (TJ) ............. +125°C θJA θJC Notes: 1) Exceeding these ratings may damage the device. 2) The maximum allowable power dissipation is a function of the maximum junction temperature TJ(MAX), the junction-toambient thermal resistance θJA, and the ambient temperature TA. The maximum allowable continuous power dissipation at any ambient temperature is calculated by PD(MAX)=(TJ(MAX)TA)/θJA. Exceeding the maximum allowable power dissipation will cause excessive die temperature, and the regulator will go into thermal shutdown. Internal thermal shutdown circuitry protects the device from permanent damage. 3) The device is not guaranteed to function outside of its operating range. 4) Measured on JESD51-7 4-layer board. Rev. 01 HT1541A ELECTRICAL CHARACTERISTICS VIN = VEN = 5V, TA = +25C unless otherwise specified. Parameters Symbol Operating Input Voltage Condition VIN Min Typ 2.5 Undervoltage Lockout 2.25 Undervoltage Lockout Hysteresis 92 Max Units 6 V 2.45 V mV Supply Current (Shutdown) VEN = 0V 0.1 1 µA Supply Current (Quiescent) VFB = 1.3V 635 850 µA 1.0 1.3 1.6 MHz Switching Frequency fSW Maximum Duty Cycle VFB = 0V 80 85 EN Threshold VEN Rising 1.0 1.3 EN Threshold VEN Rising, VIN = 2.5V EN Hysteresis EN Input Bias Current FB Voltage FB Input Bias Current (5) SW On-Resistance VFB = 1.25V RDS (ON) SW Current Limit (5) SW Leakage Thermal Shutdown V V 100 mV 1 µA 1.29 V 1.21 1.25 –100 –30 nA 0.65 Ω 1.9 A VSW = 15V (5) 1.6 1.1 VEN = 0V, 6V VFB % 1 160 µA C Note: 5) Guaranteed by design. Rev. 01 HT1541A TYPICAL PERFORMANCE CHARACTERISTICS Frequency vs Temperature 1.252 1.40 1.250 1.35 FREQUENCY (MHz) FEEDBACK VOLTAGE (V) Feedback Voltage vs Temperature 1.248 1.246 1.244 1.242 -50 0 50 100 TEMPERATURE (°C) 1.30 1.25 1.20 1.15 -50 150 640 84.0 635 83.9 630 83.8 625 83.7 620 83.6 615 610 -50 150 0.80 1.6 0.75 1.5 0.70 0.65 0.60 0.55 0.50 0 50 100 TEMPERATURE (°C) 150 Current Limit vs Duty Cycle RDS (ON) vs Input Voltage CURRENT LIMIT (A) MAXIMUM DUTY CYCLE (%) 84.1 0 50 100 TEMPERATURE (°C) 150 Supply Current vs Temperature Maximum Duty Cycle vs Temperature 83.5 -50 0 50 100 TEMPERATURE (°C) 1.4 1.3 1.2 1.1 2 3 4 5 INPUT VOLTAGE (V) 6 1.0 30 40 50 60 70 DUTY CYCLE (%) 80 Rev. 01 HT1541A PIN FUNCTIONS Pin # TSSOT QFN Name Function Power Switch Output. SW is the drain of the internal MOSFET switch. Connect the power inductor and output rectifier to SW. SW can swing between GND and 22V. 1 8 SW 2 1,4 GND 3 5 FB Feedback Input. FB voltage is 1.25V. Connect a resistor divider to FB. 4 3 EN Regulator On/Off Control Input. A high input at EN turns on the converter, and a low input turns it off. When not used, connect EN to the input source for automatic startup. The EN pin cannot be left floating. 5 2 IN Input Supply Pin. Must be locally bypassed. N.A 6,7 N/C Ground. Do not connect. Reserved for factory use. OPERATION The HT1541 uses a fixed frequency, peak current mode boost regulator architecture to regulate voltage at the feedback pin. The operation of the HT1541 can be understood by referring to the block diagram of Figure 1. At the start of each oscillator cycle the MOSFET is turned on through the control circuitry. To prevent sub-harmonic oscillations at duty cycles greater than 50 percent, a stabilizing ramp is added to the output of the current sense amplifier and the result is fed into the negative input of the PWM comparator. When this voltage equals the output voltage of the error amplifier the power MOSFET is turned off. The voltage at the output of the error amplifier is an amplified version of the difference between the 1.25V bandgap reference voltage and the feedback voltage. In this way the peak current level keeps the output in regulation. If the feedback voltage starts to drop, the output of the error amplifier increases. This results in more current to flow through the power MOSFET, thus increasing the power delivered to the output. The HT1541 has internal soft start to limit the amount of input current at startup and to also limit the amount of overshoot on the output. The current limit is increased by a fourth every 40s giving a total soft start time of 120s. CC RC SW 1 FB 3 + + - 1.25V ERROR AMPLIFIER CONTROL LOGIC M1 PWM COMPARATOR + + - 1.3MHz OSC CURRENT SENSE AMPLIFIER 2 GND Figure 1—Functional Block Diagram Rev. 01 HT1541A APPLICATIONS INFORMATION COMPONENT SELECTION Setting the Output Voltage Set the output voltage by selecting the resistive voltage divider ratio. Use 11.8kΩ for the lowside resistor R2 of the voltage divider. Determine the high-side resistor R1 by the equation: R1  R2VOUT - VFB  VFB where VOUT is the output voltage. For R2 = 11.8kΩ and VFB = 1.25V, then R1 (kΩ) = 9.44kΩ (VOUT – 1.25V). Selecting the Input Capacitor An input capacitor is required to supply the AC ripple current to the inductor, while limiting noise at the input source. This capacitor must have low ESR, so ceramic is the best choice. Use an input capacitor value of 4.7μF or greater. This capacitor must be placed physically close to the IN pin. Since it reduces the voltage ripple seen at IN, it also reduces the amount of EMI passed back along that line to the other circuitry. Selecting the Output Capacitor A single 4.7F to 10F ceramic capacitor usually provides sufficient output capacitance for most applications. If larger amounts of capacitance is desired for improved line support and transient response, tantalum capacitors can be used in parallel with the ceramic. The impedance of the ceramic capacitor at the switching frequency is dominated by the capacitance, and so the output voltage ripple is mostly independent of the ESR. The output voltage ripple VRIPPLE is calculated as: VRIPPLE  input voltage. Choose an inductor that does not saturate at the SW current limit. A good rule for determining the inductance is to allow the peakto-peak ripple current to be approximately 30%50% of the maximum input current. Make sure that the peak inductor current is below 75% of the typical current limit at the duty cycle used to prevent loss of regulation due to the current limit variation. Calculate the required inductance value L using the equations: L VIN (VOUT - VIN ) VOUT  fSW  I IIN(MAX )  VOUT  ILOAD (MAX ) VIN   I  30%  50%IIN(MAX ) Where ILOAD(MAX) is the maximum load current, ∆I is the peak-to-peak inductor ripple current, and η is efficiency. For the MP1541, 4.7µH is recommended for input voltages less than 3.3V and 10µH for inputs greater than 3.3V. Selecting the Diode The output rectifier diode supplies current to the inductor when the internal MOSFET is off. To reduce losses due to diode forward voltage and recovery time, use a Schottky diode. Choose a diode whose maximum reverse voltage rating is greater than the maximum output voltage. For output voltage less than 20V, it is recommended to choose the MBR0520 for most applications. This diode is used for load currents less than 500mA. If the average current is more than 500mA the Microsemi UPS5817 is a good choice. ILOAD VO UT  VIN  VO UT  C2  f SW Where VIN is the input voltage, ILOAD is the load current, C2 is the capacitance of the output capacitor, and fSW is the 1.3MHz switching frequency. Selecting the Inductor The inductor is required to force the output voltage higher while being driven by the lower Rev. 01 HT1541A Compensation The HT1541 uses an amplifier to compensate the feedback loop rather than a traditional transconductance amplifier like most current mode regulators. Frequency compensation is provided by an internal resistor and capacitor along with an external resistor. The system uses two poles and one zero to stabilize the control loop. The poles are fP1 set by the output capacitor and load resistance, and fP2 set by the internal compensation capacitor Cc, the gain of the error amplifier and the resistance seen looking out at the feedback node REQ. The zero fZ1 is set internally around 20kHz. These are determined by the equations: fP1  fP 2  1   C2  R LOAD 1  2    7.9  10 9  R EQ Where RLOAD is the load resistance and REQ is: Where R1, R2, and R3 are seen in Figure 2. The DC loop gain is: VIN  R LOAD  VFB VOUT 2 There is also a right-half-plane zero (fRHPZ) that exists in all continuous mode (inductor current does not drop to zero on each cycle) step up converters. The frequency of the right half plane zero is: 2 fRHPZ  VIN  R LOAD 2    L  VOUT 1     1  2    C3   R4  1 1 1       R1 R2 R3   f Z2  (R1 R2)  R3  (R1  R2) A VDC  500  For the HT1541 it is recommended that a 47kΩ to 100kΩ resistor be placed in series with the FB pin and the resistor divider as seen in Figure 2. For most applications this is all that is needed for stable operation. If greater phase margin is needed a series resistor and capacitor can be placed in parallel with the high-side resistor R1 as seen in Figure 2. The pole and zero set by the lead-lag compensation network are: fP 3  f Z1  20kHz R EQ To stabilize the regulation control loop, the crossover frequency (the frequency where the loop gain drops to 0dB or a gain of 1, indicated as fC) should be at least one decade below the right-half-plane zero and should be at most 75kHz. fRHPZ is at its lowest frequency at maximum output load current (RLOAD is at a minimum) and minimum input voltage. 1 2    C3  R1  R4  LAYOUT CONSIDERATIONS High frequency switching regulators require very careful layout for stable operation and low noise. All components must be placed as close to the IC as possible. Keep the path between L1, D1, and C2 extremely short for minimal noise and ringing. C1 must be placed close to the IN pin for best decoupling. All feedback components must be kept close to the FB pin to prevent noise injection on the FB pin trace. The ground return of C1 and C2 should be tied close to the GND pin. See the HT1541 demo board layout for reference. 2 Rev. 01 HT1541A TYPICAL APPLICATION CIRCUITS HT1541A Figure 2—VIN = 5V, VOUT = 12V, IOUT = 300mA Boost Circuit HT1541A Figure 3—Typical Application Circuit for Driving Flashlight LEDs (20mA Torch Current, 100mA Flash Current) Rev. 01 HT1541A PACKAGE INFORMATION SOT23-5(3mm*1.6mm) Rev. 01 HT1541A PACKAGE INFORMATION QFN-8 (2mmX2mm) PIN 1 ID MARKING 1.90 2.10 0.25 0.45 0.18 0.30 1.90 2.10 PIN 1 ID INDEX AREA PIN 1 ID SEE DETAIL A 0.45 0.65 8 1 1.05 1.25 0.50 BSC 5 TOP VIEW 4 BOTTOM VIEW 0.80 1.00 PIN 1 ID OPTION A 0.30x45º TYP. PIN 1 ID OPTION B R0.20 TYP. 0.20 REF 0.00 0.05 SIDE VIEW DETAIL A NOTE: 1.90 0.70 0.60 1) ALL DIMENSIONS ARE IN MILLIMETERS. 2) EXPOSED PADDLE SIZE DOES NOT INCLUDE MOLD FLASH. 3) LEAD COPLANARITY SHALL BE 0.10 MILLIMETER MAX. 4) DRAWING CONFORMS TO JEDEC MO-229, VARIATION VCCD-3. 5) DRAWING IS NOT TO SCALE. 0.25 1.20 0.50 RECOMMENDED LAND PATTERN Rev. 01