ADS7813PB

ADS7813 ADS 781 3 ADS 781 3 SBAS043C – MARCH 1997 – REVISED SEPTEMBER 2005 Low-Power, Serial 16-Bit Sampling ANALOG-TO-DIGITAL CONVERTER FEATURES ● ● ● ● ● ● DESCRIPTION 20µs max CONVERSION TIME SINGLE +5V SUPPLY OPERATION PIN-COMPATIBLE WITH 12-BIT ADS7812 EASY-TO-USE SERIAL INTERFACE 0.3" DIP-16 AND SO-16 ±2.0LSB max INL ● 87dB min SINAD ● USES INTERNAL OR EXTERNAL REFERENCE ● MULTIPLE INPUT RANGES ● 35mW max POWER DISSIPATION ● NO MISSING CODES ● 50µW POWER DOWN MODE The ADS7813 is a low-power, single +5V supply, 16-bit sampling analog-to-digital (A/D) converter. It contains a complete 16-bit capacitor-based SAR A/D with a sample/hold, clock, reference, and serial data interface. The converter can be configured for a variety of input ranges including ±10V, ±5V, 0V to 10V, and 0.5V to 4.5V. A high impedance 0.3V to 2.8V input range is also available (input impedance > 10MΩ). For most input ranges, the input voltage can swing to +16.5V or –16.5V without damage to the converter. A flexible SPI-compatible serial interface allows data to be synchronized to an internal or external clock. The ADS7813 is specified at a 40kHz sampling rate over the –40°C to +85°C temperature range. It is available in a 0.3" DIP-16 or an SO-16 package. APPLICATIONS ● ● ● ● ● ● ● MEDICAL INSTRUMENTATION DATA ACQUISITION SYSTEMS ROBOTICS INDUSTRIAL CONTROL TEST EQUIPMENT DIGITAL SIGNAL PROCESSING DSP SERVO CONTROL BUSY PWRD CONV CS Successive Approximation Register and Control Logic 40kΩ(1) Clock CDAC R1IN 8kΩ(1) EXT/INT R2IN Serial 20kΩ(1) Data Comparator R3IN BUF CAP DATACLK Out DATA Buffer 4kΩ(1) Internal +2.5V Ref NOTE: (1) Actual value may vary ±30%. REF Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. Copyright © 1997-2005, Texas Instruments Incorporated PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. www.ti.com ABSOLUTE MAXIMUM RATINGS(1) ELECTROSTATIC DISCHARGE SENSITIVITY Analog Inputs: R1IN ......................................................................... ±16.5V R2IN ................................................ GND – 0.3V to +16.5V R3IN ......................................................................... ±16.5V REF ............................................ GND – 0.3V to VS + 0.3V CAP ............................................... Indefinite Short to GND Momentary Short to VS VS ........................................................................................................... 7V Digital Inputs ...................................................... GND – 0.3V to VS + 0.3V Maximum Junction Temperature ................................................... +165°C Internal Power Dissipation ............................................................. 825mW Lead Temperature (soldering, 10s) ............................................... +300°C This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. NOTE: (1) Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. PACKAGE/ORDERING INFORMATION(1) PRODUCT MINIMUM MAXIMUM SPECIFIED SIGNAL-TOINTEGRAL NO MISSING (NOISE + LINEARITY CODE LEVEL DISTORTION) ERROR (LSB) (LSB) RATIO (DB) PACKAGE-LEAD PACKAGE DESIGNATOR SPECIFIED TEMPERATURE RANGE PACKAGE MARKING ORDERING NUMBER TRANSPORT MEDIA, QUANTITY ADS7813P ADS7813PB Tubes, 25 Tubes, 25 ADS7813P ADS7813PB ±3 ±2 15 16 85 87 Dip-16 N " " ADS7813U " ±3 " 15 " 85 " SO-16 " DW " –40°C to +85°C " ADS7813U ADS7813U/1K Tubes, 48 Tape and Reel, 1000 ADS7813UB " ±2 " 16 " 87 " SO-16 " DW " –40°C to +85°C ADS7813UB ADS7813UB " " ADS7813UB/1K Tubes, 48 Tape and Reel, 1000 –40°C to +85°C ADS7813P " ADS7813PB ADS7813U " NOTE: (1) For the most current package and ordering information, see the Package Option Addendum at the end of this data sheet, or see the TI web site at www.ti.com. ELECTRICAL CHARACTERISTICS At TA = –40°C to +85°C, fS = 40kHz, VS = +5V ±5%, using internal reference, unless otherwise specified. ADS7813P, U PARAMETER CONDITIONS MIN TYP RESOLUTION DC ACCURACY Integral Linearity Error Differential Linearity Error No Missing Codes Transition Noise(2) Full Scale Error(3) Full Scale Error Drift Full Scale Error(3) Full Scale Error Drift Bipolar Zero Error Bipolar Zero Error Drift Unipolar Zero Error Unipolar Zero Error Drift Recovery Time to Rated Accuracy from Power Down(4) Power Supply Sensitivity AC ACCURACY Spurious-Free Dynamic Range Total Harmonic Distortion Signal-to-(Noise+Distortion) Signal-to-Noise Useable Bandwidth(6) Full Power –3dB Bandwidth 2 MAX MIN TYP 16 ANALOG INPUT Voltage Range Impedance Capacitance THROUGHPUT SPEED Conversion Time Complete Cycle Throughput Rate ADS7813PB, UB 20 25 ±3 +3, –2 15 ✻ ✻ ±14 ±5 ±3 ±3 300 90 85 85 ✻ ✻ µs µs kHz ±2 +2, –1 LSB(1) LSB Bits LSB % ppm/°C % ppm/°C mV ppm/°C mV ppm/°C µs ✻ ±0.5 ±0.25 ✻ ±0.5 ±0.25 ✻ ±10 ✻ ✻ ±6 ✻ ✻ ✻ ±12 +4.75V < (VS = +5V) < +5.25 1kHz 1kHz 1kHz 1kHz pF 16 0.6 fIN = fIN = fIN = fIN = Bits ✻ 40 Ext. 2.5000V Ref Ext. 2.5000V Ref Bipolar Ranges Bipolar Ranges Unipolar Ranges Unipolar Ranges 1.0µF Capacitor to CAP UNITS ✻ ✻ ✻ ✻ See Table I See Table I 35 Acquire and Convert MAX 100 –98 89 89 130 600 ✻ 96 –90 87 87 102 –100 ✻ ✻ ✻ ✻ LSB –96 dB(5) dB dB dB kHz kHz ADS7813 www.ti.com SBAS043C ELECTRICAL CHARACTERISTICS (Cont.) At TA = –40°C to +85°C, fS = 40kHz, VS = +5V ±5%, using internal reference, unless otherwise specified. ADS7813P, U PARAMETER SAMPLING DYNAMICS Aperture Delay Aperture Jitter Transient Response Overvoltage Recovery(7) REFERENCE Internal Reference Voltage Internal Reference Source Current Internal Reference Drift External Reference Voltage Range External Reference Current Drain CAP Compensation Capacitors ESR(8) CONDITIONS Output Capacitance POWER SUPPLY VS Power Dissipation TEMPERATURE RANGE Specified Performance Derated Performance TYP MAX MIN 2.48 2.3 2.5 100 8 2.5 VREF = +2.5V –0.3 +2.0 TYP MAX ✻ ✻ ✻ ✻ 40 20 5 750 FS Step DIGITAL INPUTS Logic Levels VIL VIH(9) IIL IIH DIGITAL OUTPUTS Data Format Data Coding VOL VOH Leakage Current MIN ADS7813PB, UB 2.52 ✻ 2.7 100 3 ✻ +0.8 VS +0.3V ±10 ±10 ✻ ✻ ✻ ✻ ✻ ✻ UNITS ns ps µs ns ✻ ✻ ✻ ✻ V µA ppm/°C V µA Ω ✻ ✻ ✻ ✻ V V µA µA ✻ Serial Binary Two’s Complement ISINK = 1.6mA ISOURCE = 500µA High-Z State, VOUT = 0V to VS High-Z State +0.4 ±1 ✻ V V µA 15 15 pF ✻ ✻ V mW ✻ ✻ °C °C ✻ +4 +4.75 fS = 40kHz –40 –55 +5 +5.25 35 ✻ +85 +125 ✻ ✻ ✻ ✻ Same specification as grade to the left. NOTES: (1) LSB means Least Significant Bit. For the ±10V input range, one LSB is 305µV. (2) Typical rms noise at worst case transitions and temperatures. (3) Full scale error is the worst case of –Full Scale or +Full Scale untrimmed deviation from ideal first and last code transitions, divided by the transition voltage (not divided by the full-scale range) and includes the effect of offset error. (4) After the ADS7813 is initially powered on and fully settles, this is the time delay after it is brought out of Power Down Mode until all internal settling occurs and the analog input is acquired to rated accuracy, and normal conversions can begin again. (5) All specifications in dB are referred to a full-scale input. (6) Useable Bandwidth defined as Full-Scale input frequency at which Signal-to-(Noise+Distortion) degrades to 60dB, or 10 bits of accuracy. (7) Recovers to specified performance after 2 x FS input overvoltage. (8) ESR = total equivalent series resistance for the compensation capacitors. (9) The minimum VIH level for the DATACLK signal is 3V. ADS7813 SBAS043C www.ti.com 3 PIN CONFIGURATION PIN # NAME DESCRIPTION 1 R1IN Analog Input. See Tables I and IV. 2 GND Ground 3 R2IN Analog Input. See Tables I and IV. 4 R3IN Analog Input. See Tables I and IV. 5 BUF Reference Buffer Output. Connect to R1IN, R2IN, or R3IN, as needed. 6 CAP Reference Buffer Compensation Node. Decouple to ground with a 1µF tantalum capacitor in parallel with a 0.01µF ceramic capacitor. 7 REF Reference Input/Output. Outputs internal +2.5V reference via a series 4kΩ resistor. Decouple this voltage with a 1µF to 2.2µF tantalum capacitor to ground. If an external reference voltage is applied to this pin, it will override the internal reference. 8 GND 9 DATACLK Data Clock Pin. With EXT/INT LOW, this pin is an output and provides the synchronous clock for the serial data. The output is tri-stated when CS is HIGH. With EXT/INT HIGH, this pin is an input and the serial data clock must be provided externally. Ground 10 DATA Serial Data Output. The serial data is always the result of the last completed conversion and is synchronized to DATACLK. If DATACLK is from the internal clock (EXT/INT LOW), the serial data is valid on both the rising and falling edges of DATACLK. DATA is tri-stated when CS is HIGH. 11 EXT/INT External or Internal DATACLK Pin. Selects the source of the synchronous clock for serial data. If HIGH, the clock must be provided externally. If LOW, the clock is derived from the internal conversion clock. Note that the clock used to time the conversion is always internal regardless of the status of EXT/INT. 12 CONV Convert Input. A falling edge on this input puts the internal sample/hold into the hold state and starts a conversion regardless of the state of CS. If a conversion is already in progress, the falling edge is ignored. If EXT/INT is LOW, data from the previous conversion will be serially transmitted during the current conversion. 13 CS Chip Select. This input tri-states all outputs when HIGH and enables all outputs when LOW. This includes DATA, BUSY, and DATACLK (when EXT/INT is LOW). Note that a falling edge on CONV will initiate a conversion even when CS is HIGH. 14 BUSY Busy Output. When a conversion is started, BUSY goes LOW and remains LOW throughout the conversion. If EXT/INT is LOW, data is serially transmitted while BUSY is LOW. BUSY is tri-stated when CS is HIGH. 15 PWRD Power Down Input. When HIGH, the majority of the ADS7813 is placed in a low power mode and power consumption is significantly reduced. CONV must be taken LOW prior to PWRD going LOW in order to achieve the lowest power consumption. The time required for the ADS7813 to return to normal operation after power down depends on a number of factors. Consult the Power Down section for more information. 16 VS +5V Supply Input. For best performance, decouple to ground with a 0.1µF ceramic capacitor in parallel with a 10µF tantalum capacitor. PIN CONFIGURATION Top View DIP, SOIC R1IN 4 1 ANALOG INPUT RANGE (V) CONNECT R1IN TO CONNECT R2IN TO CONNECT R3IN TO INPUT IMPEDANCE (kΩ) ±10V VIN BUF GND 45.7 0.3125V to 2.8125V VIN VIN VIN > 10,000 26.7 ±5V GND BUF VIN 16 VS 0V to 10V BUF GND VIN 26.7 0V to 4V BUF VIN GND 21.3 ±3.33V VIN BUF VIN 21.3 0.5V to 4.5V GND VIN GND 21.3 GND 2 15 PWRD R2IN 3 14 BUSY R3IN 4 BUF 5 12 CONV CAP 6 11 EXT/INT REF 7 10 DATA GND 8 9 13 CS ADS7813 TABLE I. ADS7813 Input Ranges. DATACLK ADS7813 www.ti.com SBAS043C TYPICAL PERFORMANCE CURVES At TA = +25°C, fS = 40kHz, VS = +5V, ±10V input range, using internal reference, unless otherwise noted. FREQUENCY SPECTRUM (8192 Point FFT; fIN = 9.8kHz, 0dB) 0 0 –20 –20 –40 –40 Amplitude (dB) –60 –80 –100 –60 –80 –100 –120 –120 –140 –140 5 10 15 20 0 5 10 20 Frequency (kHz) SNR AND SINAD vs TEMPERATURE (fIN = 1kHz, 0dB) SFDR AND THD vs TEMPERATURE (fIN = 1kHz, 0dB) 92 106 –106 91 105 –105 SNR 90 89 SINAD 88 SFDR 104 –104 103 –103 THD 102 87 –102 101 86 –101 100 –50 –25 0 25 50 75 100 –100 –50 –25 0 Temperature (°C) 25 50 75 100 75 100 Temperature (°C) SIGNAL-TO-(NOISE + DISTORTION) vs INPUT FREQUENCY (fIN = 0dB) INTERNAL REFERENCE VOLTAGE vs TEMPERATURE 2.515 89 2.510 Internal Reference (V) 90 88 SINAD (dB) 15 Frequency (kHz) SFDR (dB) SNR and SINAD (dB) 0 THD (dB) Amplitude (dB) FREQUENCY SPECTRUM (8192 Point FFT; fIN = 980Hz, 0dB) 87 86 85 2.505 2.500 2.495 2.490 84 83 2.485 100 1k 10k 20k –50 Input Signal Frequency (Hz) 0 25 50 Temperature (°C) ADS7813 SBAS043C –25 www.ti.com 5 TYPICAL PERFORMANCE CURVES (Cont.) At TA = +25°C, fS = 40kHz, VS = +5V, ±10V input range, using internal reference, unless otherwise noted. ILE AND DLE AT +25°C 1 1 ILE (LSB) 2 0 –1 0 –1 –2 –2 2 2 1 1 DLE (LSB) DLE (LSB) ILE (LSB) ILE AND DLE AT –40°C 2 0 –1 –2 8000h C000h 0000h 4000h 0 –1 –2 8000h 7FFFh C000h 0000h 1 1 Linearity Degradation (LSB/LSB) ILE (LSB) 2 0 –1 –2 2 DLE (LSB) 7FFFh POWER SUPPLY RIPPLE SENSITIVITY ILE/DLE DEGRADATION PER LSB OF P-P RIPPLE ILE AND DLE AT +85°C 1 0 –1 –2 8000h 4000h Hex BTC Code Hex BTC Code 10–1 10–2 ILE 10–3 10–4 DLE 10–5 C000h 0000h 4000h 7FFFh 101 102 Hex BTC Code 103 104 105 106 107 Power Supply Ripple Frequency (Hz) WORST-CASE INL vs ESR OF REFERENCE BUFFER COMPENSATION CAPACITOR Worst-Case INL (LSB) 35 30 25 20 15 10 5 0 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6 6.5 7 7.5 8 8.5 9 Capacitor ESR (Ω) 6 ADS7813 www.ti.com SBAS043C BASIC OPERATION EXTERNAL DATACLK Figure 1b shows a basic circuit to operate the ADS7813 with a ±10V input range. To begin a conversion, a falling edge must be provided to the CONV input. BUSY will go LOW indicating that a conversion has started and will stay LOW until the conversion is complete. Just prior to BUSY rising near the end of the conversion, the internal working register holding the conversion result will be transferred to the internal shift register. The internal shift register is clocked via the DATACLK input. The recommended method of reading the conversion result is to provide the serial clock after the conversion has completed. See External DATACLK under the Reading Data section of this data sheet for more information. INTERNAL DATACLK Figure 1a shows a basic circuit to operate the ADS7813 with a ±10V input range. To begin a conversion and serial transmission of the results from the previous conversion, a falling edge must be provided to the CONV input. BUSY will go LOW indicating that a conversion has started and will stay LOW until the conversion is complete. During the conversion, the results of the previous conversion will be transmitted via DATA while DATACLK provides the synchronous clock for the serial data. The data format is 16-bit, Binary Two’s Complement, and MSB first. Each data bit is valid on both the rising and falling edge of DATACLK. BUSY is LOW during the entire serial transmission and can be used as a frame synchronization signal. C2 C1 0.1µF 10µF ADS7813 ±10V C3 1µF + C4 0.01µF C5 1µF + 1 R1IN VS 16 2 GND PWRD 15 3 R2IN BUSY 14 4 R3IN CS 13 5 BUF CONV 12 6 CAP EXT/INT 11 7 REF DATA 10 8 GND DATACLK +5V + Frame Sync (optional) Convert Pulse 40ns min 9 FIGURE 1a. Basic Operation, ±10V Input Range, Internal DATACLK. C1 C2 0.1µF 10µF ADS7813 ±10V C3 1µF + C4 0.01µF C5 1µF + 1 R1IN VS 16 2 GND PWRD 15 3 R2IN BUSY 14 4 R3IN CS 13 5 BUF CONV 12 6 CAP EXT/INT 11 7 REF DATA 10 8 GND DATACLK 9 +5V + Interrupt (optional) Chip Select (optional(1)) Convert Pulse +5V 40ns min External Clock NOTE: (1) Tie CS to GND if the outputs will always be active. FIGURE 1b. Basic Operation, ±10V Input Range, External DATACLK. ADS7813 SBAS043C www.ti.com 7 SYMBOL DESCRIPTION t1 Conversion Plus Acquisition Time 25 µs t2 CONV LOW to All Digital Inputs Stable 8 µs STARTING A CONVERSION MIN TYP MAX UNITS t3 CONV LOW to Initiate a Conversion 40 ns t4 BUSY Rising to Any Digital Input Active 0 ns t5 CONV HIGH Prior to Start of Conversion 2 µs t6 BUSY LOW 19 20 µs t7 CONV LOW to BUSY LOW 85 120 ns t8 Aperture Delay 40 t9 Conversion Time 18 20 µs t10 Conversion Complete to BUSY Rising 1.1 2 µs 5 µs t11 Acquisition Time t12 CONV LOW to Rising Edge of First DATACLK If a conversion is not currently in progress, a falling edge on the CONV input places the sample and hold into the hold mode and begins a conversion, as shown in Figure 2 and with the timing given in Table II. During the conversion, the CONV input is ignored. Starting a conversion does not depend on the state of CS. A conversion can be started once every 25µs (40kHz maximum conversion rate). There is no minimum conversion rate. Even though the CONV input is ignored while a conversion is in progress, this input should be held static during the conversion period. Transitions on this digital input can easily couple into sensitive analog portions of the converter, adversely affecting the conversion results (see the Sensitivity to External Digital Signals section of this data sheet for more information). Ideally, the CONV input should go LOW and remain LOW throughout the conversion. It should return HIGH sometime after BUSY goes HIGH. In addition, it should be HIGH prior to the start of the next conversion for a minimum time period given by t5. This will ensure that the digital transition on the CONV input will not affect the signal that is acquired for the next conversion. An acceptable alternative is to return the CONV input HIGH as soon after the start of the conversion as possible. For example, a negative going pulse 100ns wide would make a good CONV input signal. It is strongly recommended that from time t2 after the start of a conversion until BUSY rises, the CONV input should be held static (either HIGH or LOW). During this time, the converter is more sensitive to external noise. ns µs 1.4 t13 Internal DATACLK HIGH 250 350 500 t14 Internal DATACLK LOW 600 760 875 ns t15 Internal DATACLK Period t16 DATA Valid to Internal DATACLK Rising 20 ns t17 Internal DATACLK Falling to DATA Not Valid 400 ns t18 Falling Edge of Last DATACLK to BUSY Rising t19 External DATACLK Rising to DATA Not Valid t20 External DATACLK Rising to DATA Valid ns µs 1.1 800 ns 15 ns 55 85 ns t21 External DATACLK HIGH 50 ns t22 External DATACLK LOW 50 ns t23 External DATACLK Period 100 ns t24 CONV LOW to External DATACLK Active 100 ns t25 External DATACLK LOW or CS HIGH to BUSY Rising 2 µs t26 CS LOW to Digital Outputs Enabled 85 ns t27 CS HIGH to Digital Outputs Disabled 85 ns TABLE II. ADS7813 Timing. TA = –40°C to +85°C. t1 t2 t3 t4 t5 CONV t6 t7 BUSY t8 t10 t9 MODE Acquire t11 Convert Acquire Convert FIGURE 2. Basic Conversion Timing. 8 ADS7813 www.ti.com SBAS043C DESCRIPTION DIGITAL OUTPUT ANALOG INPUT BINARY TWO’S COMPLEMENT ±10V 305µV 0.5V to 4.5V 61µV BINARY CODE HEX CODE 9.999695V 4.499939V 0111 1111 1111 1111 7FFF 0V 2.5V 0000 0000 0000 0000 0000 –305µV 2.499939µV 1111 1111 1111 1111 FFFF –10V 0.5V 10000 0000 0000 0000 8000 Full-Scale Range Least Significant Bit (LSB) +Full Scale –1LSB Midscale Midscale –1LSB –Full Scale TABLE III. Ideal Input Voltage and Corresponding Digital Output for Two Common Input Ranges. Converter Core REF CDAC CONV Clock Control Logic BUSY Each flip-flop in the working register is latched as the conversion proceeds Working Register D Q D Q D Q D Q D Q ••• W0 W1 W2 W14 W15 Update of the shift register occurs just prior to BUSY Rising(1) Shift Register D Q D Q D Q D Q D Q D DATA Q EXT/INT S0 S1 S2 S14 S15 SOUT Delay DATACLK CS NOTE: (1) If EXT/INT is HIGH (external clock), DATACLK is HIGH, and CS is LOW during this time, the shift register will not be updated and the conversion result will be lost. FIGURE 3. Block Diagram of the ADS7813’s Digital Inputs and Outputs. READING DATA CONV t25 t6 – t25 BUSY NOTE: Update of the internal shift register occurs in the shaded region. If EXT/INT is HIGH, then DATACLK must be LOW or CS must be HIGH during this time. FIGURE 4. Timing of the Shift Register Update. The ADS7813 digital output is in Binary Two’s Complement (BTC) format. Table III shows the relationship between the digital output word and the analog input voltage under ideal conditions. Figure 3 shows the relationship between the various digital inputs, digital outputs, and internal logic of the ADS7813. Figure 4 shows when the internal shift register of the ADS7813 is updated and how this relates to a single conversion cycle. Together, these two figures point out a very important aspect of the ADS7813: the conversion result is not available until after the conversion is complete. The implications of this are discussed in the following sections. ADS7813 SBAS043C www.ti.com 9 INTERNAL DATACLK With EXT/INT tied LOW, the result from conversion ‘n’ is serially transmitted during conversion ‘n+1’, as shown in Figure 5 and with the timing given in Table II. Serial transmission of data occurs only during a conversion. When a transmission is not in progress, DATA and DATACLK are LOW. During the conversion, the results of the previous conversion will be transmitted via DATA, while DATACLK provides the synchronous clock for the serial data. The data format is 16-bit, Binary Two’s Complement, and MSB first. Each data bit is valid on both the rising and falling edges of DATACLK. BUSY is LOW during the entire serial transmission and can be used as a frame synchronization signal. EXTERNAL DATACLK With EXT/INT tied HIGH, the result from conversion ‘n’ is clocked out after the conversion has completed, during the next conversion (‘n+1’), or a combination of these two. Figure 6 shows the case of reading the conversion result after the conversion is complete. Figure 7 describes reading the result during the next conversion. Figure 8 combines the important aspects of Figures 6 and 7 as to reading part of the result after the conversion is complete and the remainder during the next conversion. The serial transmission of the conversion result is initiated by a rising edge on DATACLK. The data format is 16-bit, Binary Two’s Complement, and MSB first. Each data bit is valid on the falling edge of DATACLK. In some cases, it t1 CONV BUSY t13 t12 t15 DATACLK 1 2 t18 3 t16 14 15 16 1 Bit 2 Bit 1 LSB t14 t17 DATA MSB Bit 14 Bit 13 MSB FIGURE 5. Serial Data Timing, Internal Clock (EXT/INT and CS LOW). t1 t5 CONV BUSY t21 t4 DATACLK t23 1 2 3 t19 4 14 15 16 t22 t20 DATA MSB Bit 14 Bit 13 Bit 2 Bit 1 LSB FIGURE 6. Serial Data Timing, External Clock, Clocking After the Conversion Completes (EXT/INT HIGH, CS LOW). 10 ADS7813 www.ti.com SBAS043C completed and before the next conversion starts—as shown in Figure 6. Note that the DATACLK signal should be static before the start of the next conversion. If this is not observed, the DATACLK signal could affect the voltage that is acquired. might be possible to use the rising edge of the DATACLK signal. However, one extra clock period (not shown in Figures 6, 7, and 8) is needed for the final bit. The external DATACLK signal must be LOW or CS must be HIGH prior to BUSY rising (see time t25 in Figures 7 and 8). If this is not observed during this time, the output shift register of the ADS7813 will not be updated with the conversion result. Instead, the previous contents of the shift register will remain and the new result will be lost. Before reading the next three paragraphs, consult the Sensitivity to External Digital Signals section of this data sheet. This will explain many of the concerns regarding how and when to apply the external DATACLK signal. External DATACLK Active During the Next Conversion Another method of obtaining the conversion result is shown in Figure 7. Since the output shift register is not updated until the end of the conversion, the previous result remains valid during the next conversion. If a fast clock (≥ 2MHz) can be provided to the ADS7813, the result can be read during time t2. During this time, the noise from the DATACLK signal is less likely to affect the conversion result. External DATACLK Active After the Conversion The preferred method of obtaining the conversion result is to provide the DATACLK signal after the conversion has been t1 t2 CONV BUSY t21 t24 t23 DATACLK 1 2 3 t19 t25 4 15 16 1 t22 t20 DATA MSB Bit 14 Bit 13 Bit 1 LSB MSB FIGURE 7. Serial Data Timing, External Clock, Clocking During the Next Conversion (EXT/INT HIGH, CS LOW). CONV BUSY t5 t24 t4 DATACLK DATA 1 2 MSB n Bit 14 t25 n+1 Bit n Bit n-1 15 16 Bit 1 LSB FIGURE 8. Serial Data Timing, External Clock, Clocking After the Conversion Completes and During the Next Conversion (EXT/INT HIGH, CS LOW). ADS7813 SBAS043C www.ti.com 11 External DATACLK Active After the Conversion and During the Next Conversion Figure 8 shows a method that is a hybrid of the two previous approaches. This method works very well for microcontrollers that do serial transfers 8 bits at a time and for slower microcontrollers. For example, if the fastest serial clock that the microcontroller can produce is 1µs, the approach shown in Figure 6 would result in a diminished throughput (26kHz maximum conversion rate). The method described in Figure 7 could not be used without risk of affecting the conversion result (the clock would have to be active after time t2). The approach in Figure 8 results in an improved throughput rate (33kHz maximum with a 1µs clock) and DATACLK is not active after time t2. CHIP SELECT (CS) The CS input allows the digital outputs of the ADS7813 to be disabled and gates the external DATACLK signal when EXT/INT is HIGH. See Figure 9 for the enable and disable time associated with CS and Figure 3 for a block diagram of the ADS7813’s logic. The digital outputs can be disabled at any time. Note that a conversion is initiated on the falling edge of CONV even if CS is HIGH. If the EXT/INT input is LOW (internal DATACLK) and CS is HIGH during the entire conversion, the previous conversion result will be lost (the serial transmission occurs but DATA and DATACLK are disabled). CS COMPATIBILITY WITH THE ADS7812 The only difference between the ADS7812 and the ADS7813 is in the internal control logic and the digital interface. Since the ADS7812 is a 12-bit converter, the internal shift register is 12 bits wide. In addition, only 12-bit decisions are made during the conversion. Thus, the ADS7812’s conversion time is approximately 75% of the ADS7813’s. In the internal DATACLK mode, the ADS7812 produces 12 DATACLK periods during the conversion instead of the ADS7813’s 16 (see Figure 5). In the external DATACLK mode, the ADS7812 can accept 16 clock periods on DATACLK. At the start of the 13th clock cycle, the DATA output will go LOW and remain LOW. Thus, Figures 6, 7, 8, and the associated times in Table II can also be used for the ADS7812, but the last four bits of the conversion result will be zero. ANALOG INPUT RANGE (V) CONNECT R1IN TO CONNECT R2IN TO 0.3125 to 2.8125 VIN –0.417 to 2.916 VIN 0.417 to 3.750 ±3.333 t26 BUSY, DATA, DATACLK(1) t27 HI-Z Active HI-Z NOTE: (1) DATACLK is an output only when EXT/INT is LOW. FIGURE 9. Enable and Disable Timing for Digital Outputs. ANALOG INPUT The ADS7813 offers a number of input ranges. This is accomplished by connecting the three input resistors to either the analog input (VIN), to ground (GND), or to the 2.5V reference buffer output (BUF). Table I shows the input ranges that are typically used in most data acquisition applications. These ranges are all specified to meet the specifications given in the Specifications table. Table IV contains a complete list of ideal input ranges, associated input connections, and comments regarding the range. CONNECT R3IN TO INPUT IMPEDANCE (kΩ) VIN VIN > 10,000 VIN BUF 26.7 VIN VIN GND 26.7 Offset and gain not specified VIN BUF VIN 21.3 Specified offset and gain –15 to 5 VIN BUF BUF 45.7 Offset and gain not specified ±10 VIN BUF GND 45.7 Specified offset and gain 0.833 to 7.5 VIN GND VIN 21.3 Offset and gain not specified –2.5 to 17.5 VIN GND BUF 45.7 Exceeds absolute maximum VIN 2.5 to 22.5 VIN GND GND 45.7 Exceeds absolute maximum VIN 0 to 2.857 BUF VIN VIN 45.7 Offset and gain not specified VIN cannot go below GND – 0.3V COMMENT Specified offset and gain VIN cannot go below GND – 0.3V –1 to 3 BUF VIN BUF 21.3 0 to 4 BUF VIN GND 21.3 Specified offset and gain –6.25 to 3.75 BUF BUF VIN 26.7 Offset and gain not specified Specified offset and gain 0 to 10 BUF GND VIN 26.7 0.357 to 3.214 GND VIN VIN 45.7 Offset and gain not specified –0.5 to 3.5 GND VIN BUF 21.3 VIN cannot go below GND – 0.3V 0.5 to 4.5 GND VIN GND 21.3 Specified offset and gain ±5 GND BUF VIN 26.7 Specified offset and gain 1.25 to 11.25 GND GND VIN 26.7 Offset and gain not specified TABLE IV. Complete List of Ideal Input Ranges. 12 ADS7813 www.ti.com SBAS043C Likewise, three of the input ranges involve the connection at R2IN being driven below GND. This input has a reversebiased ESD protection diode connection to ground. If R2IN is taken below GND – 0.3V, this diode will be forward-biased and will clamp the negative input at –0.4V to –0.7V, depending on the temperature. Since the negative full-scale value of these input ranges exceed –0.4V, they are not recommended. Note that Table IV assumes that the voltage at the REF pin is +2.5V. This is true if the internal reference is being used or if the external reference is +2.5V. Other reference voltages will change the values in Table IV. HIGH IMPEDANCE MODE When R1IN, R2IN, and R3IN are connected to the analog input, the input range of the ADS7813 is 0.3125V to 2.8125V and the input impedance is greater than 10MΩ. This input range can be used to connect the ADS7813 directly to a wide variety of sensors. Figure 10 shows the impedance of the sensor versus the change in ILE and DLE of the ADS7813. The performance of the ADS7813 can be improved for higher sensor impedance by allowing more time for acquisition. For example, 10µs of acquisition time will approximately double sensor impedance for the same ILE/DLE performance. The input impedance and capacitance of the ADS7813 are very stable with temperature. Assuming that this is true of the sensor as well, the graph shown in Figure 10 will vary less than a few percent over the ensured temperature range of the ADS7813. If the sensor impedance varies significantly with temperature, the worst-case impedance should be used. DRIVING THE ADS7813 ANALOG INPUT In general, any reasonably fast, high-quality operational or instrumentation amplifier can be used to drive the ADS7813 input. When the converter enters the acquisition mode, there is some charge injection from the converter input to the amplifier output. This can result in inadequate settling time with slower amplifiers. Be very careful with single-supply amplifiers, particularly if their output will be required to swing very close to the supply rails. In addition, be careful in regards to the amplifier linearity. The outputs of single-supply and rail-to-rail amplifiers can saturate as they approach the supply rails. Rather than the amplifier transfer function being a straight line, the curve can become severely ‘S’ shaped. Also, watch for the point where the amplifier switches from sourcing current to sinking current. For some amplifiers, the transfer function can be noticeably discontinuous at this point, causing a significant change in the output voltage for a much smaller change on the input. Texas Instruments manufactures a wide variety of operational and instrumentation amplifiers that can be used to drive the input of the ADS7813. These include the OPA627, OPA132, and INA110. REFERENCE The ADS7813 can be operated with its internal 2.5V reference or an external reference. By applying an external reference voltage to the REF pin, the internal reference voltage is overdriven. The voltage at the REF input is internally buffered by a unity gain buffer. The output of this buffer is present at the BUF and CAP pins. REF The REF pin is the output of the internal 2.5V reference or the input for an external reference. A 1µF to 2.2µF tantulum capacitor should be connected between this pin and ground. The capacitor should be placed as close to the ADS7813 as possible. When using the internal reference, the REF pin should not be connected to any type of significant load. An external load will cause a voltage drop across the internal 4kΩ resistor that is in series with the internal reference. Even a 40MΩ external load to ground will cause a decrease in the full-scale range of the converter by 6 LSBs. LINEARITY ERROR vs SOURCE IMPEDANCE 10 9 Change in Worst-Case Linearity Error (LSBs) The input impedance results from the various connections and the internal resistor values (refer to the block diagram on the front page of this data sheet). The internal resistor values are typical and can change by ±30%, due to process variations. However, the ratio matching of the resistors is considerably better than this. Thus, the input range will vary only a few tenths of a percent from part to part, while the input impedance can vary up to ±30%. The Specifications table contains the maximum limits for the variation of the analog input range, but only for those ranges where the comment field shows that the offset and gain are specified (this includes all the ranges listed in Table I). For the other ranges, the offset and gain are not tested and are not specified. Five of the input ranges in Table IV are not recommended for general use. The upper-end of the –2.5V to +17.5V range and +2.5V to +22.5V range exceed the absolute maximum analog input voltage. These ranges can still be used as long as the input voltage remains under the absolute maximum, but this will moderately to significantly reduce the full-scale range of the converter. DLE 7 6 ILE 5 4 3 2 1 0 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 External Source Impedance (kΩ) FIGURE 10. Linearity Error vs Source Impedance in the High Impedance Mode (R1IN = R2IN = R3IN = VIN). ADS7813 SBAS043C TA = +25°C Acquisition Time = 5µs 8 www.ti.com 13 CAP The CAP pin is used to compensate the internal reference buffer. A 1µF tantalum capacitor in parallel with a 0.01µF ceramic capacitor should be connected between this pin and ground, with the ceramic capacitor placed as close to the ADS7813 as possible. The total value of the capacitance on the CAP pin is critical to optimum performance of the ADS7813. A value larger than 2.0µF could overcompensate the buffer while a value lower than 0.5µF may not provide adequate compensation. The ESR (equivalent series resistance) of these compensation capacitors is also critical. Keep the total ESR under 3Ω. See the Typical Characteristic curve, |Worst-Case INL| vs ESR of Reference Buffer Compensation Capacitor, for how the worst-case INL is affected by ESR. BUF The voltage on the BUF pin is the output of the internal reference buffer. This pin is used to provide +2.5V to the analog input or inputs for the various input configurations. The BUF output can provide up to 1mA of current to an external load. The load should be constant as a variable load could affect the conversion result by modulating the BUF voltage. Also note that the BUF output will show significant glitches as each bit decision is made during a conversion. Between conversions, the BUF output is quiet. power-down mode, CONV is taken HIGH and then PWRD is taken LOW. Note that a conversion will be initiated if PWRD is taken HIGH while CONV is LOW. While in the power-down mode, the voltage on the capacitors connected to CAP and REF will begin to leak off. The voltage on the CAP capacitor leaks off much more rapidly than on the REF capacitor (the REF input of the ADS7813 becomes high-impedance when PWRD is HIGH—this is not true for the CAP input). When the power-down mode is exited, these capacitors must be allowed to recharge and settle to a 16-bit level. Figure 11 shows the amount of time typically required to obtain a valid 16-bit result based on the amount of time spent in power down (at room temperature). This figure assumes that the total capacitance on the CAP pin is 1.01µF. Figure 12 provides a circuit which can significantly reduce the power up time if the power down time will be fairly brief (a few seconds or less). A low on-resistance MOSFET is used to disconnect the capacitance on the CAP pin from the leakage paths internal to the ADS7813. This allows the capacitors to retain their charge for a much longer period of time, reducing the time required to recharge them at power up. With this circuit, the power-down time can be extended to tens or hundreds of milliseconds with almost instantaneous power up. POWER-DOWN TO POWER-UP RESPONSE Power-Up Time to Rated Accuracy (µs) The range for the external reference is 2.3V to 2.7V. The voltage on REF determines the full-scale range of the converter and the corresponding LSB size. Increasing the reference voltage will increase the LSB size in relation to the internal noise sources which, in turn, can improve signal-tonoise ratio. Likewise, decreasing the reference voltage will reduce the LSB size and signal-to-noise ratio. POWER DOWN The ADS7813 has a power-down mode that is activated by taking CONV LOW and then PWRD HIGH. This will power down all of the analog circuitry including the reference, reducing power dissipation to under 50µW. To exit the 300 TA = +25°C 250 200 150 100 50 0 0.1 1 10 100 Power-Down Duration (ms) FIGURE 11. Power-Down to Power-Up Response. 1RF7604 + 1µF 1 8 1 R1IN VS 16 2 7 2 GND PWRD 15 3 6 3 R2IN BUSY 14 4 5 4 R3IN CS 13 5 BUF CONV 12 6 CAP EXT/INT 11 7 REF 8 GND 0.01µF Power-Down Signal DATA 10 DATACLK 9 FIGURE 12. Improved Power-Up Response Circuit. 14 ADS7813 www.ti.com SBAS043C LAYOUT For example, the timing diagram in Figure 2 shows that the CONV signal should return HIGH sometime during time t2. In fact, the CONV signal can return HIGH at any time during the conversion. However, after time t2, the transition of the CONV signal has the potential of creating a good deal of noise on the ADS7813 die. If this transition occurs at just precisely the wrong time, the conversion results could be affected. In a similar manner, transitions on the DATACLK input could affect the conversion result. For the ADS7813, there are 16 separate bit decisions which are made during the conversion. The most significant bit decision is made first, proceeding to the least significant bit at the end of the conversion. Each bit decision involves the assumption that the bit being tested should be set. This is combined with the result that has been achieved so far. The converter compares this combined result with the actual input voltage. If the combined result is too high, the bit is cleared. If the result is equal to or lower than the actual input voltage, the bit remains HIGH. This is why the basic architecture is referred to as successive approximation register (SAR). If the result so far is getting very close to the actual input voltage, then the comparison involves two voltages which are very close together. The ADS7813 has been designed so that the internal noise sources are at a minimum just prior to the comparator result being latched. However, if an external digital signal transitions at this time, a great deal of noise will be coupled into the sensitive analog section of the ADS7813. Even if this noise produces a difference between the two voltages of only 2mV, the conversion result will be off by 52 counts or least significant bits (LSBs). (The internal LSB size of the ADS7813 is 38µV regardless of the input range.) Once a digital transition has caused the comparator to make a wrong bit decision, the decision cannot be corrected (unless some type of error correction is employed). All subsequent bit decisions will then be wrong. Figure 13 shows a successive approximation process that has gone wrong. The dashed line represents what the correct bit decisions should have been. The solid line represents the actual result of the conversion. The ADS7813 should be treated as a precision analog component and should reside completely on the analog portion of the printed circuit board. Ideally, a ground plane should extend underneath the ADS7813 and under all other analog components. This plane should be separate from the digital ground until they are joined at the power supply connection. This will help prevent dynamic digital ground currents from modulating the analog ground through a common impedance to power ground. The +5V power should be clean, well-regulated, and separate from the +5V power for the digital portion of the design. One possibility is to derive the +5V supply from a linear regulator located near the ADS7813. If derived from the digital +5V power, a 5Ω to 10Ω resistor should be placed in series with the power connection from the digital supply. It may also be necessary to increase the bypass capacitance near the VS pin (an additional 100µF or greater capacitor in parallel with the 10µF and 0.1µF capacitors). For designs with a large number of digital components or very high speed digital logic, this simple power supply filtering scheme may not be adequate. SENSITIVITY TO EXTERNAL DIGITAL SIGNALS All successive approximation register-based A/D converters are sensitive to external sources of noise. The reason for this will be explained in the following paragraphs. For the ADS7813 and similar A/D converters, this noise most often originates due to the transition of external digital signals. While digital signals that run near the converter can be the source of the noise, the biggest problem occurs with the digital inputs to the converter itself. In many cases, the system designer may not be aware that there is a problem or a potential for a problem. For a 12-bit system, these problems typically occur at the least significant bits and only at certain places in the converter’s transfer function. For a 16-bit converter, the problem can be much easier to spot. External Noise SAR Operation after Wrong Bit Decision Actual Input Voltage Converter’s Full-Scale Input Voltage Range Proper SAR Operation Internal DAC Voltage Wrong Bit Decision Made Here t Conversion Clock Conversion Start (Hold Mode) 1 1 0 0 0 0 Incorrect Result (1 0 1 1 0 1) Correct Result FIGURE 13. SAR Operation When External Noise Affects the Conversion. ADS7813 SBAS043C www.ti.com 15 Keep in mind that the time period when the comparator is most sensitive to noise is fairly small. Also, the peak portion of the noise event produced by a digital transition is fairly brief, as most digital signals transition in a few nanoseconds. The subsequent noise may last for a period of time longer than this and may induce further effects which require a longer settling time. However, in general, the event is over within a few tens of nanoseconds. For the ADS7813, error correction is done when the tenth bit is decided. During this bit decision, it is possible to correct limited errors that may have occurred during previous bit decisions. However, after the tenth bit, no such correction is possible. Note that for the timing diagrams shown in Figures 2, 5, 6, 7, and 8, all external digital signals should remain static from 8µs after the start of a conversion until BUSY rises. The tenth bit is decided approximately 10µs to 11µs into the conversion. APPLICATIONS INFORMATION TRANSITION NOISE If a low-noise DC input is applied to the ADS7813 and 1,000 conversions are performed, the digital output of the converter will vary slightly in output codes. This is true for all 16-bit SAR converters. The Transition Noise specification found in the Electrical Characteristics section is a statistical figure that represents the 1 sigma (σ) limit of these output codes. Using a histogram to plot the number of occurrences of each output code, the distribution should appear bell-shaped with the peak of the curve representing the nominal output code for the given input voltage. The ±1σ, ±2σ, and ±3σ limits around this nominal code should contain 68.3%, 95.5%, and 99.7%, respectively, of the conversion results. As a rough approximation, multiplying transition noise by 6 (±3σ) will yield the number of unique output codes which should be present in 1,000 conversions. The ADS7813 has a transition noise figure of 0.6LSB, yielding approximately 4 different output codes for 1,000 conversions. However, since ±3σ is only 99.7%, up to three conversions have some chance of being outside this range. In addition, the differential linearity error of each code and the quantization performed by the converter result in histograms which can deviate from the ideal. Figure 14 shows a histogram of 5,000 conversions from the ADS7813. AVERAGING The noise of the converter can be reduced by averaging conversion results. The noise will be reduced by a factor of 1/√n, where n is the number of averages. For example, averaging four conversions will reduce transition noise by half, to 0.3LSBs. Averaging should only be used for lowfrequency signals. For higher frequency signals, a digital filter can be used to reduce noise. This works in a similar manner to averaging: for every reduction in the signal bandwidth by two, the signal-to-noise ratio will improve by 3dB. QSPI INTERFACING Figure 15 shows a simple interface between the ADS7813 and any queued serial peripheral interface (QSPI) equipped microcontroller (available on several Motorola devices). This interface assumes that the convert pulse does not originate from the microcontroller and that the ADS7813 is the only serial peripheral. Convert Pulse QSPI ADS7813 CONV 3291 PCS0/SS BUSY MOSI DATA SCK DATACLK CS EXT/INT CPOL = 0 (Inactive State is LOW) CPHA = 1 (Data valid on falling edge) QSPI port is in slave mode. FIGURE 15. QSPI Interface to the ADS7813. 832 821 0 23 FFFDh FFFEh FFFFh 0000h 0001h 33 0 0002h 0003h FIGURE 14. Histogram of 5,000 Conversions with Input Grounded. 16 Before enabling the QSPI interface, the microcontroller must be configured to monitor the slave select (SS) line. When a LOW to HIGH transition occurs (indicating the end of a conversion), the port can be enabled. If this is not done, the microcontroller and A/D converter may not be properly synchronized. (The slave select line simply enables communication—it does not indicate the start or end of a serial transfer.) ADS7813 www.ti.com SBAS043C Figure 16 shows a QSPI-equipped microcontroller interfacing to three ADS7813s. There are many possible variations to this interface scheme. As shown, the QSPI port produces a common CONV signal which initiates a conversion on all three converters. After the conversions are finished, each result is transferred in turn. The QSPI port is completely programmable to handle the timing and transfers without processor intervention. If the CONV signal is generated in this way, it should be possible to make both AC and DC measurements with the ADS7813, as the CONV signal will have low jitter. Note that if the CONV signal is generated via software commands, it will have a good deal of jitter and only low frequency (DC) measurements can be made. QSPI ADS7813 PCS0 CONV PCS1 CS +5V EXT/INT PCS2 PCS3 SCK MISO DATACLK SPI INTERFACING The serial peripheral interface (SPI) is directly related to the QSPI and both Figures 15 and 16 can be used as a guide for connecting the ADS7813 to SPI-equipped microcontrollers. For most microcontrollers, the SPI port is capable of 8-bit transfers only. In the case of Figure 15, be aware that the microcontroller may have to be capable of fetching the 8 most significant bits before they are overwritten by the 8 least significant bits. DSP56002 INTERFACING The DSP56002 serial interface has an SPI compatibility mode with some enhancements. Figure 17 shows an interface between the ADS7813 and the DSP56002. As with the QSPI interface of Figure 15, the DSP56002 must be programmed to enable the serial interface when a LOW to HIGH transition on SCI occurs. The DSP56002 can also provide the CONV signal, as shown in Figure 18. The receive and transmit sections of the interface are decoupled (asynchronous mode) and the transmit section is set to generate a word length frame sync every other transmit frame (frame rate divider set to 2). The prescale modulus should be set to produce a transmit frame at twice the desired conversion rate. DATA Convert Pulse ADS7813 CONV +5V DSP56002 EXT/INT ADS7813 CS CONV DATACLK DATA ADS7813 CONV +5V SC1 BUSY SRD DATA SCO DATACLK EXT/INT CS CS EXT/INT DATACLK SYN = 0 (Asychronous) GCK = 1 (Gated clock) SCD1 = 0 (SC1 is an input) SHFD = 0 (Shift MSB first) WL1 = 1 WL0 = 0 (Word length = 16 bits) DATA FIGURE 16. QSPI Interface to Three ADS7813s. FIGURE 17. DSP56002 Interface to the ADS7813. DSP56002 ADS7813 SC2 CONV BUSY SC0 DATACLK SRD DATA CS SYN = 0 (Asychronous) GCK = 1 (Gated clock) SCD2 = 1 (SC2 is an output) SHFD = 0 (Shift MSB first) WL1 = 1 WL0 = 0 (Word length = 16 bits) EXT/INT FIGURE 18. DSP56002 Interface to the ADS7813. Processor Initiates Conversions. ADS7813 SBAS043C www.ti.com 17 PACKAGE OPTION ADDENDUM www.ti.com 1-Dec-2023 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) ADS7813U LIFEBUY SOIC DW 16 40 RoHS & Green NIPDAU-DCC Level-3-260C-168 HR -40 to 85 ADS7813U ADS7813U/1K LIFEBUY SOIC DW 16 1000 RoHS & Green NIPDAU-DCC Level-3-260C-168 HR -40 to 85 ADS7813U ADS7813UB LIFEBUY SOIC DW 16 40 RoHS & Green NIPDAU-DCC Level-3-260C-168 HR -40 to 85 ADS7813U B ADS7813UE4 LIFEBUY SOIC DW 16 40 RoHS & Green NIPDAU-DCC Level-3-260C-168 HR -40 to 85 ADS7813U ADS7813UG4 LIFEBUY SOIC DW 16 40 RoHS & Green NIPDAU-DCC Level-3-260C-168 HR -40 to 85 ADS7813U (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of