MIC24051
12V, 6A High-Efficiency Buck Regulator
with Hyper Speed Control®
Features
General Description
• Hyper Speed Control® Architecture Enables:
- High Delta V Operation (VIN = 19V and VOUT
= 0.8V)
- Small Output Capacitance
• 4.5V to 19V Voltage Input
• 6A Output Current Capability, up to 95%
Efficiency
• Adjustable Output from 0.8V to 5.5V
• ±1% Feedback Accuracy
• Any Capacitor Stable - Zero-to-High ESR
• 600 kHz Switching Frequency
• No External Compensation
• Power Good (PG) Output
• Foldback Current-Limit and “Hiccup Mode”
Short-Circuit Protection
• Supports Safe Start-Up into a Pre-Biased Load
• –40°C to +125°C Junction Temperature Range
• Available in a 28-pin 5 mm x 6 mm QFN Package
The MIC24051 is a constant-frequency, synchronous
buck regulator featuring a unique adaptive on-time
control architecture. The MIC24051 operates over an
input supply range of 4.5V to 19V and provides a
regulated output of up to 6A of output current. The
output voltage is adjustable down to 0.8V with a
guaranteed accuracy of ±1%, and the device operates
at a switching frequency of 600 kHz.
Microchip’s Hyper Speed Control® architecture allows
for ultra-fast transient response while reducing the
output capacitance and also makes (High VIN)/(Low
VOUT) operation possible. This adaptive tON ripple
control architecture combines the advantages of
fixed-frequency operation and fast transient response
in a single device.
The MIC24051 offers a full suite of protection features
to ensure protection of the IC during fault conditions.
These include undervoltage lockout to ensure proper
operation under power-sag conditions, internal
soft-start to reduce inrush current, foldback current
limit, “hiccup mode” short-circuit protection and thermal
shutdown. An open-drain Power Good (PG) pin is
provided.
Applications
• Servers and Workstations
• Routers, Switches, and Telecom Equipment
• Base Stations
The 6A HyperLight Load® part, MIC24052, is also
available on Microchip’s web site.
Typical Application Schematic
MIC24051
28-PIN QFN
MIC24051
2.2μF
10k
4.7μF
x2
10k
EN
2016 Microchip Technology Inc.
BST
0.1μF
2.2μH
PG
PG
VIN
4.5V TO 19V
VDD
PVDD
SGND
VIN
PVIN
PGND
EN
SW
CS
VOUT
1.8V/6A
0.1μF
4.7nF
19.6k
2.49k
100μF
FB
2.00k
DS20005658A-page 1
MIC24051
Package Type
VDD
VIN
EN
PG
FB
SGND
MIC24051
28-Pin QFN (JL)
(Top View)
28 27 26 25
24 23
PVDD
PGND
NC
1
3
20
SW
PGND
PGND
PGND
PGND
4
19
5
18
22
PGND
2
6
SW
7
21
PVIN
17
16
8
13 14
PVIN
PVIN
PVIN
PVIN
PVIN
PVIN
PVIN
9 10 11 12
SW
SW
SW
SW
15
CS
PGND
BST
Block Diagram
D1
MIC24051
VDD
PVDD
2.2μF
VIN
LDO
FIXED TON
ESTIMATE
VDD
VIN
UVLO
PVIN
MODIFIED
TOFF
10k
0.1μF
CBST
HSD
CONTROL
LOGIC
TIMER
SOFT-START
EN
VIN
4.5V to 19V
4.7μF
x2
BST
SW
CL and ZC
DETECTION
2.2μH
0.1μF
SOFT
START
THERMAL
SHUTDOWN
PVDD
LSD
INTERNAL
RIPPLE
INJECTION
VDD
VOUT
1.8V/6A
CS
R1
2.49k
4.7nF
100μF
19.6k
SGND
PGND
COMPENSATION
10k
gm EA
COMP
PG
FB
R2
2.00k
8%
VREF
0.8V
92%
DS20005658A-page 2
2016 Microchip Technology Inc.
MIC24051
1.0
ELECTRICAL CHARACTERISTICS
Absolute Maximum Ratings †
PVIN to PGND ............................................................................................................................................ –0.3V to +29V
VIN to PGND ............................................................................................................................................... –0.3V to PVIN
PVDD, VDD to PGND .................................................................................................................................... –0.3V to +6V
VSW, VCS to PGND ....................................................................................................................... –0.3V to (PVIN + 0.3V)
VBST to VSW ................................................................................................................................................. –0.3V to +6V
VBST to PGND............................................................................................................................................ –0.3V to +35V
VFB, VPG to PGND ......................................................................................................................... –0.3V to (VDD + 0.3V)
VEN to PGND ...................................................................................................................................–0.3V to (VIN + 0.3V)
PGND to SGND ........................................................................................................................................ –0.3V to +0.3V
ESD Rating (Note 1) .................................................................................................................................. ESD Sensitive
Operating Ratings ‡
Supply Voltage (PVIN, VIN)......................................................................................................................... +4.5V to +19V
PVDD, VDD Supply Voltage (PVDD, VDD)................................................................................................... +4.5V to +5.5V
Enable Input (VEN) ..............................................................................................................................................0V to VIN
† Notice: Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device.
This is a stress rating only and functional operation of the device at those or any other conditions above those indicated
in the operational sections of this specification is not intended. Exposure to maximum rating conditions for extended
periods may affect device reliability.
‡ Notice: The device is not guaranteed to function outside its operating ratings.
Note 1: Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5 kΩ in series with
100 pF.
2016 Microchip Technology Inc.
DS20005658A-page 3
MIC24051
TABLE 1-1:
ELECTRICAL CHARACTERISTICS
Electrical Characteristics: PVIN = VIN = VEN = 12V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values
indicate –40°C ≤ TJ ≤ +125°C. (Note 1).
Parameters
Sym.
Min.
Typ.
Max.
Units
Conditions
Input Voltage Range (VIN, PVIN)
—
4.5
—
19
V
—
Quiescent Supply Current
—
—
730
1500
µA
VFB = 1.5V (non-switching)
Shutdown Supply Current
—
—
5
10
µA
VEN = 0V
VDD Output Voltage
—
4.8
5
5.4
V
VIN = 7V to 19V, IDD = 40 mA
VDD UVLO Threshold
—
3.7
4.2
4.5
V
VDD Rising
VDD UVLO Hysteresis
—
400
mV
—
Dropout Voltage (VIN – VDD)
—
380
600
mV
IDD = 25 mA
Power Supply Input
VDD Supply Voltage
DC/DC Controller
Output-Voltage Adjust Range
(VOUT)
—
0.8
—
5.5
V
—
—
0.792
0.8
0.808
V
0°C ≤ TJ ≤ +85°C (±1.0%)
—
0.788
0.8
0.812
V
–40°C ≤ TJ ≤ +125°C (±1.5%)
Load Regulation
—
—
0.25
—
%
IOUT = 0A to 6A (Continuous
Mode)
Line Regulation
—
—
0.25
—
%
VIN = 4.5V to 19V
FB Bias Current
—
—
50
500
nA
VFB = 0.8V
EN Logic Level High
—
1.8
—
—
V
—
EN Logic Level Low
—
—
—
0.6
V
—
EN Bias Current
—
—
6
30
µA
VEN = 12V
Switching Frequency (Note 2)
—
450
600
750
kHz
VOUT = 2.5V
Maximum Duty Cycle (Note 3)
—
—
82
—
%
VFB = 0V
Minimum Duty Cycle
—
—
0
—
%
VFB = 1.0V
Minimum Off-Time
—
—
300
—
ns
—
—
—
3
—
ms
—
7.5
11
17
6.6
11
17
—
8
—
Reference
Feedback Reference Voltage
Enable Control
Oscillator
Soft-Start
Soft-Start Time
Short-Circuit Protection
Peak Inductor Current-Limit
Threshold
—
Short-Circuit Current
—
A
A
VFB = 0.8V, TJ = 25°C
VFB = 0.8V, TJ = 125°C
VFB = 0V
Internal FETs
Note 1:
2:
3:
Specification for packaged product only.
Measured in test mode.
The maximum duty-cycle is limited by the fixed mandatory off-time (tOFF) of typically 300 ns.
DS20005658A-page 4
2016 Microchip Technology Inc.
MIC24051
TABLE 1-1:
ELECTRICAL CHARACTERISTICS (CONTINUED)
Electrical Characteristics: PVIN = VIN = VEN = 12V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values
indicate –40°C ≤ TJ ≤ +125°C. (Note 1).
Parameters
Sym.
Min.
Typ.
Max.
Units
Conditions
Top MOSFET RDS(ON)
—
—
42
—
mΩ
ISW = 1A
Bottom MOSFET RDS(ON)
—
—
12.5
—
mΩ
ISW = 1A
SW Leakage Current
—
—
—
60
µA
VEN = 0V
VIN Leakage Current
—
—
—
25
µA
VEN = 0V
PG Threshold Voltage
—
85
92
95
%VOUT
Sweep VFB from Low to High
PG Hysteresis
—
—
5.5
—
%VOUT
Sweep VFB from High to Low
PG Delay Time
—
—
100
—
µs
Sweep VFB from Low to High
PG Low Voltage
—
—
70
200
mV
Sweep VFB < 0.9 x VNOM,
IPG = 1 mA
Overtemperature Shutdown
—
—
160
—
°C
TJ Rising
Overtemperature Shutdown
Hysteresis
—
—
15
—
°C
—
Power Good (PG)
Thermal Protection
Note 1:
2:
3:
Specification for packaged product only.
Measured in test mode.
The maximum duty-cycle is limited by the fixed mandatory off-time (tOFF) of typically 300 ns.
2016 Microchip Technology Inc.
DS20005658A-page 5
MIC24051
TEMPERATURE SPECIFICATIONS
Parameters
Sym.
Min.
Typ.
Max.
Units
Conditions
TJ
–40
—
+125
°C
Note 1
Temperature Ranges
Junction Operating Temperature
Range
Maximum Junction Temperature
—
—
—
+150
°C
—
Storage Temperature
TS
–65
—
+150
°C
—
Lead Temperature
—
—
—
+260
°C
Soldering, 10s
Thermal Resistance, 5x6 QFN-28
JA
—
28
—
°C/W
Note 2
Thermal Resistance, 5x6 QFN-28
JC
—
2.5
—
°C/W
—
Package Thermal Resistances
Note 1:
2:
The maximum allowable power dissipation is a function of ambient temperature, the maximum allowable
junction temperature and the thermal resistance from junction to air (i.e., TA, TJ, JA). Exceeding the
maximum allowable power dissipation will cause the device operating junction temperature to exceed the
maximum +125°C rating. Sustained junction temperatures above +125°C can impact the device reliability.
PD(MAX) = (TJ(MAX) – TA)/JA, where JA depends upon the printed circuit layout. A 5 square inch 4 layer,
0.62”, FR-4 PCB with 2 oz. finish copper weight per layer is used for the JA.
DS20005658A-page 6
2016 Microchip Technology Inc.
MIC24051
2.0
Note:
TYPICAL PERFORMANCE CURVES
The graphs and tables provided following this note are a statistical summary based on a limited number of
samples and are provided for informational purposes only. The performance characteristics listed herein
are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified
operating range (e.g., outside specified power supply range) and therefore outside the warranted range.
FIGURE 2-1:
VIN Operating Supply
Current vs. Input Voltage.
FIGURE 2-4:
Voltage.
Feedback Voltage vs. Input
FIGURE 2-2:
Input Voltage.
VIN Shutdown Current vs.
FIGURE 2-5:
Voltage.
Total Regulation vs. Input
FIGURE 2-3:
Input Voltage.
VDD Output Voltage vs.
FIGURE 2-6:
Input Voltage.
Output Current Limit vs.
2016 Microchip Technology Inc.
DS20005658A-page 7
MIC24051
FIGURE 2-7:
Input Voltage.
Switching Frequency vs.
FIGURE 2-10:
VIN Operating Supply
Current vs. Temperature.
FIGURE 2-8:
Input Voltage.
Enable Input Current vs.
FIGURE 2-11:
Temperature.
VIN Shutdown Current vs.
FIGURE 2-9:
vs. Input Voltage.
PG Threshold/VREF Ratio
FIGURE 2-12:
Temperature.
VDD UVLO Threshold vs.
DS20005658A-page 8
2016 Microchip Technology Inc.
MIC24051
FIGURE 2-13:
Temperature.
Feedback Voltage vs.
FIGURE 2-16:
Temperature.
Switching Frequency vs.
FIGURE 2-14:
Temperature.
Load Regulation vs.
FIGURE 2-17:
VDD vs. Temperature.
FIGURE 2-15:
Temperature.
Line Regulation vs.
FIGURE 2-18:
Temperature.
Output Current Limit vs.
2016 Microchip Technology Inc.
DS20005658A-page 9
MIC24051
FIGURE 2-19:
Output Voltage.
Switching Frequency vs.
FIGURE 2-22:
Current.
Line Regulation vs. Output
FIGURE 2-20:
Output Current.
Feedback Voltage vs.
FIGURE 2-23:
Output Current.
Switching Frequency vs.
FIGURE 2-21:
Current.
Output Voltage vs. Output
FIGURE 2-24:
Output Voltage (VIN = 5V)
vs. Output Current.
DS20005658A-page 10
2016 Microchip Technology Inc.
MIC24051
FIGURE 2-25:
Output Current.
Efficiency (VIN = 5V) vs.
FIGURE 2-28:
Output Current.
Efficiency (VIN = 12V) vs.
FIGURE 2-26:
IC Power Dissipation (VIN =
5V) vs. Output Current.
FIGURE 2-29:
IC Power Dissipation (VIN =
12V) vs. Output Current.
FIGURE 2-27:
Die Temperature (VIN = 5V)
vs. Output Current (Note 1).
FIGURE 2-30:
Die Temperature (VIN =
12V) vs. Output Current (Note 1).
2016 Microchip Technology Inc.
DS20005658A-page 11
MIC24051
FIGURE 2-31:
Thermal Derating vs.
Ambient Temperature (Note 1).
FIGURE 2-34:
Thermal Derating vs.
Ambient Temperature (Note 1).
VIN
(10V/div)
VIN = 12V, VOUT = 1.8V
IOUT = 6A
VSW
(10V/div)
VOUT
(2V/div)
IL
(5A/div)
Time (2ms/div)
FIGURE 2-32:
Thermal Derating vs.
Ambient Temperature (Note 1).
FIGURE 2-35:
VIN Soft Turn-On.
FIGURE 2-33:
Thermal Derating vs.
Ambient Temperature (Note 1).
FIGURE 2-36:
VIN Soft Turn-Off.
Time (1ms/div)
Note 1: The temperature measurement was taken at the hottest point on the MIC24051 case mounted on a 5 square
inch 4 layer, 0.62”, FR-4 PCB with 2oz finish copper weight per layer, see Thermal Measurement section.
Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting components.
DS20005658A-page 12
2016 Microchip Technology Inc.
MIC24051
VEN
(5V/div)
VIN = 12V
VOUT = 1.8V
IOUT = 6A
VOUT
(1V/div)
IL
(5A/div)
Time (2ms/div)
FIGURE 2-37:
Rise Time.
Enable Turn-On Delay and
FIGURE 2-40:
Enable Turn-On/Turn-Off.
FIGURE 2-38:
Fall Time.
Enable Turn-Off Delay and
FIGURE 2-41:
Enable Thresholds.
VDD
(2V/div)
VOUT
(1V/div)
Time (2ms/div)
FIGURE 2-39:
VIN Start-Up with
Pre-Biased Output.
2016 Microchip Technology Inc.
FIGURE 2-42:
VDD UVLO Thresholds.
DS20005658A-page 13
MIC24051
VIN
(10V/div)
VIN = 12V
VOUT = 1.8V
IOUT = 6A
VOUT
(200mV/div)
IL
(5A/div)
Time (1ms/div)
FIGURE 2-43:
Power-Up Into Short Circuit.
Time (2ms/div)
FIGURE 2-46:
Circuit.
Output Recovery from Short
p
VEN
(2V/div)
VIN = 12V
VOUT = 1.8V
IOUT = SHORT
VOUT
(200mV/div)
VOUT
(1V/div)
VIN = 12V
VOUT = 1.8V
IL
(5A/div)
IOUT
(5A/div)
Time (40ms/div)
FIGURE 2-44:
Enable Into Short Circuit.
FIGURE 2-47:
Threshold.
VIN = 12V
VOUT = 1.8V
IOUT = 6A TO SHORT
Output Current-Limit
VIN = 12V
VOUT = 1.8V
IOUT = 1A
VOUT
(1V/div)
VOUT
(1V/div)
IL
(5A/div)
VSW
(10V/div)
Time (2.0ms/div)
(2ms/div)
Time
FIGURE 2-45:
DS20005658A-page 14
Short Circuit.
FIGURE 2-48:
Output Recovery from
Thermal Shutdown.
2016 Microchip Technology Inc.
MIC24051
(AC-COUPLED)
FIGURE 2-49:
= 6A).
Switching Waveforms (IOUT
(AC-COUPLED)
FIGURE 2-50:
= 0A).
Switching Waveforms (IOUT
(AC-COUPLED)
FIGURE 2-51:
Transient Response.
2016 Microchip Technology Inc.
DS20005658A-page 15
MIC24051
3.0
PIN DESCRIPTIONS
The descriptions of the pins are listed in Table 3-1.
TABLE 3-1:
PIN FUNCTION TABLE
Pin Number
Pin Name
Description
1
PVDD
5V Internal Linear Regulator output. PVDD supply is the power MOSFET gate drive
supply voltage and created by internal LDO from VIN. When VIN < +5.5V, PVDD should
be tied to PVIN pins. A 2.2 µF ceramic capacitor from the PVDD pin to PGND (Pin 2)
must be place next to the IC.
2, 5, 6, 7, 8, 21
PGND
Power Ground. PGND is the ground path for the MIC24051 buck converter power
stage. The PGND pins connect to the low-side N-Channel internal MOSFET gate
drive supply ground, the sources of the MOSFETs, the negative terminals of input
capacitors, and the negative terminals of output capacitors. The loop for the power
ground should be as small as possible and separate from the signal ground (SGND)
loop.
3
NC
No connect.
4, 9, 10, 11, 12
SW
Switch Node output. Internal connection for the high-side MOSFET source and
low-side MOSFET drain. Due to the high-speed switching on this pin, the SW pin
should be routed away from sensitive nodes.
13,14,15,
16,17,18,19
PVIN
High-Side N-internal MOSFET Drain Connection input. The PVIN operating voltage
range is from 4.5V to 19V. Input capacitors between the PVIN pins and the Power
Ground (PGND) are required and keep the connection short.
20
BST
Boost output. Bootstrapped voltage to the high-side N-channel MOSFET driver. A
Schottky diode is connected between the PVDD pin and the BST pin. A boost
capacitor of 0.1 μF is connected between the BST pin and the SW pin. Adding a small
resistor at the BST pin can slow down the turn-on time of high-side N-Channel
MOSFETs.
22
CS
Current Sense input. The CS pin senses current by monitoring the voltage across the
low-side MOSFET during the OFF-time. The current sensing is necessary for short
circuit protection. In order to sense the current accurately, connect the low-side
MOSFET drain to SW using a Kelvin connection. The CS pin is also the high-side
MOSFET’s output driver return.
23
SGND
Signal Ground. SGND must be connected directly to the ground planes. Do not route
the SGND pin to the PGND Pad on the top layer (see PCB Layout Recommendations
for details).
24
FB
Feedback input. Input to the transconductance amplifier of the control loop. The FB
pin is regulated to 0.8V. A resistor divider connecting the feedback to the output is
used to adjust the desired output voltage.
25
PG
Power Good output. Open drain output. The PG pin is externally tied with a resistor to
VDD. A high output is asserted when VOUT > 92% of nominal.
26
EN
Enable input. A logic level control of the output. The EN pin is CMOS-compatible.
Logic high = enable, logic low = shutdown. In the off state, supply current of the device
is greatly reduced (typically 5 µA). The EN pin should not be left floating.
27
VIN
Power Supply Voltage input. Requires bypass capacitor to SGND.
28
VDD
5V Internal Linear Regulator output. VDD supply is the supply bus for the IC control
circuit. VDD is created by internal LDO from VIN. When VIN < +5.5V, VDD should be
tied to PVIN pins. A 1 µF ceramic capacitor from the VDD pin to SGND pins must be
place next to the IC.
DS20005658A-page 16
2016 Microchip Technology Inc.
MIC24051
4.0
FUNCTIONAL DESCRIPTION
The MIC24051 is an adaptive ON-time synchronous
step-down DC/DC regulator with an internal 5V linear
regulator and a Power Good (PG) output. It is designed
to operate over a wide input voltage range from 4.5V to
19V and provides a regulated output voltage at up to 6A
of output current. An adaptive ON-time control scheme
is employed in to obtain a constant switching frequency
and to simplify the control compensation. Overcurrent
protection is implemented without the use of an
external sense resistor. The device includes an internal
soft-start function which reduces the power supply
input surge current at start-up by controlling the output
voltage rise time.
4.1
Theory of Operation
The MIC24051 operates in a continuous mode as
shown in the Block Diagram.
4.2
Continuous Mode
In continuous mode, the output voltage is sensed by
the MIC24051 feedback pin FB via the voltage divider
R1 and R2, and compared to a 0.8V reference voltage
VREF at the error comparator through a low gain
transconductance (gm) amplifier. If the feedback
voltage decreases and the output of the gm amplifier is
below 0.8V, then the error comparator will trigger the
control logic and generate an ON-time period. The
ON-time period length is predetermined by the “FIXED
tON ESTIMATION” circuitry:
EQUATION 4-1:
V OUT
t ON ESTIMATED = --------------------------------V IN 600kHz
Where:
VOUT
VIN
Output voltage
Power stage input voltage
The maximum duty cycle is obtained from the 300 ns
tOFF(min):
EQUATION 4-2:
t S – t OFF MIN
D MAX = ---------------------------------- = 1 – 300ns
--------------tS
tS
Where:
tS
1/600 kHz = 1.66 µs
It is not recommended to use MIC24051 with a
OFF-time close to tOFF(min) during steady-state
operation. Also, as VOUT increases, the internal ripple
injection will increase and reduce the line regulation
performance. Therefore, the maximum output voltage
of the MIC24051 should be limited to 5.5V and the
maximum external ripple injection should be limited to
200 mV. Please refer to Setting Output Voltage in the
Application Information section for more details.
The actual ON-time and resulting switching frequency
will vary with the part-to-part variation in the rise and fall
times of the internal MOSFETs, the output load current,
and variations in the VDD voltage. Also, the minimum
tON results in a lower switching frequency in high VIN to
VOUT applications, such as 18V to 1.0V. The minimum
tON measured on the MIC24051 evaluation board is
about 100 ns. During load transients, the switching
frequency is changed due to the varying OFF-time.
To illustrate the control loop operation, we will analyze
both the steady-state and load transient scenarios.
Figure 4-1 shows the MIC24051 control loop timing
during steady-state operation. During steady-state, the
gm amplifier senses the feedback voltage ripple, which
is proportional to the output voltage ripple and the
inductor current ripple, to trigger the ON-time period.
The ON-time is predetermined by the tON estimator.
The termination of the OFF-time is controlled by the
feedback voltage. At the valley of the feedback voltage
ripple, which occurs when VFB falls below VREF, the
OFF period ends and the next ON-time period is
triggered through the control logic circuitry.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases
and the output of the gm amplifier is below 0.8V, the
ON-time period is triggered and the OFF-time period
ends. If the OFF-time period determined by the
feedback voltage is less than the minimum OFF-time
tOFF(min), which is about 300 ns, the MIC24051 control
logic will apply the tOFF(min) instead. tOFF(min) is
required to maintain enough energy in the boost
capacitor (CBST) to drive the high-side MOSFET.
2016 Microchip Technology Inc.
DS20005658A-page 17
MIC24051
sensed by the gm amplifier and the error comparator.
The recommended feedback voltage ripple is
20 mV~100 mV. If a low-ESR output capacitor is
selected, then the feedback voltage ripple may be too
small to be sensed by the gm amplifier and the error
comparator. Also, the output voltage ripple and the
feedback voltage ripple are not necessarily in phase
with the inductor current ripple if the ESR of the output
capacitor is very low. In these cases, ripple injection is
required to ensure proper operation. Please refer to
Ripple Injection in the Application Information section
for more details about the ripple injection technique.
4.3
FIGURE 4-1:
Timing.
MIC24051 Control Loop
Figure 4-2 shows the operation of the MIC24051 during
a load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to
trigger an ON-time period. At the end of the ON-time
period, a minimum OFF-time tOFF(min) is generated to
charge CBST because the feedback voltage is still
below VREF. Then, the next ON-time period is triggered
due to the low feedback voltage. Therefore, the
switching frequency changes during the load transient,
but returns to the nominal fixed frequency once the
output has stabilized at the new load current level. With
the varying duty cycle and switching frequency, the
output recovery time is fast and the output voltage
deviation is small in MIC24051 converter.
VDD Regulator
The MIC24051 provides a 5V regulated output for input
voltage VIN ranging from 5.5V to 19V. When VIN < 5.5V,
VDD should be tied to PVIN pins to bypass the internal
linear regulator.
4.4
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time.
The input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC24051 implements an internal digital soft-start
by making the 0.8V reference voltage VREF ramp from
0 to 100% in about 3 ms with 9.7 mV steps. Therefore,
the output voltage is controlled to increase slowly by a
stair-case VFB ramp. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same
time or after VIN to make the soft-start function
correctly.
4.5
Current Limit
The MIC24051 uses the RDS(ON) of the internal
low-side power MOSFET to sense over-current
conditions. This method will avoid adding cost, board
space and power losses taken by a discrete current
sense resistor. The low-side MOSFET is used because
it displays much lower parasitic oscillations during
switching than the high-side MOSFET.
FIGURE 4-2:
Response.
MIC24051 Load Transient
Unlike true current-mode control, the MIC24051 uses
the output voltage ripple to trigger an ON-time period.
The output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough. The MIC24051 control loop has the advantage
of eliminating the need for slope compensation.
In each switching cycle of the MIC24051 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. If the peak inductor
current is greater than 11A, then the MIC24051 turns
off the high-side MOSFET and a soft-start sequence is
triggered. This mode of operation is called “hiccup
mode” and its purpose is to protect the downstream
load in case of a hard short. The load current-limit
threshold has a fold-back characteristic related to the
feedback voltage as shown in Figure 4-3.
In order to meet the stability requirements, the
MIC24051 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
DS20005658A-page 18
2016 Microchip Technology Inc.
MIC24051
FIGURE 4-3:
MIC24051 Current-Limit
Foldback Characteristic.
4.6
Power Good (PG)
The Power Good (PG) pin is an open drain output
which indicates logic high when the output is nominally
92% of its steady state voltage. A pull-up resistor of
more than 10 kΩ should be connected from PG to VDD.
4.7
MOSFET Gate Drive
The Block Diagram shows a bootstrap circuit,
consisting of D1 (a Schottky diode is recommended)
and CBST. This circuit supplies energy to the high-side
drive circuit. Capacitor CBST is charged, while the
low-side MOSFET is on, and the voltage on the SW pin
is approximately 0V. When the high-side MOSFET
driver is turned on, energy from CBST is used to turn the
MOSFET on. As the high-side MOSFET turns on, the
voltage on the SW pin increases to approximately VIN.
Diode D1 is reverse biased and CBST floats high while
continuing to keep the high-side MOSFET on. The bias
current of the high-side driver is less than 10 mA, so a
0.1 μF to 1 μF is sufficient to hold the gate voltage with
minimal droop for the power stroke (high-side
switching) cycle, i.e. ∆BST = 10 mA x 1.67 μs/0.1 μF =
167 mV. When the low-side MOSFET is turned back
on, CBST is recharged through D1. A small resistor RG,
which is in series with CBST, can be used to slow down
the turn-on time of the high-side N-channel MOSFET.
The drive voltage is derived from the VDD supply
voltage. The nominal low-side gate drive voltage is VDD
and the nominal high-side gate drive voltage is
approximately VDD – VDIODE, where VDIODE is the
voltage drop across D1. An approximate 30 ns delay
between the high-side and low-side driver transitions is
used to prevent current from simultaneously flowing
unimpeded through both MOSFETs.
2016 Microchip Technology Inc.
DS20005658A-page 19
MIC24051
5.0
APPLICATION INFORMATION
5.1
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine
the peak-to-peak inductor ripple current. Generally,
higher inductance values are used with higher input
voltages. Larger peak-to-peak ripple currents will
increase the power dissipation in the inductor and
MOSFETs. Larger output ripple currents will also
require more output capacitance to smooth out the
larger ripple current. Smaller peak-to-peak ripple
currents require a larger inductance value and
therefore a larger and more expensive inductor. A good
compromise between size, loss and cost is to set the
inductor ripple current to be equal to 20% of the
maximum output current. The inductance value is
calculated in Equation 5-1.
The RMS inductor current is used to calculate the I2R
losses in the inductor.
EQUATION 5-4:
2
I L RMS =
2 I L PP
I OUT MAX + -------------------12
fSW
Switching frequency, 600 kHz
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance.
The high-frequency operation of the MIC24051
requires the use of ferrite materials for all but the most
cost sensitive applications. Lower cost iron powder
cores may be used but the increase in core loss will
reduce the efficiency of the power supply. This is
especially noticeable at low output power. The winding
resistance decreases efficiency at the higher output
current levels. The winding resistance must be
minimized although this usually comes at the expense
of a larger inductor. The power dissipated in the
inductor is equal to the sum of the core and copper
losses. At higher output loads, the core losses are
usually insignificant and can be ignored. At lower
output currents, the core losses can be a significant
contributor. Core loss information is usually available
from the magnetics vendor. Copper loss in the inductor
is calculated by Equation 5-5:
20%
Ratio of AC ripple current to DC output
current
EQUATION 5-5:
EQUATION 5-1:
V OUT V IN MAX – V OUT
L = ---------------------------------------------------------------------------------------V IN MAX f SW 20% I OUT MAX
Where:
VIN(MAX) Maximum power stage input voltage
The peak-to-peak inductor current ripple is:
2
P INDUCTOR CU = I L RMS R WINDING
EQUATION 5-2:
V OUT V IN MAX – V OUT
I L PP = ------------------------------------------------------------------V IN MAX f SW L
The peak inductor current is equal to the average
output current plus one half of the peak-to-peak
inductor current ripple.
EQUATION 5-3:
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating
temperature.
EQUATION 5-6:
R WINDING HT =
R WINDING 20C 1 + 0.0042 T H – T 20C
Where:
I L PK = I OUT MAX + 0.5 I L PP
TH
T20C
Temperature of wire under full load
Ambient temperature
RWINDING(20C) Room temperature winding
resistance (usually specified by
the manufacturer)
DS20005658A-page 20
2016 Microchip Technology Inc.
MIC24051
5.2
Output Capacitor Selection
The type of the output capacitor is usually determined
by its equivalent series resistance (ESR). Voltage and
RMS current capability are two other important factors
for selecting the output capacitor. Recommended
capacitor types are ceramic, low-ESR aluminum
electrolytic, OS-CON and POSCAP. The output
capacitor’s ESR is usually the main cause of the output
ripple. The output capacitor ESR also affects the
control loop from a stability point of view.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated in Equation 5-9:
EQUATION 5-9:
I L PP
I COUT RMS = ----------------12
The maximum value of ESR is calculated:
EQUATION 5-7:
V OUT PP
ESR COUT --------------------------I L PP
The power dissipated in the output capacitor is:
EQUATION 5-10:
P DISS COUT = I COUT RMS ESR COUT
Where:
∆VOUT(PP)
∆IL(PP)
Peak-to-peak output voltage ripple
Peak-to-peak inductor current ripple
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 5-8:
EQUATION 5-8:
V OUT PP =
2
I L PP
------------------------------------- + I L PP ESR COUT 2
C OUT f SW 8
Where:
COUT
fSW
Output capacitance value
Switching frequency
As described in the Theory of Operation section, the
MIC24051 requires at least 20 mV peak-to-peak ripple
at the FB pin to make the gm amplifier and the error
comparator behave properly. Also, the output voltage
ripple should be in phase with the inductor current.
Therefore, the output voltage ripple caused by the
output capacitors value should be much smaller than
the ripple caused by the output capacitor ESR. If
low-ESR capacitors, such as ceramic capacitors, are
selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the Ripple
Injection section for more details.
2016 Microchip Technology Inc.
5.3
Input Capacitor Selection
The input capacitor for the power stage input VIN
should be selected for ripple current rating and voltage
rating. Tantalum input capacitors may fail when
subjected to high inrush currents, caused by turning the
input supply on. A tantalum input capacitor’s voltage
rating should be at least two times the maximum input
voltage to maximize reliability. Aluminum electrolytic,
OS-CON, and multilayer polymer film capacitors can
handle the higher inrush currents without voltage
de-rating. The input voltage ripple will primarily depend
on the input capacitor’s ESR. The peak input current is
equal to the peak inductor current, so:
EQUATION 5-11:
V IN = I L PK ESR CIN
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
EQUATION 5-12:
I CIN RMS I OUT MAX D 1 – D
DS20005658A-page 21
MIC24051
The power dissipated in the input capacitor is:
2.
Inadequate ripple at the feedback voltage due to
the small ESR of the output capacitors.
EQUATION 5-13:
L
SW
P DISS CIN = I CIN RMS ESR CIN
COUT
R1
MIC24051 FB
Cff
ESR
R2
5.4
FIGURE 5-2:
Ripple Injection
The VFB ripple required for proper operation of the
MIC24051 gm amplifier and error comparator is 20 mV
to 100 mV. However, the output voltage ripple is
generally designed as 1% to 2% of the output voltage.
For a low output voltage, such as a 1V, the output
voltage ripple is only 10 mV to 20 mV, and the feedback
voltage ripple is less than 20 mV. If the feedback
voltage ripple is so small that the gm amplifier and error
comparator can’t sense it, then the MIC24051 will lose
control and the output voltage is not regulated. In order
to have some amount of VFB ripple, a ripple injection
method is applied for low output voltage ripple
applications.
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1.
Inadequate Ripple at FB.
The output voltage ripple is fed into the FB pin through
a feed-forward capacitor Cff in this situation, as shown
in Figure 5-2. The typical Cff value is between 1 nF and
100 nF. With the feed-forward capacitor, the feedback
voltage ripple is very close to the output voltage ripple:
EQUATION 5-15:
V FB PP ESR I L PP
3.
Virtually no ripple at the FB pin voltage due to
the very low ESR of the output capacitors.
Enough ripple at the feedback voltage due to the
large ESR of the output capacitors.
L
SW
Cinj
SW
MIC24051 FB
L
MIC24051 FB
COUT
R1
COUT
R1
R2
Cff
R2
ESR
ESR
FIGURE 5-3:
FIGURE 5-1:
Rinj
Enough Ripple at FB.
As shown in Figure 5-1, the converter is stable without
any ripple injection. The feedback voltage ripple is:
Invisible Ripple at FB.
In this situation, the output voltage ripple is less than
20 mV. Therefore, additional ripple is injected into the
FB pin from the switching node SW via a resistor RINJ
and a capacitor CINJ, as shown in Figure 5-3. The
injected ripple is:
EQUATION 5-14:
EQUATION 5-16:
R2
V FB PP = -------------------- ESR COUT I L PP
R1 + R2
Where:
∆IL(PP)
Peak-to-peak value of the inductor
current ripple
1
V FB PP = V IN K DIV D 1 – D ----------------f SW
Where:
VIN
D
fSW
τ
DS20005658A-page 22
Power stage input voltage
Duty cycle
Switching frequency
(R1//R2//RINJ) × Cff
2016 Microchip Technology Inc.
MIC24051
5.5
EQUATION 5-17:
R1//R2 K DIV = ---------------------------------R INJ + R1//R2
Setting Output Voltage
The MIC24051 requires two resistors to set the output
voltage as shown in Figure 5-4.
The output voltage is determined by Equation 5-21:
EQUATION 5-21:
In Equation 5-16 and Equation 5-17, it is assumed that
the time constant associated with Cff must be much
greater than the switching period:
V OUT = V FB 1 + R1
-------
R2
EQUATION 5-18:
1 - = T
------------------ « 1
f SW
If the voltage divider resistors R1 and R2 are in the kΩ
range, a Cff of 1 nF to 100 nF can easily satisfy the
large time constant requirements. Also, a 100 nF
injection capacitor CINJ is used in order to be
considered as short for a wide range of the
frequencies.
VFB equals 0.8V. A typical value of R1 can be between
3 kΩ and 10 kΩ. If R1 is too large, it may allow noise to
be introduced into the voltage feedback loop. If R1 is
too small, it will decrease the efficiency of the power
supply, especially at light loads. Once R1 is selected,
R2 can be calculated using:
EQUATION 5-22:
V FB R1
R2 = ----------------------------V OUT – V FB
The process of sizing the ripple injection resistor and
capacitors is:
1.
2.
Select Cff to feed all output ripples into the feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is
1 nF to 100 nF if R1 and R2 are in kΩ range.
Select RINJ according to the expected feedback
voltage ripple using Equation 5-19.
EQUATION 5-19:
f SW
V FB PP
K DIV = ----------------------- ---------------------------V IN
D 1 – D
Then the value of RINJ is obtained as:
EQUATION 5-20:
1 - – 1
R INJ = R1//R2 ----------K
DIV
3.
FIGURE 5-4:
Configuration.
Voltage Divider
In addition to the external ripple injection added at the
FB pin, internal ripple injection is added at the inverting
input of the comparator inside the MIC24051, as shown
in Figure 5-5. The inverting input voltage VINJ is
clamped to 1.2V. As VOUT is increased, the swing of
VINJ will be clamped. The clamped VINJ reduces the
line regulation because it is reflected as a DC error on
the FB terminal. Therefore, the maximum output
voltage of the MIC24051 should be limited to 5.5V to
avoid this problem.
Select CINJ as 100 nF, which could be considered as short for a wide range of the frequencies.
2016 Microchip Technology Inc.
DS20005658A-page 23
MIC24051
FIGURE 5-5:
5.6
Internal Ripple Injection.
Thermal Measurements
Measuring the IC’s case temperature is recommended
to ensure it is within its operating limits. Although this
might seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared
thermometer. If a thermal couple wire is used, it must
be constructed of 36 gauge wire or higher then (smaller
wire size) to minimize the wire heat-sinking effect. In
addition, the thermal couple tip must be covered in
either thermal grease or thermal glue to make sure that
the thermal couple junction is making good contact with
the case of the IC. Omega brand thermal couple
(5SC-TT-K-36-36) is adequate for most applications.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, an IR
thermometer from Optris has a 1 mm spot size, which
makes it a good choice for measuring the hottest point
on the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
DS20005658A-page 24
2016 Microchip Technology Inc.
MIC24051
6.0
PCB LAYOUT
RECOMMENDATIONS
To minimize EMI and output noise, follow these layout
recommendations.
PCB layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
The following guidelines should be followed to insure
proper operation of the MIC24051 regulator.
6.1
IC
• A 2.2 µF ceramic capacitor, which is connected to
the PVDD pin, must be located right at the IC. The
PVDD pin is very noise sensitive and placement of
the capacitor is very critical. Use wide traces to
connect to the PVDD and PGND pins.
• A 1 µF ceramic capacitor must be placed right
between VDD and the signal ground SGND. The
SGND must be connected directly to the ground
planes. Do not route the SGND pin to the PGND
Pad on the top layer.
• Place the IC close to the point-of-load (POL).
• Use fat traces to route the input and output power
lines.
• Signal and power grounds should be kept
separate and connected at only one location.
6.2
Input Capacitor
• Place the input capacitors on the same side of the
board and as close to the IC as possible.
• Keep both the PVIN pin and PGND connections
short.
• Place several vias to the ground plane close to
the input capacitor ground terminal.
• Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
• Do not replace the ceramic input capacitor with
any other type of capacitor. Any type of capacitor
can be placed in parallel with the input capacitor.
• If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended
for switching regulator applications and the
operating voltage must be derated by 50%.
• In “Hot-Plug” applications, a Tantalum or
Electrolytic bypass capacitor must be used to limit
the over-voltage spike seen on the input supply
with power is suddenly applied.
6.3
to the inductor.
• Keep the switch node (SW) away from the
feedback (FB) pin.
• The CS pin should be connected directly to the
SW pin to accurate sense the voltage across the
low-side MOSFET.
• To minimize noise, place a ground plane
underneath the inductor.
• The inductor can be placed on the opposite side
of the PCB with respect to the IC. It does not
matter whether the IC or inductor is on the top or
bottom as long as there is enough air flow to keep
the power components within their temperature
limits. The input and output capacitors must be
placed on the same side of the board as the IC.
6.4
Output Capacitor
• Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
• Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the user guide.
• The feedback trace should be separate from the
power trace and connected as close as possible
to the output capacitor. Sensing a long
high-current load trace can degrade the DC load
regulation.
6.5
Optional RC Snubber
• Place the RC snubber on either side of the board
and as close to the SW pin as possible.
Inductor
• Keep the inductor connection to the switch node
(SW) short.
• Do not route any digital lines underneath or close
2016 Microchip Technology Inc.
DS20005658A-page 25
MIC24051
7.0
PACKAGING INFORMATION
7.1
Package Marking Information
28-Pin QFN*
XXX
XXXXXXXX
WNNN
Legend: XX...X
Y
YY
WW
NNN
e3
*
Example
MIC
24051YJL
6420
Product code or customer-specific information
Year code (last digit of calendar year)
Year code (last 2 digits of calendar year)
Week code (week of January 1 is week ‘01’)
Alphanumeric traceability code
Pb-free JEDEC® designator for Matte Tin (Sn)
This package is Pb-free. The Pb-free JEDEC designator ( e3 )
can be found on the outer packaging for this package.
●, ▲, ▼ Pin one index is identified by a dot, delta up, or delta down (triangle
mark).
Note:
In the event the full Microchip part number cannot be marked on one line, it will
be carried over to the next line, thus limiting the number of available
characters for customer-specific information. Package may or may not include
the corporate logo.
Underbar (_) and/or Overbar (⎯) symbol may not be to scale.
DS20005658A-page 26
2016 Microchip Technology Inc.
MIC24051
28-Pin 5 mm x 6 mm QFN Package Outline and Recommended Land Pattern
Note:
For the most current package drawings, please see the Microchip Packaging Specification located at
http://www.microchip.com/packaging.
2016 Microchip Technology Inc.
DS20005658A-page 27
MIC24051
DS20005658A-page 28
2016 Microchip Technology Inc.
MIC24051
APPENDIX A:
REVISION HISTORY
Revision A (November 2016)
• Converted Micrel document MIC24051 to Microchip data sheet DS20005658A.
• Minor text changes throughout.
• Vertical axis description updated in Figure 2-9.
• Labeling of Figure 2-42 corrected VIN to VDD.
• Corrected a naming error in Equation 5-6.
• Corrected a formatting error in Equation 5-17.
2016 Microchip Technology Inc.
DS20005658A-page 29
MIC24051
NOTES:
DS20005658A-page 30
2016 Microchip Technology Inc.
MIC24051
PRODUCT IDENTIFICATION SYSTEM
To order or obtain information, e.g., on pricing or delivery, contact your local Microchip representative or sales office.
PART NO.
Device
XX
X
–
Examples:
X.X
a) MIC24051YJL-TR:
Temperature Package Media Type
Device:
MIC24051:
12V, 6A High-Efficiency Buck Regulator
with Hyper Speed Control
Temperature:
Y
=
–40°C to +125°C (Industrial)
Package:
JL
=
28-Lead 5 mm x 6 mm QFN
Media Type:
TR
=
1,000/Reel
2016 Microchip Technology Inc.
Note 1:
12V, 6A High-Efficiency Buck
Regulator with Hyper Speed
Control, –40°C to +125°C
Temperature Range, 28-Lead
QFN, 1,000/Reel
Tape and Reel identifier only appears in the
catalog part number description. This identifier is
used for ordering purposes and is not printed on
the device package. Check with your Microchip
Sales Office for package availability with the
Tape and Reel option.
DS20005658A-page 31
MIC24051
NOTES:
DS20005658A-page 32
2016 Microchip Technology Inc.
Note the following details of the code protection feature on Microchip devices:
•
Microchip products meet the specification contained in their particular Microchip Data Sheet.
•
Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the
intended manner and under normal conditions.
•
There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our
knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data
Sheets. Most likely, the person doing so is engaged in theft of intellectual property.
•
Microchip is willing to work with the customer who is concerned about the integrity of their code.
•
Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not
mean that we are guaranteeing the product as “unbreakable.”
Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our
products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts
allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.
Information contained in this publication regarding device
applications and the like is provided only for your convenience
and may be superseded by updates. It is your responsibility to
ensure that your application meets with your specifications.
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2016 Microchip Technology Inc.
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GestIC is a registered trademarks of Microchip Technology
Germany II GmbH & Co. KG, a subsidiary of Microchip
Technology Inc., in other countries.
All other trademarks mentioned herein are property of their
respective companies.
© 2016, Microchip Technology Incorporated, Printed in the
U.S.A., All Rights Reserved.
ISBN: 978-1-5224-1076-8
DS20005658A-page 33
Worldwide Sales and Service
AMERICAS
ASIA/PACIFIC
ASIA/PACIFIC
EUROPE
Corporate Office
2355 West Chandler Blvd.
Chandler, AZ 85224-6199
Tel: 480-792-7200
Fax: 480-792-7277
Technical Support:
http://www.microchip.com/
support
Web Address:
www.microchip.com
Asia Pacific Office
Suites 3707-14, 37th Floor
Tower 6, The Gateway
Harbour City, Kowloon
China - Xiamen
Tel: 86-592-2388138
Fax: 86-592-2388130
Austria - Wels
Tel: 43-7242-2244-39
Fax: 43-7242-2244-393
China - Zhuhai
Tel: 86-756-3210040
Fax: 86-756-3210049
Denmark - Copenhagen
Tel: 45-4450-2828
Fax: 45-4485-2829
India - Bangalore
Tel: 91-80-3090-4444
Fax: 91-80-3090-4123
Finland - Espoo
Tel: 358-9-4520-820
Atlanta
Duluth, GA
Tel: 678-957-9614
Fax: 678-957-1455
Hong Kong
Tel: 852-2943-5100
Fax: 852-2401-3431
Australia - Sydney
Tel: 61-2-9868-6733
Fax: 61-2-9868-6755
China - Beijing
Tel: 86-10-8569-7000
Fax: 86-10-8528-2104
Austin, TX
Tel: 512-257-3370
China - Chengdu
Tel: 86-28-8665-5511
Fax: 86-28-8665-7889
Boston
Westborough, MA
Tel: 774-760-0087
Fax: 774-760-0088
China - Chongqing
Tel: 86-23-8980-9588
Fax: 86-23-8980-9500
Chicago
Itasca, IL
Tel: 630-285-0071
Fax: 630-285-0075
Dallas
Addison, TX
Tel: 972-818-7423
Fax: 972-818-2924
Detroit
Novi, MI
Tel: 248-848-4000
Houston, TX
Tel: 281-894-5983
Indianapolis
Noblesville, IN
Tel: 317-773-8323
Fax: 317-773-5453
Tel: 317-536-2380
Los Angeles
Mission Viejo, CA
Tel: 949-462-9523
Fax: 949-462-9608
Tel: 951-273-7800
Raleigh, NC
Tel: 919-844-7510
New York, NY
Tel: 631-435-6000
San Jose, CA
Tel: 408-735-9110
Tel: 408-436-4270
Canada - Toronto
Tel: 905-695-1980
Fax: 905-695-2078
DS20005658A-page 34
China - Dongguan
Tel: 86-769-8702-9880
China - Guangzhou
Tel: 86-20-8755-8029
China - Hangzhou
Tel: 86-571-8792-8115
Fax: 86-571-8792-8116
China - Hong Kong SAR
Tel: 852-2943-5100
Fax: 852-2401-3431
China - Nanjing
Tel: 86-25-8473-2460
Fax: 86-25-8473-2470
China - Qingdao
Tel: 86-532-8502-7355
Fax: 86-532-8502-7205
China - Shanghai
Tel: 86-21-3326-8000
Fax: 86-21-3326-8021
China - Shenyang
Tel: 86-24-2334-2829
Fax: 86-24-2334-2393
China - Shenzhen
Tel: 86-755-8864-2200
Fax: 86-755-8203-1760
India - New Delhi
Tel: 91-11-4160-8631
Fax: 91-11-4160-8632
India - Pune
Tel: 91-20-3019-1500
Japan - Osaka
Tel: 81-6-6152-7160
Fax: 81-6-6152-9310
Japan - Tokyo
Tel: 81-3-6880- 3770
Fax: 81-3-6880-3771
Korea - Daegu
Tel: 82-53-744-4301
Fax: 82-53-744-4302
Korea - Seoul
Tel: 82-2-554-7200
Fax: 82-2-558-5932 or
82-2-558-5934
Malaysia - Kuala Lumpur
Tel: 60-3-6201-9857
Fax: 60-3-6201-9859
Malaysia - Penang
Tel: 60-4-227-8870
Fax: 60-4-227-4068
Philippines - Manila
Tel: 63-2-634-9065
Fax: 63-2-634-9069
Singapore
Tel: 65-6334-8870
Fax: 65-6334-8850
Taiwan - Hsin Chu
Tel: 886-3-5778-366
Fax: 886-3-5770-955
Taiwan - Kaohsiung
Tel: 886-7-213-7830
China - Wuhan
Tel: 86-27-5980-5300
Fax: 86-27-5980-5118
Taiwan - Taipei
Tel: 886-2-2508-8600
Fax: 886-2-2508-0102
China - Xian
Tel: 86-29-8833-7252
Fax: 86-29-8833-7256
Thailand - Bangkok
Tel: 66-2-694-1351
Fax: 66-2-694-1350
France - Paris
Tel: 33-1-69-53-63-20
Fax: 33-1-69-30-90-79
France - Saint Cloud
Tel: 33-1-30-60-70-00
Germany - Garching
Tel: 49-8931-9700
Germany - Haan
Tel: 49-2129-3766400
Germany - Heilbronn
Tel: 49-7131-67-3636
Germany - Karlsruhe
Tel: 49-721-625370
Germany - Munich
Tel: 49-89-627-144-0
Fax: 49-89-627-144-44
Germany - Rosenheim
Tel: 49-8031-354-560
Israel - Ra’anana
Tel: 972-9-744-7705
Italy - Milan
Tel: 39-0331-742611
Fax: 39-0331-466781
Italy - Padova
Tel: 39-049-7625286
Netherlands - Drunen
Tel: 31-416-690399
Fax: 31-416-690340
Norway - Trondheim
Tel: 47-7289-7561
Poland - Warsaw
Tel: 48-22-3325737
Romania - Bucharest
Tel: 40-21-407-87-50
Spain - Madrid
Tel: 34-91-708-08-90
Fax: 34-91-708-08-91
Sweden - Gothenberg
Tel: 46-31-704-60-40
Sweden - Stockholm
Tel: 46-8-5090-4654
UK - Wokingham
Tel: 44-118-921-5800
Fax: 44-118-921-5820
2016 Microchip Technology Inc.
11/07/16
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